In the process of adding a PC/controls, and other related instruments, reorganized items in the lab. Threw out some boxes and stored cabling and unused power dock. Moved the sticky mat and put out large trash bin. Organized electronics rack to which a Sorensen (DCS33-33) power supply was attached. For this, took a 14 AWG wire (should be fine up to 15 A at 115 VAC) and cut plug end. Then connect neutral and live as indicated by the rear of the panel and add chassis ground. Tested DC output voltage of 3 V and it works ok.

There are now two workstations in the lab attached to the same monitor (VGA and DVI ports), and it is ok to ssh from one to the other. They both now have fresh debian 10 installs.

After getting what looked like a decent cavity reflection signal, installed RFPD yesterday. For this, removed the lens that was right before the PD because the RFPD area is large enough, but keep ND filter in place. Powered with +- 18 VDC and monitor DC out on the scope, and RF out is sent to the IF of the mixer in the PDH box. For the LO, split the Marconi RF output and connected the demodulated signal into Ch2 of the scope in hopes that there was an error signal.

A hint of the error signal is present (blue trace below), although deeply buried in line noise (and harmonics up to ~180 Hz) so there really are two things to optimize now -->

1. Line noise (hunting for ground loops or equipment, e.g. power supplies, analyze LO spectrum before/after splitters, mixers, etc...)
2. Mode matching (this was the first reaction, as improving the cav refl SNR by means of mode matching makes a better pdh err signal)

Other things attempted so far -->

• Switched mixers, splitter, and RF cables
• Bypass the phase shifter completely
• Scan LO phase
• Floated RFPD power supply
• Floated PDH box power supply (really only affecting the phase shifter if anything, though unlikely to matter at this point)

Locking update

The plan during these past few days has been to have fast control loop of the cavity (locked to laser using PZT, which succeeded using SR560s), and slow control loop where the laser temp. actuator is fed back the integrated PZT input to follow the long term cavity drift. For that, have been messing around with the high-level (GUI) API of PyRPL, with basically no success. In fact currently the RedPitaya cannot even replace the SR560 fast controls, which probably has to do with the +- 1 Volt limits on the RP input/output.

Another issue is that any loop gain depends on the REFL power, which will be at some point slowly ramped up to cross the OPO operating threshold, and while there is a (PBS + HWP) knob on how much light is hitting the RFPD, the lock is not yet good enough to keep up with the slow human action.

First light from nonlinear conversion

WIth the cavity locked, and under ~ 220 mW of pump (right before the cavity, i.e. 1.3 Amps of current on the driver), noticed a tiny green dot coming from within the crystal oven. This is pretty great news in terms of phase matching, but not necessarily so in terms of the right parametric conversion process (understanding is that SHG is easier to attain even with single pass). See tiny green spot as caught using phone camera in the attachment.

Another Heimann Sensor / Boston Electronics delivered to Paco.
This unit (purchased May 2020/ / Delivered Aug 5th, 2020) has a FZ-Si window on it.
We don't know how it is.

See http://nodus.ligo.caltech.edu:8080/Mariner/121

How to move the large engine hoist through the narrow door

See http://nodus.ligo.caltech.edu:8080/Mariner/122

I wanted to push the limits and see when NN subtraction performance starts to break by changing the number of seismometers and the size of the array. For aLIGO, 10 seismometers in a doubly-wound spiral around the test mass with outer radius 8m is definitely ok. Only if I simulate a seismic field that is stronger by a factor 20 than the 90 percentile curve observed at LHO does it start to get problematic. The subtraction residuals in this case look like

The 20 seismometer spiral is still good, but the 10 seismometer spiral does not work anymore. It gets even worse when you consider arrays with circular shape (and one seismometer at the center near the test mass):

This result is in agreement with previous results that circular arrays have trouble in general to subtract NN from locally generated seismic waves or seismic transients (wavelets).

I should emphasize that the basic assumption is that I know what the minimum seismic wavelength is. Currently I associate the minimum wavelength with a Rayleigh overtone, but scattering could make a difference. It is possible that there are scattered waves with significantly smaller wavelength.

The attached PDF shows the thermal noise of a short, low mass silicon cavity.

The thermal noise uses the gwincdev BQuad suspension noise term, I've disabled all the stages except the last (so it is a simple pendulum). The frequency noise is the noise budget of Dmass' silicon refcav.

The mass is 10g, the cavity length is 0.1m. The ribbons are 2mm by 0.05mm, 0.5m long.

I made them so thin because for some reason the gwinc model starts going nuts with the violin modes at high frequencies. (you can see that above ~150Hz). The frequency where it goes nuts comes down lower if I make them thicker. I don't know what's happening in the model but I guess it's not physical.

So this experiment looks a little like Thomas Corbitt's cantilever, it's not so similar to what would go in LIGO but it makes the thermal noise big enough to be seen above frequency noise.

In order to accentuate the thermal noise in a silicon test cavity, it would be nice to make the ribbons a little bit thicker. Sadly, the BQuad thermal noise model seems to explode when the fibers get thicker.

The three plots I will show have the following parameters in common:

4 fibers, single pendulum silicon suspension @120K. 10g mirror mass, 5cm fiber length, 2cm cavity length. The fiber width is 2mm and the fiber thickness varies in the three plots.

The first shows a fiber thickness of 0.05mm. The second has 0.1mm, and the third is 0.2mm.

As one can see, the model sort of goes more and more nuts as the thickness is increased. I don't really understand the model enough to know why this is the case, but it seems that to have a believable noise budget we might need to make a thermal noise model from scratch, rather than using gwincdev.

The source for what I've been using to calculate thermal noise.

I made an estimate for frequency noise requirement for a laser that can be used in crackle experiment. With some assumptions, I came up with df = 3x10² [^Hz/_rtHz ] for the requirement.

 The two beams from both arms are recombined at the output port of a Michelson interferometer. If it is operated at dark port, the output signal will be linear with the differential length between the two arms.

some assumptions in the calculation:

• Operating at darkport
• The laser has frequency = f0 + df  (carrier + noise)
• mismatch between the two arms is ~ 1mm
• aim for SNR = 1, no integration time.
• crackle signal is ~10^-15 m/rtHz, this is actually the shot noise limit of the current setup.

This will be a requirement for the planned ecdl.

Is a HeNe laser good enough? I'm not sure about HeNe frequency noise level, and I haven't found it in literature that much. I checked here,see fig 5, HeNe f noise is not so bad compared to NPRO noise (10^4 /f Hz/rtHz).This feels a bit counter intuitive. But if it is real, it should be ok for the measurement around 100 Hz and above.

you have to overlay the estimated displacemnt noise with the existing L1 noise bud or else we cant tell what the importance of the result is

There we go. Based on the noisemon data at L1 and H1, I calculated the DAC noises at those sites, using roughly the same approach as described in 1847.

I used the coherence between the master channel and the noisemon channel to calculate the total noise going into the coils.

Then I converted the ADC noise and noisemon noise to DAC volts and subtracted them from the total noise. I compared the result of the subtraction, which should be DAC noise, at least in the passband (20-100Hz), with the G1401399 model and made a noise budget, shown in attachment 1. We can see that, as designed, the DAC noise is sufficiently amplified so that it dominates over the noisemon noise or the ADC noise in the passband.

Next, I projected the DAC noise to strain noise and summed them up for all the four channels in all the four stations.

Finally, I compared this with the interferometer noise spectrum based on data in L1:OAF-CAL_DARM_DQ and H1:CAL-DELTAL_EXTERNAL_DQ. I calibrated these data with calibration files here. The results are shown in attachment 3. All the data and scripts are included in attachment 4, where analysis.py is the script that does the job. Based on the plots, it seems DAC noise could be potentially a limiting factor for the interferomter sensitivity.

The coil driver states for L1 is LP off, ACQ off (state 1). For H1 is LP on, ACQ off. The LISO files calculating the current transfer functions and the voltage transfer functions are attached in attachment 4. 

I used a resolution of 1mHz in the diaggui measurement. The data files are too large so I can not upload them here. I am figuring out what to do.

Note: I fell into a few traps during the calculation. Many of them was about data and transfer functions. I have been more careful about what data is used in these calculations. For example, the noisemon data downloaded from the sites when MASTER was off still has DAC noise in it. I thought it was ADC noise + noisemon noise before and used it for subtraction. Another example, the transfer function measured at the sites has all the noise in it. We do not see the noises in the passband but ADC noise dominates at high frequencies. If you use this transfer function to figure out how much noisemon noise contributes, you result will be tampered by the noises, like ADC noises at high frequency. Last example, if you use the noisemon noise data measured in the digital system in our lab, you should be aware that, although it does not have DAC noise (I disconnected DAC when measuring the noises), it also has ADC noise. Therefore, it would be better to use data from SR785 or LISO simulations (which has been shown to agree with each other). I drew a diagram in attachment 2 to help thinking about what data or transfer functions should be used. 

for some reason the DAC noise estimate is too high, it can't really be so large compared to the real DARM curve (see the noise budget curves from LLO - there are other noise sources besides DAC noise)

some possibilities:

1. maybe the DAC noise calibration into meters is wrong? I can't tell from the code where this came from. It would be good to put a comment in there.
2. perhaps most of this noise is actually angular noise. The ASC control signals are adjusted by tuning the digital coefficients (before MASTER_OUT) so that the angle to length coupling is minimized. I think something like this has to be done to remove the angular noise from the DAC noise estimate.
3. internal saturation of the DAC noise monitor?

I hve modified the code to plot nicer and also to remove some divide by zero problems. There is also still some warnings about other divide by zero - those should probably be fixed by examining how better to handle it when the coherence goes to zero.

we only care about PRM, PR2, PR3, BS, SR3, SR2, SRM, so you should eliminate all other optics from the plots.

Then just plot the HSTS to get started and we can see how they compare.

The spectra for damp out for the following optics
- small triples {PR2, PRM, SR2, SRM} (Attachment 1)
- large triples {PR3, SR3} (Attachment 2)
- beamsplitter {BS} (Attachment 3)

For pwelch(), I was not taking into account the DC offset which is why I was getting funky plots (spectral leakage). I corrected for it, no more interpolation anymore!

Edit: I have changed the yaxis label for all plots to counts/rHz for now since I do not know the exact calibration

Can you explain how you calibrated the control signals into displacement units? It would be fine to start if you could just plot them with the y-axis in counts/rHz for now, unless you have good confidence in the calibration.

Small Triple Suspension Trial Filter Design

tldr: From the Damp_OUT spectrum of the small suspensions, PRM had a relatively larger magnitude in the 10-30Hz bandwidth. I thus choose to design filters for them using PRM as a reference.
While designing for PRM, I used the site noise asd (DAMP_IN*(1 + OLG)) given as input to the controller instead of using white noise while desinging them.
The design objectives that were met are as follows:

• Lower noise by a factor ~10 in the 10-30Hz bandwidth or better
• Achieve similar ringdowns for the impulse response
• Have phase margins of ~30 degrees and gain margins of ~3dB or better

The same design principles were used for PRM as for PR3 (explained in this elog). Here’s the filter design for each DOF. (The electronics gain has not been accounted for here, it must be added to the filter design). The filters work with negative feedback with a gain of 1. 
Note: I had an an interesting time designing the Roll Mode, bumping the peaks to increase damping on them after setting the UGF at 2.75 would increase the ringdown period. I reverted to using a similar overall "magnitude" that they used at the sites and added notches for reducing noise in the required bandwidth. The new filter still better on the ringdown (15 seconds lesser) so I'll leave it at that for now and make changes if required later. 

The pdfs have been arranged as following for each DOF. (Comparisons are between the site filter design and the trial filter design (need to come up with a better terminology for this)).

• Attachment 1 shows a comparison of the sensor noise ASD (DOF sensor noise on M1 to DOF Displacement on M3) in the bandwidth of interest and ringdown periods for the impulse response. (FOR Longitudinal DOF, the impulse input was ground, for the remaining DOFs, the input was at M1). 
• Attachement 2 shows a comparison of the filter designs in the bandwidth of interest
• Attachment 3 shows the open loop transfer function (P(s)*C(s)) with phase and gain margin (minimum stability margins)
• Atttachment 4 shows a comparison of the open loop transfer functions (P(s)*C(s)).

Small Triple Suspension Trial Filter Design (SR2 Baseline)

tlpr: Referencing the triple noise breakdown from this alog , SR2 has the highest noise amongst the smaller suspensions. I had designed the trial filters eariler using PRM as a baseline, but since the optics couple against DARM loop differently, I am switching to this new baseline
While designing for SR2, I used the site noise asd (seismic noise + sensor noise = DAMP_IN*(1 + OLG)) given as input to the controller instead of using white noise while desinging them. I had to lose out on the ringdown times this time, still doing comparable to the sites. Phase and gain margins almost at design criterion now.
I've also added bumps to account for the "SECRET COSTS" at low frequencies; however, since I had to reduce the gain on the damping filters, they're relatively lower in magnitude than the site filters (most of these peaks are around the mechanical resonances at the lower frequencies; getting similar features in the ringdown could be a good measure of these bumps doing the job although being lower in magnitude, maybe :$).
The site noise asd was produced using a timeseries obtained from 11/22/2023, 12am PST for a duration of 1 hour
The design objectives that were met are as follows:

• Lower noise by a factor ~10 in the 10-30Hz bandwidth or better
• Achieve similar ringdowns for the impulse response
• Have phase margins of ~30 degrees and gain margins of ~3dB or better

The same design principles were used for SR2 as for PRM (explained in this elog). Here’s the filter design for each DOF. (The electronics gain has not been accounted for here, it must be added to the filter design). The filters work with negative feedback with a gain of 1. 
 

The pdf has been arranged as following for each DOF. (Comparisons are between the existing site filter design and the trial filter design (sticking to this terminology, psych).

• Figure 1 shows a comparison of the sensor noise ASD (DOF sensor noise on M1 to DOF Displacement on M3) in the bandwidth of interest and ringdown periods for the impulse response. (FOR Longitudinal DOF, the impulse input was ground, for the remaining DOFs, the input was at M1). 
• Figure 2 shows a comparison of the filter designs in the bandwidth of interest
• Figure 3 shows the open loop transfer function (P(s)*C(s)) with phase and gain margin (minimum stability margins)
• Figure 4 shows a comparison of the open loop transfer functions (P(s)*C(s)).

• The slotted disks that capture the wires do not have the alignment bore used to center the wire in the hole
  
• We didn't correctly route the far field QPD cable so it runs funny
  
• We didn't have a tool which could be used to get two of the DCPD preamp box mounting screws (which are M3's chub!)
  
• We don't have the cable clamps to tie off the electrical cables to the intermediate mass
  
• We don't have any of the cabling from the OMC-SUS top to the rack so we can't test anything
  
• We haven't uploaded pretty pictures for all to see
  

• The bottom plate of the EQ stops is too thick so that it bumps into the tombstones
  
• The vertical member on the "waist" EQ stops is too close to the breadboard, possibly interfering with the REFL beam
  
• The "waist" EQ stops are made from a thin plate that doesn't have enough thickness to mount helicoils in
  
• Helicoil weren't loaded in the correct bottom EQ stops
  
• The DCPD cable loops over the end EQ stop looking nasty but not actually making contact
  

• ND10 and ND20 neutral density filter
  
• EOM and mount set for 4 inch beam height
  
• Post for fiber launch to get to 4 inch
  
• Mode matching lens at 4in
  
• 3x steering mirror at 4in
  
• RF photodiode at 4in
  
• Post for camera to 4in
  
• Light sheild for camera
  
• Long BNC cable
  

• DitherLock_sweep: Sweep of the IN2/IN1 for the dither lock error point showing 600 Hz UGF
  
• HiResPZTDither_sweep: Sweep of the PZT dither input compared to the PDH error signal. I restarted the front end before the sweep was finished accounting for the blip.
  
• HiResPZTDither_sweep2: Finish of the PZT dither sweep
  

• FIgure 1: PZT dither path. Most of the features in this plot are understood: There is a 2kHz high pass filter in the PZT drive which is otherwise flat. The resonance features above 5 kHz are believed to be the tombstones. I don't understand the extra motion from 1-2 kHz.
  
• Figure 2: PZT dither path zoom in. Since I want to dither the PZT to get an error signal, it helps to know where to dither. The ADC Anti-aliasing filter is a 3rd order butterworth at 10 kHz, so I looked for nice flat places below 10 KHz and settled on 8 kHz as relatively harmless.
  
• Figure 3: PZT LSC path. This path has got a 1^2:10^2 de-whitening stage in the hardware which hasn't been digitally compensated for. You can see its effect between 10 and 40 Hz. The LSC path also has a 160 Hz low path which is visible causing a 1/f between 200 and 500 Hz. I have no idea what the 1 kHz resonant feature is, though I am inclined to point to the PDH loop since that is pretty close to the UGF and there is much gain peaking at that frequency.