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2216   Sat Jul 21 14:17:20 2018 awadeDailyProgress2micronLasersThe case of the DET10D Photodiode, featuring TIA

Cool.  You should also make a plot of the actual noise you measured of the circuit with the SR785 (even though this was only in the 1-100 kHz band).  Maybe we can stick a resistor in series and get a transfer function with the Agilent spectrum analyzer to see if C_F is chosen correctly.  Although I'm not sure if this is the best way to do it.

I've attached the jupyter notebook I made with interactive sliders.  It lets you select your op amp, drag component values up and down on sliders and see how this affects noise, signal bandwidth and OLG of the op amp circuit.  It will require pyliso to run on the backend.

There is a bunch of background theory that is not complete yet, so take it with a grain of salt.  Interactive sliders and buttons stuff is at the bottom.

 Quote: Down in the machine shop we've been developing and modifying a transimpedance amplifier to be used in conjuction with the photodiode in the 2micron experiment, schematic and noise analysis(with PD shot noise as horizontal line!) are shown below. With a dual 9V battery supply, we ran the output through our signal analyzer but quickly noticed the signal's unstable nature- a consequence of the lack of phase margin between the open loop gain and feedback factor. To address this issue and improve the circuit's stability, we simply added a phase compensator(i.e. a capacitor) in parallel with the gain resistor. Its capacitance can be calculated through a straightforward relationship between, 1. the intercept frequency, $f_i$ ,  of the open loop gain curve and the reciprocal of the feedback factor. 2. The pole corner frequency, $f_F$. 3.  The unity-gain bandwidth, $f_{GBWP}$. The intercept frequency can be expressed as, $f_i = \frac{1}{2\pi R_FC_F}$ Where R_f and C_f are the values of gain resistor and phase compensator capacitance respectivel. Additionally, it follow that the pole corner frequency is, $f_F = \frac{1}{2\pi R_F(C_F+C_i)}$ Where C_i is the capacitance of the PD's junction capacitor in parallel with the input capacitance of the op-amp. They are brought together by the following expression: $f_i = \sqrt{f_f*f_{GBWP}}$ Solving for the phase compensator capacitance, we reach, $C_F = \frac{1}{4\pi R_Ff_{GBWP}}(1+\sqrt{1+8\pi C_if_{GBWP}})$ Inputting the values of our circuit's components, we reach a capaticance of 200pF. Fortunately, the TIA proved to be more stable after this modification. Finally, we implemented our TIA circuit into our PD in the 2micron experiment and achieved better readings, which I'll be sure to elaborate more upon tomorrow.

Attachment 1: notebook_pyliso_WOPO1mmTransImp-v3.ipynb.zip
2221   Thu Jul 26 17:49:28 2018 Vinny W.DailyProgress2micronLasersPower Loss through different optic components

In our mission to characterize our 2micron laser, I calculated the changes of power at different points within the experiment- the points are shown in the schematic below. I kept the input current constant at 50.02 mA, and the temperature of the laser diode at 8.657k$\Omega$.

 Location (from laser to...) Power (mW) (error, +/- 0.002 mW) Frequency (Hz) (error, +/- 3Hz) Directly from laser 1.339 796.68 Faraday Isolator 0.894 61.09 Beam Coupler #1, A 0.528 60.09 Beam Coupler #1, B 0.707 61.20 Beam Coupler #2, A 0.762 60.55 Beam Coupler #2, B 0.480 61.40 Longer arm of interferometer 1.291 62.30

The excess power loss, $L$, at either beam splitter can be expressed in dB as:

$L = 10\log{\frac{P_{laser}}{P_{A}+P_{B}}}$

Running this through gives us an excess loss of 0.351dB at Beam Splitter #1 and 0.327dB at Beam Splitter #2.

We're finishing up the thermal sensor to be placed in the ATF! Schematics and pictures will be provided later on today.

Attachment 1: poweranalysis.JPG
2226   Thu Aug 2 18:06:54 2018 awadeDailyProgress2micronLasersThe case of the DET10D Photodiode, featuring TIA

You are missing a factor of R_F under the square root.

$C_\textrm{F} = \frac{1}{4\pi R_\textrm{F}f_\textrm{GBWP}} (1 + \sqrt{1 + 8 \pi R_\textrm{F} C_\textrm{i}f_\textrm{GBWP}})$

 Quote: Down in the machine shop we've been developing and modifying a transimpedance amplifier to be used in conjuction with the photodiode in the 2micron experiment, schematic and noise analysis(with PD shot noise as horizontal line!) are shown below. With a dual 9V battery supply, we ran the output through our signal analyzer but quickly noticed the signal's unstable nature- a consequence of the lack of phase margin between the open loop gain and feedback factor. To address this issue and improve the circuit's stability, we simply added a phase compensator(i.e. a capacitor) in parallel with the gain resistor. Its capacitance can be calculated through a straightforward relationship between, 1. the intercept frequency, $f_i$ ,  of the open loop gain curve and the reciprocal of the feedback factor. 2. The pole corner frequency, $f_F$. 3.  The unity-gain bandwidth, $f_{GBWP}$. The intercept frequency can be expressed as, $f_i = \frac{1}{2\pi R_FC_F}$ Where R_f and C_f are the values of gain resistor and phase compensator capacitance respectivel. Additionally, it follow that the pole corner frequency is, $f_F = \frac{1}{2\pi R_F(C_F+C_i)}$ Where C_i is the capacitance of the PD's junction capacitor in parallel with the input capacitance of the op-amp. They are brought together by the following expression: $f_i = \sqrt{f_f*f_{GBWP}}$ Solving for the phase compensator capacitance, we reach, $C_F = \frac{1}{4\pi R_Ff_{GBWP}}(1+\sqrt{1+8\pi C_if_{GBWP}})$ Inputting the values of our circuit's components, we reach a capaticance of 200pF. Fortunately, the TIA proved to be more stable after this modification. Finally, we implemented our TIA circuit into our PD in the 2micron experiment and achieved better readings, which I'll be sure to elaborate more upon tomorrow.

2228   Mon Aug 6 03:45:37 2018 Vinny W.DailyProgress2micronLasersThermal Sensor Circuit Update

Below are the schematics of the two added sections to the passive temperature sensor circuit we're building. Since we're trying to measure temperature differences/fluctuations throughout the ATF, we'll need to incorporate a voltage reference in our circuit- which is necessary in this analog-to-digital purpose. The addition of a Sallen Key filter in conjunction with the voltage reference is to overcome precision limitations of the component. The large values of the two resistors were chosen to counteract the decline in performace due to the component's output resistance at high frequencies. To characterize the temperature sensor, we ran the circuit through the EE shop's spectrum analyzer at points before the SK and after. I'm not too versed in extracting data from it(I was given a crash course earlier this morning, so I'll try again tomorrow). For now it's good to note that observing a 50Hz bandwidth, the Vrms noise before the SK, measured at the output of the LT1021 7V reference was 17.24nVrms/rtHz at 10Hz, and after the SK turned out to be 16.7nVrms/rtHz at 10Hz (pictured below. Apologies in advance for the picture of data, it's just to have a visual of the noise bandwidth characteristics. I'll have actual plots once I get working with the whole spectrum analyzer data-over-wifi scheme). Additionally, I'd like to update this e-log tomorrow with the transfer functions at both of those stages!

Conceptually, the device has been completed(pictured below, and a special thanks to Andrew for helping me out with molex connections!). At this point further modifications would be to switch out the film resistors for surface mounts to see if there is any improvement.

Attachment 1: scheme_voltageref.JPG
Attachment 2: scheme_thermalsensor.JPG
Attachment 3: 20180806_162441.jpg
Attachment 4: 20180806_135423.jpg
2230   Tue Aug 7 10:47:43 2018 awadeDailyProgress2micronLasersThermal Sensor Circuit Update

Great. A few things:

### Noise ASD

It looks like this spectral density is catching only below the 1/f corner frequency.  What you'll want to do for your actual measurement is to take a bunch of different frequency spans over the full range and stitch them together.  The iris plotting tools allow you to do this pretty easily

> python ~/Git/labutils/iris/iris.py --noTar NameOfBatchFiles*

The '*' is just to grab all the spans you took of that particular run. Then in the same directory something like

> python ~/Git/labutils/iris/iris.py --noTar --noStitch StitchedSpectrum_*

will make a nice plot of all the stitched spans.

### Drift and noise

I've attached a interactive notebook for the SK using LISO as backend to model the circuit.  You can see that the noise is dominated by the non-inverting pin current noise. You could choose a FET op amp.  However, I think the bigger concern is the thermally induced offset drift; for typical op amps this is in the range of 0.2 to 2 µV/K.  However given the signal is order of 7 V then maybe this is only a very small affect on the injected offsetting current: say 2 µV/K /25.6 kΩ = 78 pA/K or equivalently 0.078 mK of input referred drift per degree kelvin of drift in your chip. This would be a concern for precision thermal readout but in your case it is waaaay over spec'ed.  OP27 is probably good enough for now.

As a side note: the AD590 has a noise floor of 40 pA/rtHz, which is an equivalent input temperature noise floor of 1 uK/rtHz.  The readout circuit is not limited by the sensor but you are still much better than your requirements. You should quote the drift and noise in equivalent input units of what you are trying to measure, which is temperature.

You may well end up being dominated by drift in the TIA feedback resistor thermal sensitivity.  This will shift the gain of the circuit and therefore the readout slope of V/K.  Typical thick metal film resistors are about 50-200 ppm/K.  Thin film are on the order of 5-50 ppm/K.

### Schematics

Don't know what is going on the the second one there.  Did you mean to use an AD746? Also, the AD620 is an instrument amplifier, you probably don't want to use it in this application for a TIA.

 Quote: Below are the schematics of the two added sections to the passive temperature sensor circuit we're building. Since we're trying to measure temperature differences/fluctuations throughout the ATF, we'll need to incorporate a voltage reference in our circuit- which is necessary in this analog-to-digital purpose. The addition of a Sallen Key filter in conjunction with the voltage reference is to overcome precision limitations of the component. The large values of the two resistors were chosen to counteract the decline in performace due to the component's output resistance at high frequencies. To characterize the temperature sensor, we ran the circuit through the EE shop's spectrum analyzer at points before the SK and after. I'm not too versed in extracting data from it(I was given a crash course earlier this morning, so I'll try again tomorrow). For now it's good to note that observing a 50Hz bandwidth, the Vrms noise before the SK, measured at the output of the LT1021 7V reference was 17.24nVrms/rtHz at 10Hz, and after the SK turned out to be 16.7nVrms/rtHz at 10Hz (pictured below. Apologies in advance for the picture of data, it's just to have a visual of the noise bandwidth characteristics. I'll have actual plots once I get working with the whole spectrum analyzer data-over-wifi scheme). Additionally, I'd like to update this e-log tomorrow with the transfer functions at both of those stages! Conceptually, the device has been completed(pictured below, and a special thanks to Andrew for helping me out with molex connections!). At this point further modifications would be to switch out the film resistors for surface mounts to see if there is any improvement.

Attachment 1: LISOModel_Sallen-Key.ipynb.zip
Draft   Tue Aug 14 13:44:47 2018 Vinny W.DailyProgress2micronLasersNoise Analyses

With the two 15m optic fibers in, we can start building the thermally-actuated reference interferometer portion of the 2um experiment. To have a re

On the thermal sensor: With it powered up, this signal is from one of the output BNCs.

On the TIA: With it powered up, this signal is from the BNC output (note: the input end was open)

On the TMTF: With the photodetector and TIA powered on, this signal is

2250   Thu Sep 6 10:19:54 2018 awadeDailyProgress2micronLasersThermal Sensor Circuit: Modifications

I had another look at Vinny's AD590 temperature sensor preamplifier.  After debugging and thinking for a bit I made a number of changes that should reduce drift and minimize noise.

## Assessing the noise and drift of OP27 TIA

The main TransImpedance amplifier (TIA) stage is done by an OP27 in each channel.  We chose these for the first pass prototype for their good compromise points in noise/offset etc for general use.  The main issue of this chip in this application is its thermally induced drift and not noise.

Output noise of the AD590 is 40 pA/rtHz, where OP27 has a current noise of 1.7 pA/rtHz at 10 Hz (0.4 pA/rtHz@1kHz), so we should expect the AD590 sensor itself to dominate the current noise. Similarly, voltage noise makes a contribution but only when the sensor's lead capacitance is large(ish) and uncompensated with a feedback capacitor.  With proper compensation the preamplifiers dominant noise contribution is its current noise at the inverting pin of the op amp. For 100 kΩ of TIA gain, 300 pF lead capacitance and 21 pF of feedback capacitance, in the full band (150 kHz) the input referred noise is order 0.6 µK/rtHz with a 1/f knee-point of about 100 Hz.

Drift and offsetting seems to be more of an issue.  For the chips that Vinny had put in I found that, between the channels, they had offsets equivalent to 5ºC between them.  This would mean a voltage offset of 125 mV or current offset of 5 µA.  I suspect these units were recycled from previous undergraduate projects and had previously been abused. I swapped them all out for brand new OP27 (bin G) of same manufacture and found that the DC offset between channels was reduced.  I didn't go much further in quantifying this as it seemed that the longer term drift due to thermal affects on the box was order of a couple of degrees (common between channels) which mean that the chips or the configuration weren't right.

Absolute calibration error in the AD590 (J bin) can be up to 5.0 °C and AD592 (A bin) have a worst case calibration error of 2.5 °C. However swapping sensors between channels showed that this batch of sensors were all about the same.

## Offsetting signal with reference and removing Sallen-Key Filter

It should be added that there is a 273 µA offset provided by a LT1021-7 reference (7 V) to bring output voltage closer to 0V at 0°C. The 7 V output is converted to current via a 25.64 kΩ ± 0.005% (max 50ppm/K) resistor into the inverting pin of the TIA (in parallel with the AD590/AD592 input).  This offsetting also allows us to increase the transimpedance gain without reaching the rails of the TIA stage at room temperature.   Here the 7V reference was chosen as it has the lowest absolute error, hysteresis and drift (2ppm/K).  The 5 V and 10V versions of this voltage reference have some internal resistive divider circuitry to up or down the voltage to the spec value.  The LT1021-7 is the best choice as uses no resistive divider to set output and therefore has the best long term stability.

Voltage noise of the LT1021-7 reference is relatively high at about 70 nV/rtHz and 4.0 uVpp 0.1Hz<f<10Hz.  But when converted to current this is is order 2.7 pA/rtHz.  The shot noise of 273 µA is 9.3 pA/rtHz, so the dominant penalty here for offsetting is actually the shot noise of the current and not the reference.  For this reason I removed the Sallen-Key filter (sorry Vinny) as it was unnecessary: only adding noise and potential offset drift in the circuit. Simpler is better in this case.

## Testing drift

To test drift I took all four sensors -- each with 5 m of cable -- taped them to a ~2.5 kg copper block (with aluminum tape) and placed them in a foam box (pictures below).  These were left to reach equilibrium and then the output of the circuit was logged with an Acromag ADC to track changes.  I'll put up the data from this long term test in another post.

I found that pointing a heat gun directly at the board would induced common drift of the channels of 3-5 degrees.  At a guess this was about a 50 C rise in temperature of the board. That would place the circuit induced apparent drift at (worst case) 0.1°C/°C which isn't great if we actually want to know what is changing in the environment.  I didn't purse the OP27 further because it seemed like this drift was well in excess of what should be expected. We should expect 80 µK/K from op amp current drift, 0.8 µK/K op amp voltage drift and 125 µK/K @ 25°C for transresistance thermal drift for 50 ppm/K thin film resistors: for a total of 0.2 mK/K drift due to the TIA circuit.

Also the offset provided by the LT1021-7 should have given a drift of 0.55 mK/K for the voltage reference itself (2ppm/K) and 14 mK/K? for thermal drift of 25.64 kΩ series resistor (thin film 50 ppm/K worst case).

All up maximum worst case drift should hav been 14.8 mK/K. The main culprit here is the 25.64 kΩ offsetting resistor, if there is a lower thermal gradient resistor this would be the part of the circuit to change out.

## Choosing better TIA op amp

OP27 is not the best possible choice for this application.  We don't need a huge GBWP for these thermal sensing applications and often the lower GBWP chips seem to have better drift specs.  The below table summarizes​ some specs of OP27, OP07 (OP27s lower bandwidth cousin) and LT1012.

Summary of offsets and thermal coefficients op amps*
Parameter OP27 OP07 LT1012
Input voltage offset 30 µV 60 µV 30 µV
Thermal coefficient of input offset 0.4 µV/K 0.5 µV/K 0.2 µV/K
Input current offset 12 nA 0.8 nA 30 pA (250 pA worst)
Thermal coefficient of input offset 80 pA/K 12 pA/K 0.3 pA/K
Long term offset drift 0.4 µV/month 0.4 µV/month  0.4 µV/month
Input bias current ±15 nA ±1.8 nA ±80 pA (±300 pA worst)
Voltage noise

3.8 nV/rtHz @ 10 Hz

(0.09 µV p-p @ 0.1 Hz to 10 Hz)

10.5 nV/rtHz @ 10 Hz

(0.38 μV p-p @ 0.1 Hz to 10 Hz)

17 nV/rtHz @ 10 Hz

(0.5 µVp-p@ 0.1 Hz to 10 Hz)

Current noise 1.7 pA/rtHz @ 10 Hz

0.35 pA/rtHz @ 10 Hz

15 pA p-p

20 fA/rtHz @ 10 Hz

LT1012 seems like a much better choice.

*Some FET based op amps had comparable noise but thermal drift wasn't much better. I didn't look too deep into this, maybe there is a much better FET op amp out there.

## Other issues with the circuit

I also found that unplugging just one of the sensors from the temperature sense box lead to all of the channels having an output change of order 0.3 V (or 3 °C).  So I think there were a number of issues with the board relating to the grounding plane and some other parasitic effects that were giving the above excessive drift.

I looked at a few remedies for improving op amp grounding and isolation as well as routing of wires but decided that it would be faster to rebuild.  The 0.1" pitch eurocard boards from Busboard (POW3U) are not the best as they don't have a common ground across the board.  You have to choose between routing power or ground to the connected traces that go along the rows.  The surface mount busboard boards (SMT3UT) are much better as they have a copper plane covering the whole underside.

## Rebuild with surface mount design

Rebuild is shown below.  I used all soic-8 chips and thin film 0805 resistors.  These resistors should have a tolerance of 0.005% and a thermal slope that is better than 50ppm/K. I changed the OP27 for LT1012: these are ultralow drift that outperform OP07.

Schematic of a single channel is sketched below:

The sensor leads have a length of 5 m.  I measured the capacitance to be 300 pF, this seems high but the wires is probably higher gauge than it needs to be. I added 62 pF of feedback capacitance across all the TIA stages to compensate the lead capacitance. Stability and noise plots modeled in liso are shown below.

## Thermal stability numbers for LT1012

For the given above circuit with LT1012 we should expect a drift of 10 µK/K from op amp voltage drift, 0.3 µK/K from op amp current drift and 125 µK/K @ 25 for transresistance thermal drift for 50 ppm/K thin film resistors.

As before offset of the LT1021-7 should have given a drift of 0.55 mK/K for the voltage reference itself (2ppm/K) and 14 mK/K? for thermal drift of 25.64 kΩ series resistor (thin film 50 ppm/K worst case)

Thus total worst case drift of the circuit itself will be 14.7 mK/K, dominated by the offseting circuit. Resistors are the worst culprit in environmental drift of the pre-amplifier circuit.

Quote:

Great. A few things:

### Noise ASD

It looks like this spectral density is catching only below the 1/f corner frequency.  What you'll want to do for your actual measurement is to take a bunch of different frequency spans over the full range and stitch them together.  The iris plotting tools allow you to do this pretty easily

> python ~/Git/labutils/iris/iris.py --noTar NameOfBatchFiles*

The '*' is just to grab all the spans you took of that particular run. Then in the same directory something like

> python ~/Git/labutils/iris/iris.py --noTar --noStitch StitchedSpectrum_*

will make a nice plot of all the stitched spans.

### Drift and noise

I've attached a interactive notebook for the SK using LISO as backend to model the circuit.  You can see that the noise is dominated by the non-inverting pin current noise. You could choose a FET op amp.  However, I think the bigger concern is the thermally induced offset drift; for typical op amps this is in the range of 0.2 to 2 µV/K.  However given the signal is order of 7 V then maybe this is only a very small affect on the injected offsetting current: say 2 µV/K /25.6 kΩ = 78 pA/K or equivalently 0.078 mK of input referred drift per degree kelvin of drift in your chip. This would be a concern for precision thermal readout but in your case it is waaaay over spec'ed.  OP27 is probably good enough for now.

As a side note: the AD590 has a noise floor of 40 pA/rtHz, which is an equivalent input temperature noise floor of 1 uK/rtHz.  The readout circuit is not limited by the sensor but you are still much better than your requirements. You should quote the drift and noise in equivalent input units of what you are trying to measure, which is temperature.

You may well end up being dominated by drift in the TIA feedback resistor thermal sensitivity.  This will shift the gain of the circuit and therefore the readout slope of V/K.  Typical thick metal film resistors are about 50-200 ppm/K.  Thin film are on the order of 5-50 ppm/K.

### Schematics

Don't know what is going on the the second one there.  Did you mean to use an AD746? Also, the AD620 is an instrument amplifier, you probably don't want to use it in this application for a TIA.

Attachment 1: plot20180906_TemperatureSense_TIA_OP27Noise.pdf
Attachment 6: 20180907_TIAStage_ThermalSenseCircuit.pdf
Attachment 7: 18-09-07_13-06-35_3721_rot.jpg
2307   Wed Mar 20 15:57:55 2019 AnjaliNoise Budget2micronLasersNoise budget for the frequency discriminator

I was trying to prepare the noise budget for the frequency stabilisation setup for 2 micron laser. In this test, we use a fiber based Mach-Zehnder interferometer as a frequency discriminator to convert frequency noise to amplitude(voltage) fluctuations. The different noise sources to be considered in this analysis are the following

1. Shot noise
2. Dark current noise
3. Room thermal noise
4. Fiber thermal noise
5. Fiber acoustic noise
6. Laser intensity noise
7. Photothermal noise
• Attachment #1 shows the magnitude spectrum of the transfer fucntion for the frequency discriminator. I assumed the delay fiber length in one of the arms of the Mach-Zehnder interferometer is 10 m. This corresponds to a time delay of 50 ns. The corner frequency is inversely proportional to the time delay.
• The transfer fucntion can be interpreted as, if the frequency variation is very slow- it can't be distiguished by the discriminator and we dont get any signal.
• Attachment # 2 and # 3 shows the shot noise and fiber thermal noise respectively in rad/rtHz
• The shot noise is calculated  considering thorlabs DET10D detector with a responsivity of 1.2 A/W at 2 micron. The laser output power from Eblana laser is assumed to be 2 mW and the power reaching the detector is estimated to be 1.23 mW.
• The fiber thermal noise is calculated based on Wanser, K. H. (1992). Fundamental phase noise limit in optical fibres due to temperature fluctuations. Electronics letters28(1), 53-54.
• The dark current noise and room thermal noise are calculted to be a constant value (3.97x10-12 rad/rtHz and 1.20x10-12 rad/rtHz respectively)- but this is not correct
• The noise in rad/rtHz is converted to Hz/rtHz by mutiplying with the frequency (f) and it is shown in attachment #4
Attachment 1: High_pass_transfer_function.pdf
Attachment 2: Shot_noise.pdf
Attachment 3: Thermoconductive_noise_of_fiber_Glenn_parameter.pdf
Attachment 4: noise_budget_combined_log_log.png
2308   Wed Mar 20 19:17:50 2019 ranaNoise Budget2micronLasersNoise budget for the frequency discriminator

I think the dark noise should be very close to the shot noise and also have the same transfer function.

Also, the room temperature fluctuations are probably large at low frequencies. What PSD for room temperature did you use?

2309   Thu Mar 21 00:49:35 2019 AnjaliNoise Budget2micronLasersNoise budget for the frequency discriminator

I didn't use the transfer funtion in the calculation of dark current noise thinking dark current noise is a characteristic of the detector alone

Regarding room thermal fluctuation, I was plotting the thermal noise of the detector which is given in A/rtHz as

$=\sqrt{\frac{4k_bT}{R_f}}$

2310   Thu Mar 21 09:43:19 2019 AnjaliNoise Budget2micronLasersNoise budget for the frequency discriminator

Also, I have a doubt that whether the transfer function should be of low pass in nature. So, if the laser phase is fluctuating faster than the time it takes to propagate throught the delay fiber, we will not be able to discriminate .

If that is the case, the noise budget looks like as shown in attachment #1

Attachment 1: Noise_budget_with_low_pass_transfer_function.pdf
2311   Fri Mar 22 10:00:30 2019 AnjaliNoise Budget2micronLasersNoise budget for the frequency discriminator

Attachement #1 shows the modified noise budget with modification on dark current noise.

I still have to find out the PSD for room temperature to plot the room thermal noise

Attachment 1: Noise_budget_modified.pdf
2312   Mon Mar 25 09:40:16 2019 AnjaliNoise Budget2micronLasersNoise budget for the frequency discriminator
• Different fiber parameters are very important in determining the thermal noise of the fiber. I was considering SM1950 fiber for the calculations, but all the characteristic parameters for this fiber are not known.
• Glenn ( IEEE Journal of Quantum Electronics25(6), 1218-1224 ) and Jing Dong (Applied Physics Letters108(2), 021108) had given the set of fiber parameters. Attachment #1 shows the comparison of thermoconductive noise of the fiber with fiber parameters taken from Glenn and Jing Dong. The parameters from Jing Dong are for Corning standard SMF 28 fibers and they are the recent resuts compared to Glenn. Hence I think it would be good to follow the parameters given in Jing Dong's paper.
• It is also found that the thermal noise of fiber has contributions from the thermoconductive noise (https://ieeexplore.ieee.org/stamp/stamp.jsp?tp=&arnumber=255919) and thermomechanical noise (https://ieeexplore.ieee.org/stamp/stamp.jsp?tp=&arnumber=5614023) . Thermoconductive noise  (in rad/rtHz) predicts a frequency independent characteristics at lower frequencies where as thermomechanical noise (in rad/rtHz) has a 1/f characteristics over a certain frequency range. The thermomechanical noise is of concern only over a certain frequency range it is given by the ratio of velocity of sound with the total length of the fiber. This indicates that, in the case of thermomechanical noise, the fluctuation of concern is much slower than the time it takes for the sound wave to travel across the length of the fiber.
• Attachment #2 shows the noise budget including the thermomechanical noise of the fiber. With 10 m length of the fiber, the thermoconductive noise is of concern for frequencies lower than 364 Hz . Thus a low pass transfer function is assumed with corner frquency of 364 Hz in the calculation of thermomechanical noise.
Attachment 1: Comparison_of_thermoconductive_noise_in_Hz_per_rtHz.pdf
Attachment 2: Noise_budget_with_thermomechanical_noise.pdf
2313   Tue Mar 26 17:13:53 2019 AnjaliUpdate2micronLasersFrequency stabilization of 2 micron source

Experimental plan

• The aim is to repeat the homodyne measurement setup with  balanced detection scheme for the frequency stabilisation of 2 micron laser source.
• Attachment # 1 shows the schematic of the experimental setup.
• All the components specified in the schematic are available.
• For the balanced detection, we can initialy use SR560, but we need to design a new circuitry for the same.
Attachment 1: Homodyne_setup_for_2_micron.pdf
2323   Wed Apr 10 12:17:34 2019 AnjaliUpdate2micronLasersIntensity noise stabilization of 2 micron source

[Aidan, Anjali]

• Attachment #1 shows the cleaned up setup for  2 micron experiment
• We also have received the AOM . The initial test is to do the intensity stabilisation of 2 micron laser diode using AOM
• Today, we started with the laser and AOM characterisation
• Attachment #2 shows the schematic of the experimental setup for the laser characterisation. Laser output is connected to a Faray isolator and the isolator output is measured using Thorlabs thermal power meter.
• The current limit on the current controller is set to 95 mA (maximum allowed current to the laser diode is 120 mA) and thermistor is set to 10 kohm (maximum value of thermistor resistance is 10 kohm)
• Attachment #3 shows the output power as a function of input current to the laser diode.
• From the data sheet of the laser diode, the threshold current is 20 mA.For a given inut current,  the output power measured is less compared to the value given in the data sheet. This is beacuse of the insertion loss of the isolator in the path.
• The slope efficieny is calculted to be 0.018 mW/mA. The value given in the data sheet is 0.3 mW/mA. The reduction is because of the insertion loss of the isolator
• The output of isolator is then connected to the AOM. Attachement #4 shows the schematic of the setup. The zeroth order port of the AOM is connected to the power meter and the output power from the first order port is blocked.
• We then applied a dc voltage to the modulation input port of the AOM driver. We expect a decrease in the power from the zeroth order port with increase in the input voltage to the  AOM driver as the light is getting deflcted and power is geting coupled to the first order port.
• Attachmet #5 shows the variation in power from zeroth order port with input voltage to the AOM driver.As expected, the power levels are decreasing with increase in input voltage
• It is also observed that the output power levels are fluctuating. This could be because of temperature fluctuation and the thermal power meter is sensitive to that.
Attachment 1: 2_micron_setup.jpg
Attachment 2: Laser_chara_setup.png
Attachment 3: Laser_characteristics.pdf
Attachment 4: AOM_chara_setup.png
Attachment 5: AOM_characteristics.pdf
2325   Wed Apr 10 18:58:48 2019 ranaUpdate2micronLasersIntensity noise stabilization of 2 micron source

OK...but how will the amplitude stabilization be done? How about a diagram showing the feedback loop and electronics?

1. what is the feedback circuit?
2. Can it drive the low impedance of an AOM driver?
3. What is the photodiode?
4. It has to be an AC coupled feedback (DC coupled is always terrible).
5. What is the requirement on the intensity noise as a function of frequency?
 Quote: [Aidan, Anjali] Attachment #1 shows the cleaned up setup for  2 micron experiment
2328   Wed Apr 24 09:01:49 2019 AnjaliUpdate2micronLasersIntensity noise stabilization of 2 micron source

Attachment #1 shows the schematic of the experimental setup for amplitude stabilization using AOM. The proposed idea is as follows

• The output power from the  Zeroth order port of the AOM depends on the RF power supplied to the AOM and the  RF power from the RF driver is dependent on the input voltage to the driver.
• When amplitude of the laser chnages to due to amplitude noise, the output from photodiode will also changes. The output of one of the photodiodes is connected through a gain element (G) to the RF driver of AOM. So, when PD output changes , the gain of the gain element should be adjusted such that the input voltage to the AOM driver changes and thus the RF power which affect the output power from the AOM.
• The gain element could be SR560 (I still have to understand more about it) and the photodiodes that we have are thorlabs DET10D detector which are DC coupled. We also need additional TIA stage after the PD. There were two of them built by the summer student. We tested both and one of them only works fine now.
Quote:

OK...but how will the amplitude stabilization be done? How about a diagram showing the feedback loop and electronics?

1. what is the feedback circuit?
2. Can it drive the low impedance of an AOM driver?
3. What is the photodiode?
4. It has to be an AC coupled feedback (DC coupled is always terrible).
5. What is the requirement on the intensity noise as a function of frequency?
 Quote: [Aidan, Anjali] Attachment #1 shows the cleaned up setup for  2 micron experiment

Attachment 1: Intensity_stabilisation.png
2333   Sun May 5 15:17:33 2019 AnjaliNoise Budget2micronLasersFrequency noise measurement of 2 micron source

We are going to start the heterodyne configuration for the frequency noise measurement of 2 micron laser source. The schematic of the same is shown in attachment #1. The initial plan was to separate the photodetector output to in phase , quadrature component and then extract the frequency noise information from the same. The new plan suggested is to use a phase locked loop for the demodulation part. We can use IFR Marconi 2023A as a VCO and SR560 as the amplifier. The RF generator for the AOM and the VCO need to be locked through Rubidium frequency standard (FS725). One of the issues is that the band width of the photodetector that we have is 14 MHz and the AOM is at 80 MHz. The response of photodetector at 80 MHz is about 20 % of that at DC.I have modified the noise budget after discussing with Anchal. Attachment #2 shows the noise budget. The Marconi data is taken from Anchal. The data plotted in attachment #2 corresponds to Marconi #539 at 48 MHz and actuation slope of 500 Hz. We may have to find the actual numbers based on our settings for the experiment.  From the data sheet, the input noise of SR 560 is 4nV/rtHz. The frequency mixer (Minicircuit ZAD-1-1+) has a conversion loss of 4.83 dB and the thermal noise of it is not given in the data sheet and thus it is not included in the noise budget. The length of the fiber considered for MZI is 10 m.

Attachment 1: schmeatic_heterodyne.png
Attachment 2: Noise_budget_modified.pdf
2334   Sun May 5 20:52:16 2019 ranaNoise Budget2micronLasersFrequency noise measurement of 2 micron source

1. The LPF should be a 1.9 MHz LP from Mini-circuits
2. The SR560 should be set with a 1 MHz low pass only
3. No Rubidium needed. We just want to use two IFR/Marconi synthesizers so that they can share the same time standard.
4. We will need a RF Amp to drive the AOM hard enough - check the manual and see what level is required to get the max diffraction.
2338   Fri May 10 16:22:45 2019 AnjaliUpdate2micronLasersCharacterization of RF driver and AOM .

Following are the results from RF driver and AOM characterisation.

• Attachment #1 shows the results from  characterisation of Brimrose RF driver . The RF power and frequency are measured on Agilent spectrum analyser. A 30 dB attenuator was also used in the path from RF driver to spectrum analyser. This attenuation value is taken care in RF power output calculation. The RF driver has two BNC connectors labelled as “Modulation” and “Frequency” , located on the front panel. Varying the modulation input (in the range 0 V to 1 V) changes the RF output power from the RF driver as shown in attachment #1 (a). The maximum RF output power is about 0.6 W and the input RF power to the AOM is limited to this value as exceeding the same might cause damage to the AOM. Varying the frequency input (in the range 0 V to 10 V) changed the RF frequency from the RF driver as shown in attachment #1 (b). The AOM centre frequency is at 80 MHz with a frequency shift range of 8 MHz.

• Attachment #2 and #3 shows the output power from the zeroth order and first order port of the AOM when the frequency input voltage to the RF driver (thus the RF frequency from the driver) is varied. The output power from the first order port is maximum (output power from zeroth order is minimum) when the frequency is about 78.8 MHz.  As expected, the power in the zeroth order port is completely transferred to the first order port. This happens when the frequency input voltage to the RF driver is about 8.8 V.

• Attachment #4 and #5 shows the output power from the zeroth order port of the AOM when the Modulation input voltage to the RF driver (thus the RF power from the driver) is varied. The output power from the first order port is maximum (output power from zeroth order is minimum) when the RF power to the AOM is maximum. This happens when the modulation input voltage to the RF driver is 1 V.

• Attachment #5 shows the diffraction efficiency to the first order port (we use output power from first order port for the heterodyne measurement) as a function of frequency and RF power. The diffraction efficiency is calculated from the ratio of Power in first order port to the input power to the AOM.The power at the input end of AOM is 1.29 mW. So the percentage value calculated includes the insertion loss ( as per data sheet : 3-4 dB for the first order port ) as well . So the conclusion is , inorder to get maximum diffraction efficiency to the first order port of AOM, we should supply RF power of about 0.6 W at 78.8 MHz . If we are using the Brimrose driver, this can be set by giving modulation input voltage of 1 V and frequency input voltage of 8.8 V.

Attachment 1: RF_driver_characterisation.pdf
Attachment 2: AOM_zero_order_output_Vs_Frequency.pdf
Attachment 3: AOM_first_order_output_Vs_frequency.pdf
Attachment 4: AOM_zeroorder_output_Vs_RF_power.pdf
Attachment 5: AOM_first_order_output_Vs_RF_power.pdf
Attachment 6: Diffraction_efficiency.pdf
2343   Wed May 15 20:08:01 2019 AnjaliUpdate2micronLasersPLL loop for frequency noise measurement
• We are trying to setup the phase locked loop (PLL) for the frequency noise measurement of 2 micron laser. We started with a sample experiment in which we were trying to lock an arbitrary function generator (AFG) with the Marconi using PLL. Attachment #1 shows the schematic of the PLL setup. We are using a level 7 mixer (ZFM-3-S+). The RF port is connected to  AFG and LO port to Marconi. Output of mixer is passing through a low pass filter with cut off frequency at 1.9 MHz (SLP-1.9 +). The output of LPF is fed into input A of SR 560. SR560 is set with 1 MHz low pass. Initially, we set a gain value of 10 dB and actuation slope of 100 kHz/V in SR 560 and Marconi respectively. The 50 Ohm output  of SR 560 was connected to Marconi and the 600 Ohm output was connected to an oscilloscope to check the performance of PLL. The carrier frequency from AWG and Marconi were set close to each other (~ 11 MHz). We observed  a dc output at about 60 mV on the oscilloscope. This ensures that PLL is working.
• We then attempted to measure the band width. To do that, the source output from SR785 was fed into input B of SR560. Part of the source output was fed into channel 1 of SR 785, through T connector, for the transfer function measurement. We also used a T connector at the input A port of SR 560 and one of the ports of this T connector was fed into channel B of SR785. I still must interpret most of the results that we got.

• Attachment # 2: Closed loop transfer function (a) Magnitude (b) Phase, at different gain values  in SR 560 when the Marconi actuation slope is 10 kHz/V.

• Attachment # 3: Closed loop transfer function at different actuation slope value in Marconi when the gain is 7 dB.  The increase in noise at lower frequency in phase plot (b) may indicate that the phase/frequency noise of the Marconi increases if the actuation slope value is increased.

• Attachment # 4: Closed loop transfer function at different actuation slope value set in Marconi when the gain is 10 dB. The transfer function measured for the case of gain = 10 dB and actuation slope = 100 kHz/V (that is the product of gain and actuation slope is larger) shows significantly different characteristics.

• Using the SSUserFn option in SR785, we tried to get the open loop transfer function as well from SR 785. The functional form was $\frac{x}{1-x}$

• Attachment # 5: Open loop transfer function at different gain values set in SR 560 when the Marconi actuation slope is 10 kHz/V. The unity gain band width are 0.9 kHz, 2.8 kHz and 5.8 kHz respectively when the gain values are 3 dB, 7 dB and 10 dB

• Attachment # 6 : Closed loop transfer function at different actuation slope value set in Marconi when the gain is 7 dB. The unity gain band width are 3 kHz, 9 kHz and 30 kHz respectively when the actuation slope values are 10 kHz/V, 30 kHz/V, and 100 kHz/V.

• We also tried to estimate the open loop transfer function from the closed loop transfer function using the equation $G_{ol}=\frac{G_{cl}}{1-G_{cl}}$

• Attachment # 7 : Comparison of Open loop transfer function that is measured from SR 785 and that is estimated from the closed loop transfer function using the above expression. These two values are significantly different. Kindly correct me.

Attachment 1: PLL_setup.png
Attachment 2: closed_loop_FM_dvn_10kHz_diff_gain.pdf
Attachment 3: closed_loop_gain_5_diff_FM_dvn.pdf
Attachment 4: closed_loop_gain_10_diff_FM_dvn.pdf
Attachment 5: open_loop_FM_dvn_10kHz_diff_gain.pdf
Attachment 6: open_loop_gain_5_diff_FM_dvn.pdf
Attachment 7: Comparison_measured_estimated_open_loop.pdf
2344   Thu May 16 13:40:13 2019 anchalUpdate2micronLasersPLL loop for frequency noise measurement

I think you have made some coding error in your attachment 7 plot. Just pick a point in your plot and calculate by hand if your estimate is correct. Otherwise, we need to see your code to pinpoint the error. You can attach your code in a .zip file here.

2346   Sat May 18 22:25:30 2019 AnjaliUpdate2micronLasersCharacterization of new detector

The new photo detector has arrived (https://www.newport.com/p/818-BB-51F) . We did the DC and AC characterization of the same.

• Following table shows the results from DC characterisation
•  Laser diode current (mA) Input power (mW) Output voltage(mV) 50 0.5 7.2 70 0.9 13.6 90 1.3 19.7
• In this case, the input power was measured after the isolator.The power to voltage conversion is linear. The voltage levels are very low because this is a non-amplified detector. Also, the detector is coupled to a FC/UPC patch cord and we have all FC/APC fiber connectors. So, there could be some coupling loss from FC/APC to FC/UPC. FC/APC to FC/UPC conversion patch cord is ordered. We can check the performance again after it is arrived.

• We then assembled the Mach-Zehnder interferometer (MZI) for the 2-micron laser source. Attachment #1 shows the schematic of the same. We measured a power level of 0.37 mW when the AOM was not turned on (RF power to AOM off). When AOM is turned ON, the power level measured at the output of MZI is 0.5 mW. Power meter was then replaced with the new photodetector and the beat note was observed on spectrum analyser.

• Attachment # 2 shows the RF beat note at 78.8 MHz on spectrum analyser. So, the band width of the detector is enough to work at this frequency range. The signal to noise ratio is about 37 dB. There are some small peaks appearing at around 77 MHz and 80 MHz, but they are about  29 dB below the main peak.
• Attachment # 3 shows the noise floor. There are no other RF interferences at this frequency range.
Attachment 1: setup.png
Attachment 2: beat_note.pdf
Attachment 3: Noise_floor.pdf
2347   Sun May 19 22:08:09 2019 AnjaliUpdate2micronLasersPLL loop for frequency noise measurement
• After the discussion with Prof.Rana, we realised the mistake in our analysis. It was also suggested to make the measurement at the output of SR560. Attachment #1 shows the schematic of the setup for the measurement of closed loop transfer function. The RF power from AFG is 0 dBm and that  from Marconi is 7 dBm.

• The open loop transfer function is calculated from closed loop transfer function using the expression $G_{ol}=\frac{G_{cl}+G_{560}}{G_{cl}}$ , where $G_{560}}$ is the gain value set in SR560.

• Attachment #2 : Closed loop and open loop transfer functions at different values of gain in SR 560 when the actuation slope in Marconi is 10 kHz/V. The unity gain frequencies are respectively 1kHz, 3 kHz and 5 kHz when the gain values are 3 dB, 7 dB and 10 dB.

• Attachment #3 : Closed loop and open loop transfer function at different values of actuation slope in Marconi when the gain in SR 560 is 10 dB.The unity gain frequencies are respectively 5kHz, 16 kHz and 43 kHz when the actuation slope values are 10 kHz/V, 30 kHz/V and 100 kHz/V. It can be seen that the characteristics are significantly different for a larger value for the product of gain and actuation slope (G=10, S=100 kHz/V).

• The probing signal from SR 785 was then disconnected. In this case, the oscilloscope measure the error signal in time domain and the measurement from SR 785 essentially gives the frequency noise of AFG. The measurement from SR 785 has the unit of V/rt Hz, which is then multiplied with actuation slope to get the frequency noise in Hz/rt Hz. During our measurements, the oscilloscope signal was showing a low level ( mV) DC line , confirming that the PLL is  locked.

• Attachment # 4 : Frequency noise of AFG at different gain value in SR 560 when the actuation slope in Marconi is 30 kHz/V. In the full span mode, the line width (resolution) is 128 Hz where as the line width is 2 Hz in short span mode. The peak at 60 Hz visible in short span plot corresponds to AC mains.

• Attachment # 5 : Frequency noise of AFG at different  actuation slope in Marconi when the gain in SR 560 is 10 dB.  I was thinking, we should measure the same frequency noise irrespective of the setting in the PLL. It can be seen from attachment # 5 that the frequency noise measurement is affected by the value of actuation slope in Marconi. It was earlier observed that the phase noise of Marconi increases with increase in the actuation slope and , from these measurment shown in attachment #5, we are seeing increase in frequency noise value at larger values of actuation slope in Marconi.

 Quote: We also tried to estimate the open loop transfer function from the closed loop transfer function using the equation $G_{ol}=\frac{G_{cl}}{1-G_{cl}}$ Attachment # 7 : Comparison of Open loop transfer function that is measured from SR 785 and that is estimated from the closed loop transfer function using the above expression. These two values are significantly different. Kindly correct me.

Attachment 1: setup_close_loop_transfer_function.png
Attachment 2: Actuation_slope_10kHzV_diff_gain.pdf
Attachment 3: gain_10_different_actuation_slope.pdf
Attachment 4: FM_noise_AFG_Actuation_30kHzV_diff_gain.pdf
Attachment 5: FM_noise_AFG_gain_10_different_actuation_slope.pdf
2348   Mon May 20 02:23:14 2019 AnjaliUpdate2micronLasersFrequency noise measurement of 2 micron source using PLL
• Attachment # 1 shows the schematic of the experimental setup for the frequency noise measurement of 2-micron laser source using PLL. Instead of Brimrose driver, another Marconi is used to provide the RF power to the AOM. We know from the characterisation of AOM that we need to give RF power of 28 dBm at 78.8 MHz to achieve maximum diffraction efficiency to the first order port of AOM. The maximum output power from Marconi is 13 dBm. Hence, we used another RF amplifier (ZHL-3-A+) to amplify the RF power from Marconi. We initially tested the RF output from RF amplifier on spectrum analyser (RF power fed into spectrum analyser with proper attenuation in the path) and adjusted the RF frequency and power in Marconi such that we get 28 dBm output power from the RF amplifier at 78.8 MHz. The two marconis are set such that they are share the same time standard.

• Now, the output power from the photodetector in MZI (Laser diode operated at input current of 90 mA) is fed into the RF input port of the mixer, instead of AFG. The 600 Ohm output of SR 560 is observed on oscilloscope and SR 785 simultaneously.

• We observed dc line in the oscilloscope when the gain in SR 560 is set to 13 dB (20 times). Gain value below this ( 10 dB) or above this (17 dB) was showing oscillations in the oscilloscope with frequency varying with the actuation slope in Marconi. Attachment #2 shows the frequency noise measurement from SR 785 (V/rt Hz value from SR 785 multiplied with the actuation slope).

• It is observed that, the time domain trace on the oscilloscope was not very stable. In between, we could see the oscillation was popping up. Also, the trace on SR 785 was swinging a lot (attached the video). As we observed in the case of AFG, the FM noise measured increses with the value of actuation slope in Marconi.

• In this case, the RF power that is fed into the RF port of the mixer is very small (~ -40 dBm) compared to our previous experiment with AFG. So, I would like to repeat the sample experiment (locking AFG to Marconi) with AFG set to RF power comparable with that from the actual experiment.I should then find out the unity gain frequency of that particular combination of gain and actuation slope,which would help us to find the frequency range upto which the PLL measurment is valid.

•  I also need to measure the actual delay line length. I will also clean up the fiber connectors again and we can also use the FC/UPC to FC/APC patch cord for the detector after it is arrived. I still must understand the results better.Since we are using Non-PM fibers, the polarisation fluctuation might have also affected the measurement .Kindly give me further suggestions.

Attachment 1: FM_noise_measurement_setup.png
Attachment 2: 2_micron_FM_noise.pdf
Attachment 3: Oscilloscope.mp4
Attachment 4: SR_785.mp4
2350   Thu May 23 02:53:19 2019 AnjaliUpdate2micronLasersCharacterization of new detector

FC/UPC to FC/APC patch cord has arrived. I repeated the DC characterisation of the photodetector with this patchcord. The couping is improving by about 2 dB (Table below shws the result)

 Laser diode current (mA) Input power (mW) Output voltage (mv) 50 0.5 11 70 0.9 20.7 90 1.3 30.3
 Quote: In this case, the input power was measured after the isolator.The power to voltage conversion is linear. The voltage levels are very low because this is a non-amplified detector. Also, the detector is coupled to a FC/UPC patch cord and we have all FC/APC fiber connectors. So, there could be some coupling loss from FC/APC to FC/UPC. FC/APC to FC/UPC conversion patch cord is ordered. We can check the performance again after it is arrived.
2351   Thu May 23 03:00:24 2019 AnjaliUpdate2micronLasersFrequency noise measurement of 2 micron source using PLL
• The delay line length is measured to be 15.7 m.
• As suggested by Prof.Rana, I used a DC block after the photodetector and the output of it was then fed into a RF amplifier (ZHL-3 A+). Attachment #1 shows the output of the detector (a) MZI output- we see the beat note at 78.8 MHz and SNR is about 37 dB (b) when light is bloked- we see  radiation from RF amplifier with a small amplitude of about -63 dBm at 78.8 MHz  (c) when RF is turned off.
• It can be seen that, after the RF amplifier, the noise level is also increased with the signal.
• The RF level that is now fed into the mixer is about -23 dBm. But the PLL was not getting locked-oscilloscope trace is oscillating. A video is attached, which is captured when the there were no gain in SR 560 (G=1) and actuation slope was 10 kHz/V. Attachment 2 and 3 shows the corresponsing traces from SR 785 in full span mode and short spn mode respectively. It is observed that , the oscillation strength was increasing as I increased the gain value in SR 560. I doubt, this is happening because of the higher noise level at the RF port of the mixer in this case. The trace on SR 785 was more stable and it was not swinging as much as we observed before.
Attachment 1: Detector_output.pdf
Attachment 2: FM_noise_full_span.pdf
Attachment 3: FM_noise_short_span.pdf
Attachment 4: oscilloscope.mp4
2353   Sun May 26 01:12:33 2019 AnjaliUpdate2micronLasersFrequency noise measurement of 2 micron source using PLL
• I added one more amplifier stage (ZFL-500 LN) after the detector. Since noise figure of ZFL-500LN (2.9 dB) is lower than that of ZHL-3A (5 dB), ZFL-500LN is the first amplifier stage after the photo detector and it is followed by ZHL-3A.

• Attachment # 1 shows the beat note spectrum measured from the spectrum analyser. There was a 30 dB attenuator in the path during the measurement. So, the output RF power from the MZI (with two stages of amplification) is now about 3 dBm and the SNR of 37 dB is preserved even after two stages of amplification.

• So, now the RF power to the RF port of the mixer is 3 dBm. I have attached the video of signal from the PLL loop at different gain (G=1, G=2,G=5) values in SR 560. The time domain trace seems to very noise. I suspect this is because of the inherent large noise in 2-micron laser diode with a broad line width of 2 MHz.

• I then attempted to do the closed loop transfer function in the present PLL configuration by injecting the signal from SR 785. Attachment 2 shows the closed and open loop transfer functions at different gain values in SR 560 when the actuation slope is 10 kHz/V. Attachment 3 shows the closed and open loop transfer functions at different values of actuation slope when the gain is 5. The magnitude and phase traces are not very smooth as we observed when we did the similar measurement with an arbitrary function generator (AFG) as the RF source. In this case, when MZI output is fed in as the RF source, the RF power is fluctuating.

• I also tried to do the frequency noise measurement. Attchement # 4 is the FM noise at different gain values when the actuation slope is 10 kHz/V. Attachement 5 is the FM noise at different actuation slope values when the gain is 5. This time, depending on the gain value and the actuation slope value, a short frequency span was considered in SR 785 for the frequency noise measurement. The frequency span is considered based on the value of unity gain frequencies  that are approximated from the open loop transfer functions measured from attachment # 2 and #3

Attachment 1: beat_note.pdf
Attachment 2: Transfer_function_S_10kHzV_different_gain.pdf
Attachment 3: transfer_function_G_5_different_actuation.pdf
Attachment 4: FM_noise_diff_gain.pdf
Attachment 5: FM_noise_G_5_different_actuation_slope.pdf
Attachment 6: Gain_1.mp4
Attachment 7: Gain_2.mp4
Attachment 8: Gain_5.mp4
2355   Wed May 29 08:16:47 2019 AnjaliUpdate2micronLasersFrequency noise measurement of 2 micron source using PLL

Attachment #1 shows the oscilloscope traces at different gain values when the actuation slope is 100 kHz/V. It also shows the base line when there is no input to the oscilloscope. Even in the absence of any signal to the oscilloscope, there is an offset with mean value, RMS value and peak to peak value respectively of 35 mV, 42 mV and 200 mV.

Table below summarises the mean value, RMS value and peak to peak value for different combinations of actuation slope and gain.

 Actuation slope (kHz/V) Gain Mean value (mV) RMS value (mV) Peak-peak (V) 10 1 -22.5 44 0.3 10 2 -33 74 10 5 -70 205 1.52 30 1 2.7 33 0.26 30 2 -3.8 71 0.52 30 5 -85 189 1.08 100 1 42 67 0.42 100 2 -32 88 0.6 100 5 -65 207 1.36

The RMS value and the peak to peak value is increasing with increase in gain and the mean value is not showing any trend. I was pressing the Run/stop button before saving the data. I press the same to make the trace alive after saving the data as well. But the mean value read out from the oscilloscope shows different /random values in either case. If I don’t save the data, but only increases the gain, the mean value readout from oscilloscope shows almost the same.

I saw the beat note on the oscilloscope and I was trying to find the change in frequency. The frequency readout from oscilloscope was showing very large fluctuation (60-100 MHz). I feel its not a reliable measurement, but I don’t know whether we have an option to measure the frequency jitter in this oscilloscope (TDS 3032).

Attachment 1: oscilloscope_trace_s_100kHz.zip
2357   Thu May 30 07:43:21 2019 AnjaliUpdate2micronLasersFrequency noise measurement of 2 micron source using PLL
• The new 2 micron detector has a 50 ohm termination resistance, which is the transimpedance resistor, and hence the gain is very low 50 Ohms.
• We observed the output of photodetector (beat note from MZI) on oscilloscope. Initially, we observed a large DC value (20 mV) and a small AC value (peak-peak=5 mV). 20 mV of DC with 50-ohm termination corresponds to a photo current of 0.4 mA.  This corresponds to optical power of 0.42 mW (responsivity = 0.95 A/W). (I'm not sure if its really as good as 0.95 A/W at 2 microns)
• This ratio of  small AC value to large DC value says that the contrast is poor (25 %). We were then trying to improve the contrast. The RF power and RF frequency to the AOM was already set to have the maximum contrast. Prof. Rana then removed the fiber connection from laser to isolator and redid the connection. Surprisingly, the contrast became very good. We measured a AC peak-peak of 23 mV, which indicates almost 100% contrast.
• We then added a DC block and an amplifier stage (ZFL-500 LN) after the photodetector. Looking at the RMS value of the sinusoidal signal on oscilloscope, we realised that the RMS value is increasing as the thermistor resistance value set on the temperature controllerfor the laser diode increases. From the oscilloscope readout, we measured RMS value of 80 mV, 89 mV and 105 mV respectively when the thermistor resistance values are 10 kΩ, 11 kΩ and 13 kΩ. But as per the data sheet of the laser, the typical value of thermistor resistance is 10 k kΩ and the maximum value is 10.5 kΩ. Also, as per the data sheet, the thermistor temperature coefficient is -4.4 %/oC. I suppose, from the negative value, the temperature of the laser diode is reducing as the thermistor resistance value increases. Kindly correct me. This laser diode also has a current tuning coefficient of 0.01 nm/mA and temperature tuning coefficient of 0. 1 nm/ oC.
• Attached the videos of oscilloscope signal when the thermistor resistance is 10 kΩ. From the oscilloscope trace, there is no significant amplitude fluctuation, but the frequency seems to be fluctuating a lot.
• I have attached another video when the beat note observed on spectrum analyser (this video was captured when we had a dc block and two amplifier stage after the photo detector), but the characteristics are the same, except the power evel is different now. The peak is fluctuating a lot (sometime, the peak is even disappearing). We observed these fluctuations even with a resolution band width of 1 MHz. This indicates that the frequency of the laser diode is fluctuating by a large factor which is even greater than the maximum actuation Marconi can provide. Hence, we will not be able to lock the PLL with this laser having large frequency fluctuation. We need to find another method to measure the frequency noise of the laser. One option is to perform RF herterodyne, but we don’t have a deep memory oscilloscope or ADC with large sampling rate to capture the data. Kindly give further suggestions to perform the frequency noise measurement.

RXA: we have a few options for measuring large frequency fluctuations:

1. make an electronic delay line frequency discriminator. This is what is used at the 40m lab to track the ALS beat note. This requires the use of a medium power RF amp (ZHL-3A or similar), a splitter, 1 short cable, 1 long cable, and a mixer/LP. Two of these setups to get I & Q.
2. a VCO with a much larger range than the Marconi - maybe 10 MHz p-p would be enough
3. Use the heterodyne setup that Anjali mentions: a 90 deg hybrid splitter to get I & Q and then record the two mixer outputs with the Moku
Attachment 1: Oscilloscope.mp4
Attachment 2: Spectrum_analyser.mp4
2358   Sat Jun 1 09:46:00 2019 AnjaliUpdate2micronLasersFrequency noise measurement of 2 micron source

[Anchal, Anjali]

We tried the Lock in amplifier in Moku lab for the frequency noise measurement. In this case, the output of the photo detector , after dc block and one stage of amplification, is fed into  input 1 of Moku. It gives out the inphase and quadrature component. We have saved the data. I will process the data offline and update later.

2359   Wed Jun 5 14:16:07 2019 KojiUpdate2micronLasers2um laser / cryo stat: setup inspection and action item updates

[Aidan, Chris, Koji]

We went down to the lab to check the situation of the setups for 2um laser measurement and stabilization and the new cryostat.

[2um laser frequency noise measurement]

• Looked at the add-on transimpedance amps: Something was wrong with them. The power bypass caps are attached to the "hot" supply lines in parallel (both sides of the caps are soldered to a same line). And some power supply lines have no voltage. This circuit is not necessary to be bipolar. To be fixed (KA)
• We temporarily connected one of the thorlabs 2um InGaAs biased detectors to an SR560. It showed reasonable output: DC/AC response OK & no nonsense.
• The AOM was bypassed and the homodyne fringe was checked.
The fringe visibility was low (~10%) and was dependent on the stress applied to the delayline fiber.
Suspected polarization rotation somewhere -> ToDo: Check the polarization states of the output beams.
• ToDo: Check should be done with each component. how much are the output power, output polarization, dependence/fluctuation of the polarization, etc.
We might be able to use the 2um Faraday Isolators (as PBSs) for the measuement.
• Checked the fringing of the fiber delayline Mach Zehnder. We observed one fringe per sec level fluctuation.
• Laser current actuation was checked and it turned out that it is so strong and sufficient to lock the delayline fringe.

[2um AOM]

• The fiber coupled AOM gave us a reasonable amount of DC/AC actuation of the laser intensity.
• The power of the 1st order output has the dependence on the "freq input" of the driver. This is probably because of the matching between the fiber coupling and the deflection angle, which is freq dependent.
• When the freq input is 8.8V_DC, the 1st order output has the maximum efficiency. The efficiency was 96%@990mV_DC input to the modulation in.
• The AOM actuation bandwidth was tested to be ~MHz, at least.
• We are not supposed to give more than 1V to the modulation in while we want to apply 8.8V to the freq input. Incorrect plugging may cause the damage of the modulation input port. The setup needs to be improved with a protection circuits / AOM driver circuit.
• Our understanding is that the modulation input has a 50Ohm input impedance while the freq input has high-Z
• The next step towards the intensity stabilization is low noise photodetector circuits and proper interface to the AOM driver.
• Also we want to set up TECs and other circuits for the LaserComponents PDs.

[Cryostat]

• Cleaning: there are many components are scattered on the table.
• Plan:
• Move the Zack rack to the next of the optical table (or somewhere)
• Move the yellow chemical cabinet to the place where the rack was. We can pile up some plastic boxes on it.
• Remove the delicate optics from the steel table.
• Place heavy cryostat components on the steel table.
• Connect the cryo cooler to the cryostat. How do we do that? Fisrt rigidly attach for testing and then move to soft attaching?
• Replace the optical windows to the 2um ones (2"). The current ones are for 1.5um.
• We need a 2" 50/50 BS at 2um. Lenses and steering mirrors are in hand.
2360   Fri Jun 7 16:32:59 2019 AnjaliUpdate2micronLasersFrequency noise measurement of 2 micron source

Attachment # 1 show the schematic of the lock in amplifier configuration used in Moku lab. We saved the in phase and quadrature components.

In phase =$cos\left ( \right \omega_o\tau+\Delta\phi(t))$

quadrature =$Sin\left ( \right \omega_o\tau+\Delta\phi(t))$

where $\omega_0$  corresponds to 2 -micron , $\tau$ is the delay time in the delay fiber  and $\Delta\phi(t)=\phi(t) - \phi(t-\tau)$.

$\phi(t)$ is the phase noise of the laser.

From the inphase and quadrature component $\Delta \phi(t)$value is extracted. So, we are actually extrating the combined effect of phase noise of laser as well the phase noise due to fiber length fluctuations due to environmental fluctuations. ASD of this is converted to frequency noise in Hz/rt Hz. Attachment # 2 shows the frequency noise estimated from two sets of measurements. This curve exhibit a 1/f characteristics from about 240 Hz upto 30 kHz

 Quote: [Anchal, Anjali] We tried the Lock in amplifier in Moku lab for the frequency noise measurement. In this case, the output of the photo detector , after dc block and one stage of amplification, is fed into  input 1 of Moku. It gives out the inphase and quadrature component. We have saved the data. I will process the data offline and update later.

Attachment 1: Lock_in_amplifier_moku_lab.jpg
Attachment 2: FM_noise.pdf
2362   Thu Jun 13 22:53:25 2019 AnjaliUpdate2micronLasersFrequency noise measurement of 2 micron source

There was a correction in the script I used to estimate the frequency noise from the inphase and quadrature component. Attachment #1 shows the frequency noise estimated after the correction.

I have also attached the Matlab script ( I am not able to attach the zip file with data files). I remember, while saving the data, we gave the time duration as 70 s. But while processing the data only I realised that the  time domain data is captured only upto 2 s. Even in this case, I would expect the frequency axis to start from 0.5 Hz, but I don't see that in the FM noise plot. Kinldy let me know whther I am doing anything wrong in data proocessing.

Quote:

Attachment # 1 show the schematic of the lock in amplifier configuration used in Moku lab. We saved the in phase and quadrature components.

In phase =$cos\left ( \right \omega_o\tau+\Delta\phi(t))$

quadrature =$Sin\left ( \right \omega_o\tau+\Delta\phi(t))$

where $\omega_0$  corresponds to 2 -micron , $\tau$ is the delay time in the delay fiber  and $\Delta\phi(t)=\phi(t) - \phi(t-\tau)$.

$\phi(t)$ is the phase noise of the laser.

From the inphase and quadrature component $\Delta \phi(t)$value is extracted. So, we are actually extrating the combined effect of phase noise of laser as well the phase noise due to fiber length fluctuations due to environmental fluctuations. ASD of this is converted to frequency noise in Hz/rt Hz. Attachment # 2 shows the frequency noise estimated from two sets of measurements. This curve exhibit a 1/f characteristics from about 240 Hz upto 30 kHz

 Quote: [Anchal, Anjali] We tried the Lock in amplifier in Moku lab for the frequency noise measurement. In this case, the output of the photo detector , after dc block and one stage of amplification, is fed into  input 1 of Moku. It gives out the inphase and quadrature component. We have saved the data. I will process the data offline and update later.

Attachment 1: FM_noise_new_.pdf
Attachment 2: Mokulab.m
close all
clear all
clc
lam=2e-6;%wavelength
c=3e8;%velocity of light
n=1.5;%refractive index of fiber
len=15;%length of delay fiber
omeg=2*pi*(c/lam);%optical frequency corresponds to 2-micron
tau=(len*n)/c;%time delay due to delay fier
Fs=500e3;%acquisition rate

... 27 more lines ...
2366   Wed Jun 26 11:45:09 2019 KojiSummary2micronLasers2um PD transimpedance

The transimpedance amps for the 2um (unamped) InGaAs detectors were made and evaluated.

Attachment 1: The circuit diagram

The usual transimpedance configuration. The detector (Thorlabs DET-10D) is an extended InGaAs which is sensitive up to 2.2um. I believe the detector is biased to 1.8V although it is not obvious and the 12V battery is used. The feedback resistor was chosen to be 5kOhm so that the circuit can handle up to ~2mA (~1.7mW). The feedback capacitance pf 100pF for compensation was chosen kind of arbitrary to keep the circuit stable and also the RC cut off to be more than 100kHz. The output resistance is 100Ohm. The selection of the opamp is described below.

Attachment 2: The amplifier noise Part I

The amplifier noise (the first unit called Amp #2)  was evaluated with the opamp swapped with OP27 (BJT), LT1128 (BJT), OPA604 (FET), and LT1792 (FET), chosen from the 40m stock. For the given environment, the FET amps exhibited better performance while the BJT amps suffered from more line noise coupling and the larger 1/f noise. Particularly, LT1792 reached at the level of ~2pA/rtHz, with lower line noises. This looks the best among them. Note that the 5kOhm feedback resistor gives 1.8pA/rtHz current noise.

Attachment 3: The amplifier noise Part II

Then the second unit (called Amp #1) was made. This unit has more high-frequency noise. It turned out that the noise was coming from the power supply which was the +/-12V from the rear panel of an SR560 which was connected to the AC power. The noise dramatically went away with the battery mode operation of SR560 (by disconnecting the AC power). The floor level was 2.2pA/rtHz and it was slightly higher than the quadratic sum of Johnson noise of 5kOhm and the voltage noise of the amp (4nV/rtHz). This noise level was just sufficient for the purpose of the 2um detector.

Attachment 4: The detector noise levels

Now the detector #1 and #2 were paired with the amp #1 and #2, respectively. In fact the detector 1/f noise was way too large compared to the amplifier noise. There is no hope to detect shot noise level of the mA photocurrent.

Attachment 5: The detector response

The detector response of each PD+AMP pair was measured using Jenne's laser and Thorlabs PD10A (~150MHz). There was some systematic error of the absolute level calibration, therefore the transfer functions were adjusted so that they have 5kOhm transimpedance at ~1kHz. The phase delay is ~30deg at 100kHz. This partially comes from the combination of 100pF//5kOhm and the ~4MHz bandwidth gain of the opamp. If we want faster response we need to modify these.

Attachment 1: PD_model.pdf
Attachment 2: AMP2_Noise.pdf
Attachment 3: AMP_Noises.pdf
Attachment 4: PD_noise.pdf
Attachment 5: PD_TF.pdf
2368   Mon Jul 1 21:20:55 2019 KojiSummary2micronLasersThe PDs delivered to the lab

The amplifier sets for the thorlabs 2um PDs were delivered to the lab.

- PD1 and Amp1, PD2 and Amp2 are the proper combination. If a high quality power supply is used, it is not an issue.
- The cables for the external bench supply or the 9V batteries have been made.

2372   Mon Jul 22 20:21:33 2019 ShalikaNoise Budget2micronLasersNoise Analysis of Circuit using SR785 Spectrum Analyzer and Zero Simulation

The transimpedance amps, differential circuit and the whitening filter for the 2um Extended InGaAs detectors were made and their noise level was evaluated. The examination of noise was done without the diodes. For every analysis, SR785 spectrum analyzer was used and a simulation using zero in python was also done. The SR785 was controlled using the python program to get the data. The input was AC coupled and Hanning window function was used during the task of getting data.

**The noise of SR785 spectrum analyzer is also mentioned, which was measured by deploying a terminator at one channel. The noise labelled as SR785noise mentions the spectrum analyzer noise alone.

**The noise of the components, with SR785 mentioned along with them, indicates the noise observed using the Spectrum analyzer.

Attachment 1: The circuit diagram

The usual transimpedance configuration was made using the OP27 IC. There are two TIA as we will be using two PD in our final circuit. The two TIA are connected to Whitening Filter respectively. The Whitening filter has a gain of 10. Apart from that, the output of the two TIA is connected to a differential circuit and whose output is in turn connected to another independent whitening filter.

Attachment 2: The amplifier noise Part I

The noise of TIA1 was analyzed using the SR785 spectrum analyzer and also by simulating the circuit using zero in python.

Attachment 3: The amplifier noise Part II

The noise of TIA2 was analyzed using the SR785 spectrum analyzer and also by simulating the circuit using zero in python. It was observed to be exactly similar to that observed of TIA1.

Attachment 4: The differential circuit noise levels

The noise level of the differential circuit was measured by shorting both the input terminals and observing the output at the pin6 of IC.

Attachment 5: The Whitening Filter noise level.

The noise of the Whitening filter was measured by grounding the input terminal (pin 3) at taking the measurement at the output. The gain of the filter is 10.

Attachment 6: The TIA and Whitening Filter, connected together, noise level.

The TIA and Whitening filter were connected in series and the noise was observed at the output of the whitening filter.

Attachment 7: The Simulation of Circuit in Zero

The complete circuit was configured using zero. It will help us analyze the noise that we are not expecting. Please open it using Jupyter Notebook.

Attachment 1: full_circuit.pdf
Attachment 2: Noise_at_node_naout.pdf
Attachment 3: Noise_at_node_nbout.pdf
Attachment 4: Noise_at_node_ndout.pdf
Attachment 5: Noise_at_node_nout.pdf
Attachment 6: Noise_at_node_nw1out.pdf
Attachment 7: noise_simulation.zip
2373   Mon Jul 22 20:36:35 2019 KojiElectronics2micronLasersSockets for LaserComponents PDs

We received the TO-66 sockets for LaserComponents PDs (Andon Electronics F425-1009-01-295V-R27-L14 Qty.10). It is made of FRP. It is very nicely made.

Attachment 1: IMG_8793.jpg
Attachment 2: IMG_8792.jpg
Attachment 3: IMG_8791.jpg
2374   Tue Jul 23 21:02:14 2019 Shalika SinghDailyProgress2micronLasersTransfer Function using SR785 spectrum analyzer

The transfer function of the circuit was analyzed.

Attachment 1: The circuit diagramThe usual transimpedance configuration was made using the OP27 IC. There are two TIA as we will be using two PD in our final circuit. The two TIA are connected to Whitening Filter respectively. The Whitening filter has a gain of 10. Apart from that, the output of the two TIA is connected to a differential circuit and whose output is in turn connected to another independent whitening filter.

Attachment 2: The Transfer Function

The transfer function of the transimpedance amplifier, Differential Circuit(at both its input nodes), Whitening Filter was analyzed by using the SR785 spectrum analyzer. A 1V source was applied at the input of the respective circuit under consideration.

**A 1k resistor was kept in series with every node which was given a source of 1V.

The terminology used in the graph:

TIA- Transfer function of the transimpedance amplifier

Differential at node1- TF of Differential circuit by providing source at node1

Differential at node2- TF of Differential circuit by providing source at node2

Whitening Filter- TF of Whitening filter by providing source at node1

TIA and Whitening filter- TF of transimpedance amplifier in series with the Whitening filter, and the source were at node1 of the TIA.

Attachment 1: full_circuit.pdf
Attachment 2: Transfer_Function_of_Circuit.pdf
2375   Wed Jul 24 18:50:27 2019 Shalika SinghDailyProgress2micronLasersCorrection in Transfer Function using SR785 spectrum analyzer

There were some errors in the previously obtained transfer function of the part where TIA was in series with the Whitening Filter. So, below is the updated transfer function plot.

Attachment 1: The circuit diagram

The usual transimpedance configuration was made using the OP27 IC. There are two TIA as we will be using two PD in our final circuit. The two TIA are connected to Whitening Filter respectively. The Whitening filter has a gain of 10. Apart from that, the output of the two TIA is connected to a differential circuit and whose output is in turn connected to another independent whitening filter.

Attachment 2: The Transfer Function

The transfer function of the transimpedance amplifier, Differential Circuit(at both its input nodes), Whitening Filter was analyzed by using the SR785 spectrum analyzer. A 1V source was applied at the input of a 10k resistor which was in turn connected in series with the respective circuit under consideration. The averaging mode is RMS and the input ground node is floating. The input is AC coupled.

The terminology used in the graph:

TIA- Transfer function of the transimpedance amplifier

Differential at node1- TF of Differential circuit by providing source at node1

Differential at node2- TF of Differential circuit by providing source at node2

Whitening Filter- TF of Whitening filter by providing source at node1

TIA and Whitening filter- TF of transimpedance amplifier in series with the Whitening filter, and the source were at node1 of the TIA.

Attachment 1: full_circuit.pdf
Attachment 2: Transfer_Fucntion.pdf
2376   Thu Jul 25 11:15:40 2019 Nathan HollandDailyProgress2micronLasersPhase Noise of Moku.

Attached are the results from measuring the phase noise of the Moku with a SRS DS345, which was the available signal generator in the QIL lab. Attachment 1 is the estimate of the Moku phase noise, with attachement 2 containing the data. Attachment 3 shows the data this was obtained from, and attachment 4 the experimental setup used to measure this. 15.625 kHz is the maximum frequency that the Moku will let you acquire two channels of data, in the phasemeter setup. In this setup I generated a 20 MHz signal on the DS345. This was then split, via a T piece, and sent through cables of length 0.6414 m to inputs 1 and 2 of the Moku. Concurrently the 10 MHz synchronisation signal was sent, also through a 0.6414 m cable, with an additional elbow piece, from the DS345 output to the Moku 10 MHz input. The phase difference between input 1 and input 2, shown in yellow orange in attachment 3, should be sqrt(2) times the phase noise of the Moku.

Attachment 5 shows the same setup, but with the spectrum straight from the Moku, which can be acquired at a maximum speed of 488 Hz. They are relatively consistent in the overlapping frequencies. Attachment 6 shows what happens when the 10 MHz synchronisation is removed. The low frequency performance of individual channels suffers but there is not a substantial change in the difference, yellow orange.

Attachment 8 shows the phase difference performance at 10 MHz, with attachment 7 showing the setup for this measurment. The cable length for this measurment was 0.9993 m for all paths. Again the difference, yellow orange, is comparable to other measurements.

Attachment 1: 20190725__Moku_phase_noise_SRS-DS345.pdf
Attachment 2: Moku_Phase_Noise_Estimate__20190724.csv
Attachment 3: 20190725__Moku_phase_diff_fast_1.pdf
Attachment 4: 20190725__Moku_phase_noise_setup.pdf
Attachment 5: Moku_phasediff_SRSDS345_20MHz_Sync_Screenshot.png
Attachment 6: Moku_phasediff_SRSDS345_20MHz_NoSync_Screenshot.png
Attachment 7: 20190725__Moku_10MHz_phase_diff_setup.pdf
Attachment 8: Moku_phasediff_SRSDS345_10MHz_SyncSig_Screenshot.png
2377   Mon Jul 29 14:46:09 2019 Nathan HollandDailyProgress2micronLasersOptical Phase noise of 2 um Mach Zehnder Interferometer.

Figure 1, attached, shows the phase noise I measure from the 2 um Mach Zehnder interferometer. This phase noise is from the optical path and is affected by:

• Intensity noise.
• Laser frequency noise.
• Differential polarisation shifts through the IFO arms.
• Changes in differential arm length.
• Changes in refractive index.
• Detector noise, in which I am also grouping noise due to the amplifier.

The next step is to determine contributions from each of these sources.

Attachment 2 shows the setup used to measure this noise. Attachment 3 shows the measured phase spectra from both phasemeters of the Moku. Attachment 4 contains the both time series and spectra. The spectra shown here are:

1. Converted to radians, from cycles.
2. Subtracted, in time domain.
3. Converted into power spectra, using Welch's method, 30 averages and Hanning windows.
4. Converted into amplitude spectra, by taking the square root.
Attachment 1: Optical_Phase_Noise_Spectra_2um_Mach_Zehnder__20190729.pdf
Attachment 2: 20190729__Initial_Phase_Noise_Measurement_setup.pdf
Attachment 3: Measured_phase_spectra_2um_Mach_Zehnder__20190729.pdf
Attachment 4: 2um_Mach_Zehnder_data__20190729.hdf5
2378   Mon Jul 29 15:46:50 2019 KojiLaser2micronLasersOptical Phase noise of 2 um Mach Zehnder Interferometer.

Great! Can you convert this into the laser frequency noise Hz/rtHz? I believe this [rad/rtHz] was still the measured phase noise and was neither the laser phase noise nor frequency noise yet.

2379   Mon Jul 29 17:52:33 2019 Shalika SinghDailyProgress2micronLasersNoise Analysis of Circuit using SR785 Spectrum Analyser and Zero Simulation

Attachment 1: The Circuit Diagram,

>>  the TIA with a gain of 5.1k

>>  Differential Circuit with a gain of 100.

Attachment 2: Noise across TIA

The input-referred current noise across the trans-impedance amplifier was measured using SR785 and was compared against the incoherent sum of input-referred current noise graph obtained from ZERO simulation.

Attachment 3: Noise across Differential circuit

The input-referred current noise across the Differential circuit was measured using SR785 and was compared against the graph obtained from ZERO simulation.

During measurement of transfer function using SR785, a source of 1V was given to a 10k resistor which was connected in series with the circuit taken into consideration. The channels of SR785 were set to AC coupling and input was set to Ground. Apart from that Hanning window function was used for measurements from SR785.

Attachment 1: full_circuit.pdf
Attachment 2: Noise_across_TIA.pdf
Attachment 3: Noise_across_Differential_Circuit.pdf
2380   Tue Jul 30 20:42:06 2019 ranaDailyProgress2micronLasersNoise Analysis of Circuit using SR785 Spectrum Analyser and Zero Simulation

Oh no! We've lost all of the low frequency data (which is the whole point of this experiment) and all of the contributing noise sources to the circuit!

2381   Thu Aug 1 11:25:27 2019 Shalika SinghDailyProgress2micronLasersNoise Analysis of Circuit using SR785 Spectrum Analyser and Zero Simulation

Input Referred noise to be calculated for trans-impedance.

Attachment 1: The Setup,

The setup was made on a breadboard, as per the circuit diagram.

Attachment 2: The Circuit Diagram on paper,

>>  the TIA with a gain of 5.1k

Attachment 3: Noise across TIA on SR785 screen

This is the noise as seen on the screen of the SR785

Attachment 4: Noise across TIA captured using the SR785 template file

The input-referred current noise across the TIA was measured using SR785 and was compared against the graph obtained from ZERO simulation.

** There is a difficulty in capturing the exact form of data as displayed on the SR785. Previously, I had captured the full span by keeping the start frequency as 1Hz. This gave only two points between 1Hz and 100Hz, all the rest of 798 points out of 800 were plotted at a higher frequency. This gave a straight line of higher magnitudes at low frequency(1Hz to 100Hz) since there were only 2 points. But the plot on the SR785 screen looks different.

** This time, I divided the measurements into 3 parts, 1Hz to 100 Hz, 100Hz to 10kHz and 10kHz to 100kHz. This left me with a graph which looks clumsier than the previous ones. I guess if there is a way that the template files of SR785 can be modified then it can give a graph which will align properly to the simulated results. For the time being, I can try again to take measurements in 3 sections but this time with fewer(300 instead of 800) points.

** It's difficult to avoid 60Hz harmonics with a circuit kept in open as this one. Lots of its effects are visible in the plot.

Attachment 1: Circuit.pdf
Attachment 2: full_circuit.pdf
Attachment 3: SR785_display_screen.pdf
Attachment 4: Noise_across_TIA.pdf
2382   Thu Aug 1 13:01:15 2019 ranaDailyProgress2micronLasersNoise Analysis of Circuit using SR785 Spectrum Analyser and Zero Simulation

always keep the start frequency at 0 Hz no matter what the span, as I was showing you yesterday in the lab. Otherwise, the bin centers end up in weird places.

2383   Fri Aug 2 16:15:01 2019 Nathan HollandDailyProgress2micronLasersFrequency Noise Measured by 2um Mach Zehnder.

Here I follow, essentially, the same setup as in attachment 2 of elog 2377. The difference is that I measured the LO path and IFO paths sequentially, not concurrently. This allowed me to measure at 125 kilosamples per second. Note that there is an error in the pymoku conversion tool when recording on channel 2 only. To circumvent this record both paths with channel 1.

Shown in attachment 1 is the frequency noise of the 2um Mach Zehnder interferometer. Attachment 2 shows the measured phase noises of the individual paths. Attachment 3 contains the plotted frequency spectra, and phase to frequency conversion factor. The conversion factor from phase, in radians, to frequency is:

$\frac{c}{2 \, \pi \, n \, { \Delta L }_{\text{IFO}} }$

The table below gives the values necessary to calculate this. It is 2.73 x 106 Hz rad-1.

 Parameter Value Notes LO arm length 2.103 m Delay line arm length 14.6 m Approximation. Awaiting exact value from Aidan. Refractive Index 1.4 Assuming SiO2. From refractiveindex.info Awaiting manufacturer response for precise value.

The smoothed spectra below were generated by this code. The full time series data is too large to upload here, at 272 MB when doubly compressed. Low frequency data is difficult to obtain due to the length of time that there is sufficient SNR for the phasemeter to lock onto the 80 MHz signal. Locking the Mach Zehnder could resolve this problem.

Attachment 1: Measured_Mach_Zehnder_frequency_noise_SmoothMedian__20190805.pdf
Attachment 2: Measured_phase_spectra_2um_Mach_Zehnder_BB_SmoothMedian__20190805.pdf
Attachment 3: 2um_MachZehnder_frequency_noise__20190805.hdf5
2384   Thu Aug 8 17:19:12 2019 Shalika SinghNoise Budget2micronLasersNoise Analysis of TIA using SR785 Spectrum Analyser and Zero Simulation

Input Referred noise to be calculated for trans-impedance.

Attachment 1: The Circuit Diagram on paper,

>>  the TIA with a gain of 5.1k

Attachment 2: Noise across TIA

The input-referred current noise across the TIA was measured using SR785 and was compared against the graph obtained from ZERO simulation.

** This time, I divided the measurements into 7 parts, 0-800, 800-2.4k, 2.4k-5.6k, 5.6k-12k, 12k-24.8k, 24.8k-50.4k, 50.4k-101.6k. The number of points for each was 800. Hanning Window function was used in the template file and the Input channels were grounded.

Attachment 3: Noise across TIA

The input-referred current noise across the TIA was measured using SR785 and was compared against the graph obtained from ZERO simulation.

** This time, I divided the measurements into 1 part, 10-6.4k. The number of points for each was 800. Hanning Window function was used in the template file and the Input channels were grounded.

** To measure the noise the output was measured at the pin6 of the OpAmp.

** A source of 1V was applied to the circuit by keeping a 10k in series with the input of the circuit when Transfer Function was being measured.

** It's difficult to avoid 60Hz harmonics with a circuit kept in open as this one. Lots of its effects are visible in the plot.

Attachment 4: Noise of the power supply used

The power supply was observed to be used to be noisy.

Attachment 1: TIA.pdf
Attachment 2: Noise_across_TIA.pdf
Attachment 3: Noise_across_TIA.pdf
Attachment 4: powersupply.pdf
2385   Fri Aug 9 21:10:48 2019 Shalika SinghNoise Budget2micronLasersNoise Analysis of Circuit using SR785 Spectrum Analyser and Zero Simulation

Input Referred noise is calculated for the circuit that is to be used for characterization of photodiodes. For the biasing 12V was used from SR560 as it provides cleaner voltage as compared to other voltage supplies.

Attachment 1: The Circuit Diagram TIA

>>  the TIA with a gain of 5.1k

Attachment 2: Input Referred noise of TIA

The input-referred current noise across the TIA was measured using SR785 and was compared against the graph obtained from ZERO simulation.

Attachment 3: Differential Circuit

>> gain of 100

Attachment 4: Input Referred noise of Differential Circuit

The input-referred voltage noise measured using SR785 and was compared against the graph obtained from ZERO simulation.

Attachment 5: Whitening Filter Circuit

>> gain of 10

Attachment 6: Input Referred noise of Whitening Filter Circuit

The input-referred voltage noise was measured using SR785 and was compared against the graph obtained from ZERO simulation.

Some points that were observed:

*** I am observing deviation from simulated results at higher frequencies. Presently, I am unable to understand the cause of this deviation.

*** At low frequencies deviation from simulated results is perhaps caused due to 60Hz harmonics and 1/f noise.

Attachment 1: TIA.png
Attachment 2: Noise_across_TIA.pdf
Attachment 3: Differential_Circuit.png
Attachment 4: Noise_across_Differential_Circuit.pdf
Attachment 5: Whitening_Filter.png
Attachment 6: Noise_across_Whitening_Filter.pdf
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