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ID Date Author Type Categoryup Subject
  2048   Mon Mar 7 15:34:46 2016 KojiNMiscPD QEBack reflection measurement with the chopper


We are still suffering from excess amount of back reflection compared with the predicted number from the BRDF measurement. Assuming this light is not coming from the target PD, we want to reduce the scattering from other optics. As an attempt, we decided to use a BS instead of the combination of a cube PBS and a QWP. This way we can reduce the number of the optics involved, particularly the PBS which may cause more scattering than others.

What we did:

The PBS and the QWP were removed, and then a 50:50 BS was inserted as shown in Fig. 1.

The distance between the target PD and Iris2 was measured to be 38.1 cm. The diameter of Iris2 was set to be 7 mm.

As a part of the calibration, the incident power on the target PD was measured using PDA100A (Gain 0dB). The measured value was 3.973 V with no chopper blocking the beam.

An HR mirror was then placed before the target PD to measure the reflected light power with the chopper still halted and the iris 1 is oped using PDA100A (Gain 0dB) which is placed at the position as shown in Fig. 1. It was measured to be 1.253 V. This means that the reflectivity of the BS is about 0.315. This reduction is considered to be the clipping at Iris1 as the returning beam has the bigger beam size (i.e. the iris worked as a sort of mode cleaner).

(==> There might be a misunderstanding because of my not clear explanation. So, I edited this paragraph again. KN)

Then, the chopper was started at 253 Hz. The reflected light from the HR mirror was measured using a FFT analyzer to be 642 mVrms (PDA100A gain 0 dB).
(==> mVrms? KA ==> mVrms. I fixed. KN)

Throughout the measurement, the opening of Iris1 was adjusted such that the clip level was less than 0.1% of the incident power (see also elog2047).This yielded the Iris1 diameter of 1.8 mm.

Finally, the back reflection was measured in the same way as elog 2040 , and also with Iris1 opend as much as possible.


The measured voltage with Iris1 with the diameter of 1.8 mm: 720 uVrms (PDA100A gain 70 dB)
==> This corresponds to the BRDF of 1.4 x 10-3 [1/sr].

The measured voltage with Iris1 completely open: 640 uVrms (PDA100A gain 70 dB)
==> This corresponds to the BRDF of 1.2 x 10-3 [1/sr].

These values are much (x40) larger than the BRDF value expected from the BRDF measurement (5x10-5 [1/sr], see also elog2045).
In fact, these are much (x2) larger than the ones with the PBS & the QWP (8x10-4 [1/sr], see also elog 2047).

(It is not so clear how these numbers were calculated. How these calibrations were used? How did you incorporate the BS transmissivity and reflectivity??? KA)

==> For calculating the BRDF, the fomular as shown elog2034 is used. Seeing the fomular, we need the ratio of the incident light power and the scattered (reflected, in this measurement) light power, the incident angle, and the solid angle. The ratio of the incident light power and the scattered light power can be obtained by taking the ratio of the measured voltage with the chopper as a part of the calibration, Vin (642 mVrms), and the measured voltage with Iris1 with the diameter of 1.8 mm (720 uVrms) or with Iris1 completely open (640 uVrms), Vsca, considering the difference of the PDA100A gain. In this calculation the BS reflectivity and transmissivity do not appear, see also Fig. 2. In addition, using the incident angle, 15 deg, and the solid angle from the PD(C30665GH) to the Iris2, 2.6x10-4 sr, the BRDF can be calculated.

And when the Iris was closed, the value was larger. This means that the Iris may make a scattering.
(==> This didn't make sense compared with the numbers above. KA ==> The numbers were written conversely. I fixed. KN)

Fig. 1 Setup for measuring the back reflection using the BS.
Fig. 2 BRDF calculation with chopper and the BS. The left side shows that the setup of the measurement of the back reflection light. The right side shows that the setup of the measurement of the incident light. The bottom fomular shows how to obtain the ratio between the incident light power and the scatterd light power from the voltage measured with the FFT analyzer. 

[Ed.: KA]

Attachment 1: setup.pdf
Attachment 2: BRDF_cal.pdf
  2049   Mon Mar 7 15:47:03 2016 KojiNMiscPD QEBack reflection measurement with the chopper

To reduce the scattering, we did some try and error.

The setup was same as the elog2048.


First, to reduce the scattering from the Iris 1, in front of the Iris 1 the aluminum foil which has the hole for the laser was placed.

We made several foils, different hole sizes and different shapes.

We cannot found the reasonable tendency, but with the best foil the measured scatterd value became 350 uVrms (BRDF = 7.1 * 10^(-4) [1/sr]).

This is the comparable value to the BRDF measured with PBS and QWP (elog2047).


After that, we changed the steering mirror #2 from Newfocus 5104 to Newport 10Q20HE.1.

Then, without aluminum foil, i.e. in completely same setup as elog2048, the measured scatterd value became 11 mVrms.

And we tried to place the best aluminum foil and the measured valued as reduced to 820 uVrms.

  2093   Fri Apr 28 13:54:35 2017 AidanLaserPD QEDET10D QE at 2004nm measured (it was 0.79)

I set up a measurement of the DET10D QE at 2004nm. I supplied the Eblana 2004nm fiber-coupled laser diode with 45mA of current. I first measured the output with the Thorlabs power meter and then measured output with the DET10D photodiode. Both systems are fiber-coupled (although both fiber couplers are screwed on rather than glued on).

The power was 0.34mW = 3.425E15 photons per second

The photodetector output was run through a 50 Ohm resistor which was then run through a SR560 with 100x gain at DC. The measured voltage from the SR560 was 2.183V. Hence the photocurrent was 4.36E-4A  = 2.7E15 electrons per second

Therefore, the QE was 0.79 at 2004nm. This agrees with the manufacturers curve. However, I notice that there:

  • is a window over the photodiode (so we might lose a few percent there). It's not clear from the manual whether this is AR coated or not.
  • the voltage from the SR560 fluctuates a bit when I rotate the fiber coupler that is attached to the front (I measured the QE when this was maximized). This tells me that we’re not centered on the photodiode and a fraction of the fiber output isn’t getting onto the active area of the PD.  



Attachment 1: DET10D_QE.pdf
Attachment 2: IMG_9396.JPG
Attachment 3: IMG_9395.JPG
Attachment 4: IMG_9394.JPG
  2094   Tue May 2 08:26:30 2017 Aidan, AndrewLaserPD QEQE of regular InGaAs photodetectors is 16ppm at 2004nm

Last night, Andrew and I measured the QE of two regular InGaAs photodiodes at 2004nm using the fiber-coupled Eblana laser diode as a source.

The two diodes in question were a Thorlabs PDA10CS (adjustable gain set to +60dB) and a fiber-coupled NewFocus 1811

Incident power:

We set the current to 55mA which provided 495 micro Watts of power at 2004nm. Each photon has an energy of 9.9E-20J. This corresponds to 5.0E15 photons per second.

Photodiode response:

Each diode had internal transimpedance gain (R_T) and the output was connected to an SR560. The voltage difference between the diode being OFF and ON was measured. 

Diode Voltage difference SR560 Gain Voltage across R_T R_T Photo-current e- per second
1811 87 mV 2000 43.5E-6 V 1E4 Ohms 4.35E-9 A 2.7E10
PDA10CS 384 mV 20 19.2 mV 1.5E6 Ohms 1.28E-8 A 8E10

Electrons per Amp = 6.25E18

Quantum Efficiency:

  • 1811: 2.7E10 / 5E15 = 5.4E-6 electrons per photon
  • PDA10CS: 8E10 / 5E15 = 16E-6 electrons per photon
  2097   Tue May 2 14:08:52 2017 Aidan, AndrewLaserPD QEQE of regular InGaAs photodetectors is 16ppm at 2004nm

The silver lining here is that the 'extended' InGaAs only has a 3x higher QE than your standard InGaAs.

Are you sure you got all the light on the 1811? Don't you have a larger aperture diode to use than that little thing?

Also, I think the new focus 1811 DC transimpedance is 1 k, not 10 k.

Attachment 1: Screen_Shot_2017-05-02_at_2.10.30_PM.png
  2421   Tue Sep 17 23:42:41 2019 Shalika SinghLaserPD QEMeasuring Quantum Efficiency of Extended InGaAs Photodiode

**[Internal Quantum Efficiency added]

[Koji, Shalika]

Further measurements were done after elog:2419 for Quantum Efficiency of Extended InGaAs Photodiodes(X8906). A Laser of wavelength 2um was used with an incident power of 0.80+0.02mW.  The Ophir RM9 power meter was used to check the incident power and also measure the reflectivity.

Attachment 1: The Setup. A Fibre launcher was used to project the laser along with a converging lens of the focal length of 40.0 mm which was further arranged with a subsequent converging lens of  150mm focal length. A mirror was used to reflect the laser light on the photodiode at an angle of 45o. The bias voltage was provided to pin 4 of photodiode using a Sallen Key low pass filter and the output at pin 3 of the photodiode was fed to a transimpedance amplifier (with a gain of 5.1k) which converted the photocurrent to voltage.

Attachment 2: The Quantum Efficiency is plotted with respect to different bias voltages, It was observed that the quantum efficiency increases with an increase in bias voltage. An External Quantum Efficiency of 77.4% was observed at 1V(maximum bias voltage for the photodiode). The Internal Q.E was observed to be 83.8% taking into account Reflectivity of (60.0+1) uW at an angle of 17deg. 

Attachment 3: To recreate all data

Attachment 1: IMG_8915.JPG
Attachment 2: QE_X8906.pdf
Attachment 3: Extended_InGaAs.zip
  2423   Mon Sep 23 10:49:27 2019 KojiSummaryPD QEQE and dark current of InAsSb sensors

The QE and dark current of all the InAsSb sensors were measured. All the measurements were done in room temperature.
- The incident beam power of the 2004nm beam was 0.95mW.
- The beam was focused down to 50um gaussian radius, which was confirmed by DataRay BeamR.
- The angle of incidence was ~0deg.

- The element side (nominally Pin 2, 3, or 6) were connected to the vias boltage (negative) and the common ground was connected to the transimpedance amplifier (Shalika OP140 R=5100Ohm)
- The dark current was highly dependent on the reverse bias voltage. The QE was also bias dependent.
- Sb3512 A2 have different behavior compared to others. Alex mentioned that Sb3512 is the test batch. We can exclude this sensor from the test.
- The best QE was ~0.7 for Sb3513 A3 P2 (Pink) and Sb3513 A2 P6 (Purple). Both have the area of 500um^2. These two particular elements have low dark current of <1mA. The dark noise of this specific sensor should be measured.

Some issues of the measurements

- The transimpedance amp (TIA) has suspicious behavior. The saturation voltage was ~17V rather than <-15V. This indicates that the voltage regulators possibly have leakage of the input voltage (+/-18V) to the output line. This needs to be checked, particularly before the dark noise test.

- TIA saturation: The bias voltages could not be raised to ~1V for some PDs because of the dark noise and the saturation of the TIA. The transimpedance should be lowered by a factor of ~5.
- Because of the low bias voltages of these saturated cases, the max QEs were not reached. This also prevented from checking if there was any clipping loss. This should be checked again with the lower transimpedance.

- TBD: The angular dependence and the reflectivity of the sensor should be checked. It is difficult to carry out these tests without a sensor card.

Attachment 1: InAsSb_QE.pdf
Attachment 2: InAsSb_DarkCurrent.pdf
Attachment 3: 190921_SbPD_QE.zip
  2425   Wed Sep 25 01:05:30 2019 KojiSummaryPD QEQE and dark current of InAsSb sensors

The lenses were arranged so that the spot on the PD can become smaller. A quick measurement on a (500um)^2 element showed the QE of ~80%

With the strong focusing lens of f=40mm, the beam was once expanded to a few mm. Then f=75mm lens focuses the beam to ~30um (radius). (See Attachments 1&2)

With this new beam, the QE was quickly checked. The new measurement is indicated as "Sb3513 A2P6new" in the plot. It showed the QE of ~80%.
The AOI was scanned to find any maximum, but the AOI of 0deg was the best at least with the given beam. I'm not sure yet why 500umx500um requires such small beam radius like 30um. Awesome

Attachment 1: P_20190924_233507_vHDR_On.jpg
Attachment 2: P_20190925_003614_vHDR_On.jpg
Attachment 3: InAsSb_QE.pdf
Attachment 4: InAsSb_DarkCurrent.pdf
  2438   Thu Oct 31 18:31:10 2019 KojiLaserPD QEPD EQE vs Spot size

InAsSb PD QE Test

The relationship between the spot radius and the apparent QE (EQE) was measured.

1) The spot size was checked with DataRay Beam'R2. The beam scanner was mounted on the post with a micrometer stage in the longitudinal direction. (Attachment1 upper plot)
It was confirmed that the beam is focused down to ~22um. The incident power was about 0.9mW.

2) The InAsSb detector (Sb3513A2) was mounted on the PD holder and then mounted on the stage+post. The photocurrent was amplified by a FEMTO's transimpedance amp (V/A=1e3Ohm). The dark current and the total photocurrent were measured at each measurement point with the beam aligned to the PD every time. The estimated EQEs were plotted in the lower plot of the attachment.

Note that P2, P3, and P6 elements have the size of (500um)^2, (750um)^2, and (1000um)^2, respectively.

The absolute longitudinal position of the sensor was of course slightly different from the position of the beam scanner. So the horizontal axis of the plots was arbitrary adjuted based on the symmetry.

The remarkable feature is that the QE goes down with small spot size. This is suggesting a nonlinear loss mechanism such as recombination loss when the carrier density is high.

With the present incident power, the beam size of 100um is optimal for all the element sizes. For the larger elements, a bigger beam size seems still fine.

The next step is to estimate the clipping loss and the saturation threshold with the Gaussian beam model.

Attachment 1: QE_vs_spotsize.pdf
  2439   Fri Nov 1 12:47:18 2019 KojiLaserPD QEPD EQE vs Spot size

Clipping and saturation were investigated by the semi-analytical model. In the analysis, the waist radius of 20um at the micrometer position of 8mm is used.

1) Clipping

Firstly, the clipping loss was just geometrically calculated. Here the saturation issue was completely ignored. The elements P6, P3, and P2 have the sizes of (500um)^2, (750um)^2m, and (1000um)^2, respectively. However, these numbers could not explain the clipping loss observed at the large spot sizes. Instead, empirically the effective sizes of (350um)^2, (610um)^2, and (860um)^2 were given to match the measurement and the calculation. This is equivalent to have 70um of an insensitive band at each edge of an element (Attachment 1). These effective element sizes are used for the calculation throughout this elog entry.

2) Saturation modeling

To incorporate the saturation effect, set a threshold power density. i.e. When the power density exceeds the threshold, the power density is truncated to this threshold. (Hard saturation)

Resulting loss was estimated using numerical integration using Mathematica. When the threshold power density was set to be 0.85W/mm^2, the drop of QE was approximately matched at the waist (Attachment 2). However, this did not explain the observed much-earlier saturation at the lower density. This suggests that the saturation is not such hard.

In order to estimate the threshold power density, look at the beam size where the first saturation starts. The earlier sagging of the QE was represented by the threshold density of 0.1W/mm^2. (Attachment 3)

Attachment 1: QE_vs_spotsize_no_saturation.pdf
Attachment 2: QE_vs_spotsize_saturation_0_85.pdf
Attachment 3: QE_vs_spotsize_saturation_0_1.pdf
  2443   Tue Nov 12 03:40:39 2019 KojiLaserPD QEPD EQE vs Spot size

The QE of the (500um)^2 element has been tested with a half-power (0.51mW) instead of 0.92mW.
It is clear that the central dip depth is reduced by the lower power density.


Attachment 1: QE_vs_spotsize_half_power.pdf
  2459   Mon Nov 25 15:03:34 2019 KojiUpdatePD QEIn solder and PD mounts are in

The PD mounts were delivered from ProtoLabs. The order was sent on Tue last week and it's here on Monday. Excellent!
And the quality looks pretty good.

The surfaces are sandblasted. Do we want to do any process on the bottom surface to reduce the thermal resistance?

An indium solder string also came in.

Attachment 1: PB259778.JPG
Attachment 2: PB259780.JPG
Attachment 3: PB259781.JPG
  2460   Mon Nov 25 21:46:56 2019 KojiSummaryPD QESystem Diagram

System diagram of the PD QE test with the IRLabs cryostat.


PT-SE (MS/PT-SE) connector data sheets

Connector/receptacles/tools https://www.peigenesis.com/images/content/pei_tabs/amphenol/pt-ptse-series/new-thumbs/123-146_pt_series.pdf
Amphenol catalog http://www.amphenol-industrial.com/images/catalogs/PT.pdf

Detoronics Hermeic Sealed Connectors (DT02H-18-*PN) http://www.hselectronics.com/pdf/Detoronics-Hermetic-Connectors.pdf

AF8 crimping tool (expensive!) https://www.mouser.com/ProductDetail/DMC-Tools/AF8?qs=gvhpkjpQEVSjrLbsepewjg%3D%3D
AF8 alternative https://www.jrdtools.com/?gclid=Cj0KCQiA2vjuBRCqARIsAJL5a-IQ9ztCEYKdo645v_RhUBJS3eMIars1LubjlKZoorS-lnx6ClDDiMUaAlZiEALw_wcB


Thermistor link: https://www.tec-microsystems.com//Download/Docs/Thermistors/TB04-222%205%25%20Thermistor_Specification_upd2018.pdf

TEC spec: Mounted TEC type: 2MD04-022-08/1 https://www.tec-microsystems.com/products/thermoelectric-coolers/2md04-series-thermoelectric-coolers.html
2MD04-022-08/1 dTmax = 96, Qmax = 0.4W, Imax = 0.7A,  Umax = 2.0, ACR = 2.29 Ohm

Attachment 1: cryo_pd_test.pdf
cryo_pd_test.pdf cryo_pd_test.pdf
Attachment 2: InAsSb_PD_mount_short.PDF
Attachment 3: PD_pin.pdf
  2462   Tue Nov 26 18:49:11 2019 KojiUpdatePD QESocket soldering test piece made

Normal solder (Sn63 Pb37): with flux, wetting o

Pure Indium - In 99.995: no flux, wetting x, low melting temp, like paste

Pb93.5 Sn5 Ag1.5: with flux, wetting o, high melting temp (soldering iron setting 380~430F)

Cryo solder In97 Ag3: no flux, wetting x, low melting temp, like paste

Attachment 1: IMG_9118.jpeg
Attachment 2: IMG_9120.jpeg
Attachment 3: IMG_9121.jpeg
Attachment 4: IMG_9123.jpeg
Attachment 5: IMG_9125.jpeg
Attachment 6: socket.pdf
  2463   Wed Nov 27 20:38:57 2019 KojiSummaryPD QESystem Diagram

The external Dsub cable is ready except for the 32pin connector to be plugged-in to the chamber. See QIL ELOG 2460 for the pin assignment.


Attachment 1: P_20191127_203612_vHDR_On.jpg
  2464   Sun Dec 1 01:32:19 2019 KojiSummaryPD QEPD TEC cooling test

While I'm still waiting for the proper connector for the vacuum feedthru of the IRLabs cryostat, I have connected to the Dsub9/15 split cable to another Dsub9 connector so that I can test the cooling of the InAsSb sensor in air. Also, the 2004nm laser, a fiber-coupled faraday isolator, and 90:10 beam splitter was moved to the cryostat table and fixed on a black al breadboard. [Attachment 1]

The InAsSb TEC was controlled by the TEC controller of ITC-50. I didn't change the PID parameters of the controller but the temperature nicely setteled to the setpoint. The sensor has a 2.2kOhm thermister. And the max current for the TEC was unknown. The TEC driver had the current limiter of 0.3A and it was not changed for now. With this current limit, the thermistor resistance of 10Kohm was realized. This corresponds to the temperature of about -20degC. According to the data sheet given by Alex, the resistance/temperature conversion is given by the formula

1/T = 7.755e-4 + 3.425e-4*log(R)+1.611e-13*log(R)^3


  • What is the max current for the TEC?
  • What is the calibration formula of the thermister (TB04-222) at cryogenic temperature?
    -> Thermistor datasheet link (pdf)

To satisfy the curiosity, the dark current of a (500um)^2 element was measured between -250K and -300K. At -254K, the dark current went down to the level of 40uA (1/15 of the one at the room temp). For the measurement, the bias voltage was set to be 0.5 and 0.6V. However, it was dependent on the diode current. (Probably the bias circuit has the output impedance). This should be replaced by something else.

To Do

  • 2um laser beam setup (w=100um beam)
  • Bias circuit
  • Quick check of the QE and dark noise at -20degC


Attachment 1: 20191129191814_IMG_9146.JPG
Attachment 2: cooling_dark_current.pdf
  2465   Tue Dec 3 13:52:04 2019 KojiUpdatePD QESocket soldering test piece made

[Raymond and Koji]

We dunked the PD socket test piece into LN2 and repeated heat cycle 8 times. No obvious change was observed. Then the wires were pulled to find any broken joint or etc.
None of the solder joints showed the sign of failure.

For cleanliness, we are going to use In-Ag solder (no flux) for the actual wiring.

Attachment 1: Frozen connector

Attachment 2-4: Inspection after thawing.

Attachment 1: PC029784.jpeg
Attachment 2: PC029788.jpeg
Attachment 3: PC029786.jpeg
Attachment 4: PC029787.jpeg
  2466   Tue Dec 3 15:32:39 2019 KojiSummaryPD QEPD TEC cooling test

The quantities we want to measure as a function of the temperature:

- Temperature: 2.2k thermister resistance / 100ohm platinum RTD

- QE (Illuminating output / Dark output / Reference voltage / Reference dark output)

- Dark current (vs V_bias) -> Manual measurement or use a source meter

- Dark noise (PSD) 100kHz, 12.8k, 1.6kHz, 100Hz


  2468   Thu Dec 5 13:50:59 2019 KojiSummaryPD QEDark current measurement with the sourcemeter

I borrowed KEITHLEY 2450 source meter from Rich. The unit comes with special coaxial cables and banana clips. Most of the peripherals are evacuated in the OMC lab.

The dark current of A2P2, A2P3, A2P6 were measure with different temperatures (300K, 270K, 254K). The plot combined with the previous measurement ELOG QIL 2425.

== How to use the source meter ==

- Two-wire mode: Connect the wires to the diode

- Over voltage protection: [MENU] button -> SOURCE / SETTINGS->Over Voltage Protectiuon 2V

- Sweep setting: [MENU] button -> SOURCE / SWEEP -> e.g. Start -750mV, Stop +500mV, Step 10mV, Source Limit 1mA -> Select Generate

- Graph View: [MENU] button -> VIEWS / GRAPH

- Start measurement: Note: The response of [TRIGGER] button is not good. You need to push hard
  This starts the sweep, or a menu shows up if your push is too long -> Select "Initiate ..."

- Data Saving: [MENU] button -> MEASURE / READING BUFFERS -> Save to a USB stick

Attachment 1: InAsSb_DarkCurrent_markedup.pdf
  2470   Mon Dec 9 02:15:39 2019 RaymondDailyProgressPD QEIR Labs Cryostat

The IR Labs cryostat has its internals wired and attached to the baseplate. PD A2 was clamped and the vacuum pumps turned on for the first cooling test.

[in the morning I will update with a detailed pin-out and label the attached photo (labeled 12/13, pin out in separate post)]

Attachment 1: IRLabscryointernals.pdf
  2471   Mon Dec 9 12:34:14 2019 KojiDailyProgressPD QE 

I can see some screws are not vented. You also need to use a vented screw for the additional temp sensor if the face screws of the PD mounts are not vented.

You can use a bunch of clean clamps and screws I brought. They are in a mylar bag.
If you need more vented screws, please specify the size and length. I can grab some from the 40m cleanroom.

  2472   Mon Dec 9 14:03:52 2019 ChrisDailyProgressPD QEToward PD testing automation
  • A Keithley 2450 source meter will be used to measure QE and dark current vs bias.  It is on the QIL network as  A command interface is available via telnet for running measurements and grabbing data.
  • A FEMTO transimpedance amp and SR785 will be used to take dark noise spectra, with bias supplied by a DAC channel filtered by an SR560.  The 785 is on the QIL network as, and the usual GPIB scripts should be sufficient.
  • A relay has been installed to control the switching between the Keithley and SR785 readouts.
  • BNC cables and breakouts connect the electronics on the table with the cymac (see table below for the channel assignments at the moment).
Cymac channel Assignment
ADC 29 Reference PD out
ADC 30 TED200C temperature out
ADC 31 ITC510 TEC temperature out
DAC 12 LDC201C laser current ctrl
DAC 13 SR560 (PD bias control for dark noise measurement)
DAC 14 ITC510 TEC temperature tune
DAC 15 Relay control (0 ct = Keithley, 30k ct = SR785)

Note: in the preceding table, channel numbers use the digital convention (numbered starting from zero), which is not the convention used by the AA/AI chassis front panel (numbered starting from one).

  2473   Mon Dec 9 14:45:45 2019 KojiDailyProgressPD QEToward PD testing automation

Wow. This is great, thanks Chris.


  2475   Wed Dec 11 01:29:26 2019 KojiSummaryPD QESb3513 A2P6 Dark Current / QE / Dark Noise measurement @77K

[Raymond, Aidan, Chris, Koji]

P6 element (500um)^2

- We looked at the current amp (FEMTO) output. The amplifier saturated at the gain of 10^3 V/A. Looking at the output with a scope, we found that there is a huge 1.2MHz oscillation. Initially, we thought it is the amplifier oscillation. However, this oscillation is independent of the amplifier bandwidth when we tried the our-own made transimpedance amp.

- Shorting the cryostat chamber to the optical table made the 1.2MHz significantly reduced. Also, connecting the shield of the TEC/Laser controller made the oscillation almost invisible. This improvement allowed us to increase the amp-gain up to 10^7.

- Then the dominant RMS was 60Hz line. This was reduced by more grounding of the cable shields. The output was still dominated by the 60Hz line, but the gain could be increased to 10^8. This was sufficient for us to proceed to the careful measurements.


- The dark current was measured by the source meter, while the photocurrent (together with the dark current) was measured under the illumination of the ~1mW light on the PD.

- Attachment 1 shows the dependence of the dark current against the swept bias voltage. We had ~mA dark current at the room temp. So, this is ~10^5 improvement.

- Attachment 2 shows the dependence of the apparent QE against the swept bias voltage. The dark current was subtracted from the total current, to estimate the contribution of the photocurrent in the measurement.

- Attachment 3 shows the dark noise measurement at the reverse bias of ~0.6V. Up to 1kHz, the noise level was below the equivalent shotnoise level of 1mA photocurrent.


All the data and python notebook in the attached zip file.

Attachment 1: Sb3513_A2P6_DarkCurrent_77K.pdf
Attachment 2: Sb3513_A2P6_QE_77K.pdf
Attachment 3: Sb3513_A2P6_DarkNoise_77K.pdf
Attachment 4: 191210_3513A2P6.zip
  2478   Thu Dec 12 02:10:37 2019 Raymond, ChrisDailyProgressPD QEPD test dry run

(Raymond, Chris)

  • Baseplate and shield RTD temperature sensors are working, and digitally recorded.  The temperature sensing diode on the PD mount seems to be unconnected.
  • We made dark current vs bias measurements on the P6 and P2 elements at room temperature.  We also checked P3 but it appears to be unconnected.  Then, after realigning the P6 element, we measured its photocurrent vs bias at room temperature.  The photocurrent and dark current data looks much like what Koji previously reported at room temperature.
  • Going from 77 K to room temperature, the beam had to be moved downward to correct the alignment.  The magnitude was about 1 clockwise turn of the vertical alignment knob on the steering mirror.  The horizontal shift was very small.
  • Armed with this knowledge, we re-wetted the cryostat with LN2, bringing the baseplate and shield temperatures down into the 80 K range.  Our plan was to let it warm up overnight while running an automated test series of dark current and photocurrent vs bias and temperature.  We offset the vertical alignment by a half turn, in hopes of having the PD well aligned near the expected QE sweet spot of 150-200 K.
  • The procedure for adding LN2 to the cryostat involves temporarily disconnecting the fiber coupler.  Sadly, it appears in our haste we did not fully recover the alignment after this step, or else the calibration somehow shifted by a lot.  The photocurrent is at the 0.1 mA level instead of the ~1 mA we expect.  Accordingly, this run will primarily be useful as a test of the automation, as an indication of dark current vs temperature, and as a very rough, qualitative indication of QE vs temperature.
  • The script is running on qil-ws1 in a screen session.  It can be interrupted with screen -RAad autorun to connect to the session, followed by Ctrl-C to kill the script.
  • In the morning, we need to decide whether to open the cryostat and fix the wiring, or to repeat this run with tighter control over the alignment.
  2479   Fri Dec 13 01:51:15 2019 RaymondDailyProgressPD QECryostat wiring fixes
  • Monitored an autorun of the photocurrent for A2P6 from about 77K-300K and adjusted the laser alignment as necessary. I've not yet looked through the data, as I was keen on fixing the assorted wiring issues in time to pump down for tomorrow. Chris found the laser power issue rooted in my sub-optimal diode-collimator mating connection, ie I didn't fully plug back in the laser. I've learned my lesson and will simply shield the fiber with Al foil for future nitrogen pouring endeavours .
  • Opened the cryostat and confirmed the pin assignments (see attached list).
  • Cathode 2 was shorting on the thermal anchors for the PD plug. I re-wrapped the quad twist wire around the bobbins and this has fixed the issue, though I did not see any portion of the quad twist wires that was missing the Formvar insulation. 
  • The DT-670 silicon diode thermometer was busted, this was likely the case from the start and should have been checked (by me) prior to wiring. I've replaced this diode with another 4-lead platinum resistor, it is plugged into input 3 on the CTC100 (still labeled 'PDdiode'). 
  • The thermistor on A2P6 is still intact, showing about 1.9 kΩ at room temp

Still to do:

  • Move the radiation shield PT RTD to the outer shield
  • Re-wire the baseplate pins in the DSUB9 connection to the CTC100 temp controller (switched V+ and V-)
  • Check feasibility of adding 25Ω heater to the sample mount to stabilize the temp of the PD rather than measuring while sweeping 


Attachment 1: 18_32malepinsinairAssigned.pdf
  2480   Mon Dec 16 18:16:31 2019 RaymondDailyProgressPD QECryostat wiring fixes

Opened the cryostat to resolder the heater and re-wrap the thermal anchor for the sample RTD and PD connection. All connections are working as expected at room temperature. A2 is still in the sample mount. 

  2481   Fri Dec 20 13:20:53 2019 AidanSummaryPD QEQE results from A2P6 (500um) and A2P2 (1mm)

The QE measurements from the first couple of photodiodes are attached below.

  • plot_JPL_diode_results.m - A2P6 analysis
  • plot_JPL_A2P2_diode_results.m - A2P2 analysis

QE = [I_photocurrent]/[P_PD] * h *nu/e

P_PD = Power incident on photodetector = 0.966*power_incident on cryo window

Power incident on cryo window = F(voltage on reference PD)

Attachment 1: PC_DC_v_T.pdf
Attachment 2: A2P2_001_test.pdf
Attachment 3: PC_DC_v_T.pdf
Attachment 4: A2P6_001_test.pdf
Attachment 5: plot_JPL_diode_results.m
% load JPL data
f0 = dir('*dark*.txt');
f1 = dir('*photo*.txt');
f2 = dir('*cond*.txt');

% get temperature vs time
tempList = [];
pList = [];
for ii = 1:numel(f2)-1
... 102 more lines ...
Attachment 6: plot_JPL_A2P2_diode_results.m
close all 
clear all
% load JPL data
f0 = dir('*dark');
f1 = dir('*bright*');

% get temperature vs time
tempList = [];
refPDList = [];
for ii = 1:numel(f1)
... 113 more lines ...
  2482   Fri Dec 20 21:58:14 2019 KojiUpdatePD QEPD TEC driver / A2P6 aligned / Lens moved

== Currently, A2P6 is aligned ==

1) I've brought another TEC driver fro the PD temp control. This unit was borrowed from the 2um ECDL setup. Eventually, we need to return this to ECDL. (Attachment 1)
The PID loop of the TEC control works. But it is not well optimized yet. If you change the target temp too quickly, the TEC out seemed oscillating. Watch the TEC out carefully and change the temp setpoint slowly.
So far I have tried to cool the thermister up to 30kOhm (~232K) and I_TEC was 0.33A.
I did not try further. I felt it was better to cool the PD base for further trial.

2) A part of the alignment study, the beam is aligned to A2P6. Also, the lens position was investigated, and I decided to move the lens ~1 inch away from the window.  (Attachment 2)
    In fact, this allowed us to insert the power meter between the lens and the window. 

Attachment 1: P_20191220_192440_vHDR_On.jpg
Attachment 2: P_20191220_180929_vHDR_On.jpg
  2483   Fri Dec 20 22:26:19 2019 KojiUpdatePD QEPD TEC driver / A2P6 aligned / Lens moved

The QEs were measured at 293K, 239K, 232K, and 293K again. The cooling was provided by the PD TEC.  At each temperature, the incident power was changed from 30uW to 1mW to see the dependence of the QE on the incident power to check the possible saturation.

The QE was 79~81% (the window T=96.6% was already compensated). I'm not 100% sure this 1% variation in the plateau is real or due to insufficient calibration of the REF PD.
The REF PD was calibrated at 1mW at 100mA injection current to the laser.

No obvious saturation was observed.

We can cool the PD with LN2 and we should make a careful alignment of the beam at each temperature.

Attachment 1: Sb3513_A2P6_DarkCurrent_293K.pdf
Attachment 2: Sb3513_A2P6_DarkCurrent_239K.pdf
Attachment 3: Sb3513_A2P6_DarkCurrent_232K.pdf
Attachment 4: Sb3513_A2P6_DarkCurrent_293K_2.pdf
Attachment 5: Sb3513_A2P6_DarkCurrent_Comparison.pdf
Attachment 6: 191220_3513A2P6.zip
  2500   Fri Jul 24 05:22:30 2020 RaymondLab InfrastructurePD QERound 2 of JPL PD's in lab

Alex dropped off the new round of 2um PD's, they're on the north table accompanied by his data sheet.  

Attachment 1: JPL_PDs_2.jpg
  1937   Wed Jun 17 18:05:42 2015 ArjunElectronicsPD noiseResistor noise observations,analysis and inference

This eLog pertains to the resistor noise setup, which is a prelogue to the photodiode noise measurement setup. Which is expected to have noise with physics similar to that of resistors.


Resistors have been observed to contain excess noise in addition to the Johnson-Nyquist noise present in them. But, unlike the Johnson noise which has a flat power spectrum this noise has been observed to have a spectrum with a power law dependence \propto f^\beta, where \beta has the value very close to -1. Additionally, the power of this noise depends on the current passing through it. This excess resistance fluctuations in resistor is a very well documented subject for both DC and AC techniques(see [1] [4] [9],[10] for DC and [3] [8] for AC). The simplest of DC measurement would be to bias a given test resistor at a constant current and the measure the volatge fluctuations about its average value. A schematic of this image can be seen below.

The simplest of excess noise measurement topologies(taken from [9])


 Measurements with such a setup have been done(see [9]). But in such topologies the fluctuation in the sources need to be extremely small as it would manifest itself as voltage fluctuations. A upgrade of this setup would be to use a balanced bridge setup, as a perfectly balanced bridge would get rid of any source fluctuations and the resistance fluctuation can then be read out directly(see [4] [5] [8] [3] [1]). But doing this experiment with DC sources has additional challanges like the setup requires additional themal shielding as low frequency noise sources like thermal drift etc would also manifest themselves. Additionally the readout components(like amplifier etc) also have a noise spectrum which increases at low frequencies, various methods have been used to remove this additional amplifier noise(see [1] [4] [5] [10]). The best method for such a measurement would be using a AC drive to shift the spectrum and perform operations at higher frequencies where our electronics have much less noise and the demodulate back to retrieve the spectrum. The method used is partly inspired by the analysis done for crackle noise(see [2]) and partly by [8].


Model Used

Consider the schematic below which is a balanced bridge with a AC drive.

Basic Bridge Setup


Let the resistance fluctuations in R1,R2,R3 and R4 be{\delta R}_1, \delta R_2, \delta R_3 and \delta R_4 respectively. The volage difference between the two nodes 1 and 2 can be then wriiten as:

\Delta V_{12}=\frac{\delta R_4-\delta R_3+\delta R_1-\delta R_2}{2R_0}V_d

This is arrived by basic nodal analysis of the circuit and using the approximation that \frac{\delta R}{R_0}\ll1 along with binomial approximations. Now based on previous experiments on excess resistor noise at DC, it was conclusively proven that the  excess voltage noise(\Delta V_{12})  depends linearly on the drive voltage (see [1] [7] [8]). Using that we can conclude that the resistance fluctuations then cannot contain any drive dependence as drive dependence is already covered by the V_d in the above equation.  Based on this conclusion we can conclude that the resistance model can then be modeled as:

\delta R_i=\alpha g(t)

Where g(t) refers to the underlying stochastic process, which is well known to have a  \frac{1}{f} spectrum(see: basically any of the listed references). 

As a result the net bridge differential voltage would be:

\Delta V_{12}=\frac{\alpha \delta R_{net} V_d}{R_0}

Where \delta R_{net} refers to the inoherent addition of the fluctuations in the 4 resistors.The spectrum of \delta R_{net} will be the same as the spectrum of resistance fluctuations of of any one , as they are independent and uncorrelated.

For the AC measurement technique the drive would be a sinusoidal one. Let the drive be V_d=V_0 \sin\omega_0t. The bridge differential volatge would then be:

\Delta V_{12}=\frac{\alpha\delta R_{net}V_0\sin\omega_0t}{R_0}

The spectrum for this quantity would be the spectrum of \delta R_{net} shifted to \omega_0 plus the additional background noise in the system.It can be proved by a liitle bit of math that if such a signal is demodulated in a lock-in amplifier( ie multiplied with \sin\omega_0t followed by a low-pass filter) that the resultant spectrum(for a detailed  derivation see appendix of [8])

S_v(\omega,i_0)\approx G^2[{S{_v}}^{0}(\omega_0-\omega)+\frac{i_0^2}{2}}S_R(\omega)\cos^2\Delta \theta]

Where \omega_0 is the frequency of the drive, G is the gain of the lockin, S_v^0(\omega) is the spectrum of the noise at zero drive and \Delta\theta is the phase difference between the lockin's reference signal and the signal which is to be mixed and i_0 is the peak current or it can also be written as \frac{V_0}{2R_0}. The thing to note is that at \Delta\theta=0 that is spectrum of the in-phase component is the sum of both the excess resistor noise and the background noise whereas at \Delta\theta=90^{\circ}, that is in the quadrature phase spectrum only the background noise is present. Hence on finding the spectrum of in-phase and quadrature component we could subtract the two to obtain the desired noise which is S_ R(\omega). This is the primary method that has been employed to try and find the excess noise present. Additionally we could band-pass the bridge differential voltage so as to suppress unwanted frequencies, this band-pass would be around \omega_0. The next section deals with the specifics of the experimental setup used and associated results.

Experimental Schematic and Observations

The following image is an experimental schematic of the setup used to measure excess noise in resistors.


Experimental Schematic

 The first 'sample' bridge consisted of 5% carbon film resistors with a nominal value of  R_0=1.5 k\Omega, these were hand matched to 5 \times 10^{-3}\%. The differential bridge nodes were connected to a low-noise instrumentation amplifier- AD620,the gain was adjusted using a gain resistor and was set to 100(R_{gain}=499\Omega) the output of the bridge was then connected to a cascaded low pass and high-pass filter to produce a bandpass filter using SRS560 dual filter. The output of the filter is connected to the input of the SRS830 DSP lockin amplifier. Apart from the other amazing features, this lockin has both in-phase and quadrature outputs this is particularly useful as the two outputs can be recorded simultaneously. The two data out of the two output was accquired using a DAQ with a sampling rate of 16384 samples/s.


The following plots are for the results that were obtained through this scheme of excess noise characterization. In a nut-shell all the properties of this excess noise could be verified(roughly) except for the exact shape of the noise spectrum, rather then the expected -10dB/decade the observed slope was closer to -20dB/decade. This was the only inconsistancy observed and possible reasons are also mentioned in the end.

1.)The plot below is for the spectrum of the bridge differential voltage for a drive voltage of 5V_{p} and frequency of  1.5 kHz.

Diffrential bridge voltage


One can clearly see the mismatch component( the huge peak) and the noise spectrum is expected to be around it. The mismatch voltage (after amplification) recorded was \approx 20mV. Which can also be verified in the above output.

2.) The above output was bandpassed and demodulated using the SRS830 lockin amplifier. The demodulated output is shown below, both the in-phase and the quadrature component are shown.

Demodulated spectrum

The reason for the quadrature component not being completely flat is that the resistors used will have a small capacitance which at 1.5 kHz will also produce an additional phase mismatch. The lockin was set at a phase difference of 0^{\circ} with respect to the reference but the phase was not adjusted to make one of the quadratures completely zero. That hardly makes a difference as one can clearly see the two spectrums are almost an order apart which would mean their power spectra would be two orders apart and subtraction of which produce almost no noticable change. Also, subtracting the spectra at two arbitrary phases will also retrieve the excess noise upto a scale factor. The equation below explains this:

S_v(\omega,i_0,\Delta\theta_1)= G^2[S_v^0(\omega_0-\omega)+\frac{i_0^2}{2}S_R(\omega)\cos^2\Delta\theta_1]

S_v(\omega,i_0,\Delta\theta_2)= G^2[S_v^0(\omega_0-\omega)+\frac{i_0^2}{2}S_R(\omega)\cos^2\Delta\theta_2]

S_v(\omega,i_0,\Delta\theta_2)-S_v(\omega,i_0,\Delta\theta_1)= G^2[\frac{i_0^2}{2}S_R(\omega)(\cos^2\Delta\theta_2-\cos^2\Delta\theta_1)]

That is the spectrum of the excess noise S_R(\omega) can be determined within a scale factor of (\cos^2\Delta\theta_2-\cos^2\Delta\theta_1).

One can clearly see that this is not a \frac{1}{\sqrt f} but the roll off is much steeper and almost double of what is expected. But as the consequent plots will demonstrate it does seem to follow all the other features expected from resistor excess noise. The background noise is limited by the amplifier and this can be verified with numbers as well. 

Confirmation that this is indeed excess noise

 The way this was confirmed to excess noise is rather unconventional, but I suppose it is logically sound. The idea is as follows- let us begin by assuming that there exsists no excess noise, then the output spectrum of the bridge voltage would be a pure sine wave(coming from the mismatch) , that is, the spectrum on demodulating the bridge output(which is pure sine wave according to our current assumption) would then be equivalent to that of demodulating a sine wave from the source itself of same magnitude(as both are pure sine waves with no noise around them). Now , if this is not the case and we see excess noise much greater than that when the source wave is directly demodulated then we can confirm that it is excess noise and this indicates that the noise around the frequency \omega_0 is something not coming from source but coming from resistors themselves. 

Consider the plot below which is obtained by demodulating the sine wave from the source itself, now if there was no excess noise this would be the exact spectrum expected on demodulating the bridge output as well, as without the excess noise its just a pure mismatch sine wave. But we see a lot of excess noise, indicating that noise around the sine wave is not the same as that of a pure sine wave produced by the source. Hecne we can then conclude that it must come from the resistors.

The above plot indicates that what were are seeing is not something that is expected of a pure mismatch wave, as a pure mismatch sine should produce the blue spectrum. Which is not the case , hence we can conclude that this noise is indeed coming from the circuit. Even though the excess noise is not exactly as what is expected it certainly does seem to be 'excess' noise.

Characterstics of the detected excess noise:

This excess noise was detected as extensively as possible in this process its response to drive voltages, drive frequency, type of resistor(Metal Film or Carbon Film) and resistor value all of these aspects were studied. All the results in for this were a decent match to the results in already documented literature.


The following plot is the dependence of the detected excess noise on drive voltage 1.5 k\Omega5\% Carbon Film Resistors, the drive voltage was varied from 0 to 5V (peak amplitude). 

A assuring fact is that the thermal noise level agrees very well with what is calculated.


where 4k_bTR_0 is the usual Johson thermal noise which is 4nV/\sqrt Hz  coming from the resistor and from the datasheet of the amplifier AD620 it can be seen that the the input referred noise of this amplifier is 9nV/\sqrt Hz at 1 kHz. So the predicted total background noise would be \sqrt{16+81}nV/\sqrt{Hz}\approx 10nV/\sqrt{Hz}. The following plot has not been caliberated, but the calculations of thermal noise do match very well.

The following is a similar plot for driving frequency of 500Hz:

 The following plot relates the noise amplitude at 500 Hz as a drive voltage this has then fitted linearly.The slope gives us information about the parameter \alpha

The straight line fit yields a slope of 2.54 \times 10^{-3} \frac{arb.units}{V\sqrt{Hz}} 

We can infer from the above data that a  linear feature is being seen in the power spectrum on drive voltage. This linear behaviour has been previously also noted in [1],[8]

Dependence on excess noise on resistor type

After verifying the linear behaviour I went on to test my setup to see the dependence on material type. I then hand matched 1.5 k\Omega1\% Metal Film resistors. Metal Film resistors have been shown to have roughly two orders of noise lower than the carbon film ones and this was observed in our data as well.

The plot below is for the excess noise spectrum for 1.5 k\Omega1\% Metal film resistors. The interferences noises were too high at frequencies above 20 Hz , hence the lock-in filter cutoff was kept low, the sharp roll off that is being seen is a result of that.

MF resistors are roughly 2 orders of magnitude less noisier than Carbon Films. This is much more evident in the plot below where a comparison at identical drive conditions(5V,1kHz) is shown. 


Another test

The next thing I did was to try and observe the dependence of this excess noise on the resistor value. The dependence of the bridge noise voltage of resistor can be seen from the very first equation:

\Delta V_{12}=\frac{\delta R_4-\delta R_3+\delta R_1-\delta R_2}{2R_0}V_d

Which I have quoted above. There is a \frac{1}{R_0} depedence on nominal resistance so ,if I were to measure excess noise in 200 \Omega resistors then I should see it to be roughly an order of magnitude more noisier than the 1.5 k\Omega resistors. This is exactly what was seen. I hand matched 200 \Omega metal film resistors and then repeated the same form of analysis. The results are in agreement with what we would expect owing to the above equation.

1) The following plot is for excess noise at various drive voltages for a drive frequency of 1 kHz

Excess noise in 200 ohm CF resistors

The following plot is for comparing excess noise in 1.5 k\Omega and 200 \Omega Metal Film Resistors, the 1.5 k\Omega  has roughly an order of magnitude less noise than 200 \Omega resistors.

Excess noise in 1.5 kohm and 200 ohm MF resistors

The following plot is for comparing excess noise in three resistor types tested.

 Excess noise dependence as a function of frequency

The dependence of this excess noise was also studied as a function of changing frequency. One would expect that this should not change with frequency, but these resistors have their own capacitances and that as preeviously mentioned could cause additional effects especially for increasing frequency. This is readily observed in the plot below where excess noise in 1.5 k\Omega CF resistors is shown. As frequency increases the effect due to this capacitance will become more dominant leading to a net phase change in the bridge output voltage. The plots below are for the in-phase component of the demodulated signal and as the phase changes with increasing frequency, the total magnitude of in-phase component is expected to come down.

Troubleshooting Attempts:

The fact that the shape of the spectrum is not what is expected of excess noise in reisistors was wierd, but the fact it did follow(roughly) other expected trends meant that the circuit was(almost) correct but there was something in the circuit that gave this spectrum instead of the 1/sqrt(f) . So I decided to troubleshoot using a few ideas which occured to me. What follows are the results that came out of these ideas. 

Digital Domain Processing of Signal

One possible thought that struck me was analyzing  the time series of the bridge output and demodulating it in MATLAB rather then using a lock-in this was to see if anything wierd is going on in the lock-in itself. The drive and the bridge output were recorded simultaneously and demodulated on MATLAB and the resultant spectrum was then studied. The result of this analysis was the same as when lock-in was used implying that there was nothing wierd going on in the lock-in and that whatever was happening was happening in my circuit( consisting of a amplifier and the resistor bridge).

These plots show the spectrum obtained directly from the time series of differential bridge voltage and drive.     

Digitally Obtained spectrum of the drive, drive freq=1kHz
Digitally Obtained Spectrum of the differential bridge voltage










The drive seen(left) was the used to demodulate(i.e. direct multiplication) the deifferential bridge voltage(right), the striking result is that the spectrum obtained also has the same slope as that which was observed earlier! 

I suppose this conclusively proves that there is nothing wrong or wierd going on in the lock-in and that the problem can be narrowed down to being in the amplifier or the bridge setup itself. Any problem in the bridge setup is highly unlikely(after all it just consists of 4 reisistors). The only conclusion one can reach is that the problem must lie in the way the amplifier is amplifying the signal! Probably other amplifiers could be tried(AD8221 or INA103?) and tested to see if we get the same kind of results.

Another, reason could be that, we havent shielded our circuit,at all! Most of authors in the literature have taken immense care to very carefully shield their circuits both thermally and electrically(see [3],[4],[5],[8],[10]), some have even gone to the lengths of using PID controllers to control the temperature of their setup(see [10]) or some have also used  \mu-Metals to shield it from electromagnetic interferences(see [3]), but at the least they have used a simple aluminium box to electromagnetically shield it. This could be one possible reason for such a behaviour, also I had to long BNC cables and these could have large amounts of noise, an alternative could be to used low noise miniature BNC cables for this purpose. Particularly, in [12] a dedicated setup was custom made for this. 

My shielding attempts on the other hand have been non-exsistent! I think this is could be one of the major reasons that we could not observe the correct spectra. But, fortunately we are taking all of the necessary precautions for the phototdetector setup, and I am confident that this method of measurement should give us some good results. 


1) Frank Siefert, Resistor Current Noise Measurements, LIGO-T0900200-v1

2) Eric Quintero, Eric Gustafson, Rana Adhikari-Experiment to Investigate Crackling Noise in Maraging Steel Blade Springs, LIGO-T1300465-v2

3) Arindam Ghosh, Swastik Kar, Aveek Bid, A.K.Rayachaudhuri- A Set-up for Measurement of Low Frequency Conductance Fluctuations Using Digital Signal Processing Techniques, axXiv:cond-mat/0402130v1

4) S.Demolder, A.Van Calster, M.Vandendriessche- Current Noise in Thick and Thin Film Resistors

5) S.Demolder, M.Vandendriessche, A.Van Calster- The Measuring of 1/f Noise of Thick and Think Film Resistors

6) Edoardo Milotti- 1/f Noise: A Pedagogical Review

7) F.N. Hooge - 1/f Noise Sources- IEEE Transactions on Electronic Devices, Vol.41, No.11 November 1994

8) John H.Scofield- AC Method for Measuring Low-Frequency Resistance Fluctuation Spectra- AIP, Review of Scientific Instruments.58(6), June 1987

9) J.Heefner- Resistance noise measurement Summary, LIGO-T070019-01-C

10)Patrick Barry and Steven Errede, Measurement of 1/f Noise in Carbon Composition and Thick Film Resistors, Senior thesis- University of Illinois at Urbana-Champaign, Fall 2014

11) Peter M. Marchetto-1/f and Johnson-Nyquist Noise in metal-film and carbon resistors

12) Walter C. Pflanzl, Ehrenfried Seebacher- 1/f Noise Temperature Behaviour of Poly Resistors , 19th International Conference "Mixed Design of Integrated Circuits and Systems", May 24-26, 2012.



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  1938   Thu Jun 18 22:24:31 2015 ranaElectronicsPD noiseResistor noise observations,analysis and inference

This is possibly nice, but please replace the EPS with PDF and make all of the fonts (including superscripts) larger than the text font. Then we'll be able to read this stuff.


  1941   Sat Jun 20 00:47:40 2015 Arjun MiscPD noiseSetting up of mechanical parts for PD noise measurement

I have started building the PD noise measurement setup today and was able to complete all the mechanical parts. The intended diagram as well as a picture of my setup can be found. I am not sure about how well they are oriented with respect to each other, but I was able to adjust the photodetector mounts with respect to beam splitter mount decently well enough using a laser pointer. Also, the postioning of beam dumps are also an estimate. On the bright side, all the mechanical parts that were made in the shop are perfect, and the metal box fits perfectly on top of the setup.On Monday we should be able to set this up completely in 058D and install all the optics as well.

Basic measurement topology


PD noise Schematic
  1942   Mon Jun 22 18:22:26 2015 ArjunMiscPD noiseCross-Correlation scheme to calculate PD noise

This eLog is a for a new cross correlation method I have been exploring and will be soon trying it on resistors, this log presents some simulations I was able to do and also a schematic for such a measurement.

Cross-Correlation Technique

The Technique itself is extremely simple but it seems to be a very promising technique and I hope to implement this in the PD noise measurement setup. It makes use of two identical set of electronics through which a common signal is passed, and  the final output of the two is cross correlated to obtain the spectrum of the signal where as the uncorrelated background setup noise averages out to zero. Using this method one can go as far 2 orders(roughly) of magnitude below the background noise arising from electronics (readout components, amplifiers etc). The math behind it is extremely simple yet elegant-


Where n_{tot}(t) refers to the total noise as measured at the output, n_{sig}(t) refers to the signal whose spectrum we wish to assess and n_{back}(t) refers to the background noise arising solely out of the setup.

Consider two identical setups then we can write-



The signal through the two identical systems is the same where as the background noise in the system and hence the total noise changes.

We then calculate the cross correlation of n_{tot,1} and n_{tot,2} . Its very easy to see that the terms R_{n_{b1},n_{sig}}(R_{xy} is the cross correlation of x and y signals) R_{n_{b2}n_{sig}}R_{n_{b2}n_{b1}} will all be zero as the neither of the noises are correlted with any other noise. Hence the cross correlation of the two signals is actually nothing but the auto correlation of the required signal. The fourier transform will then give the required power spectrum. There will be fluctuation in this estimated power spectrum but by an ensemble average over many realisations one can converge to the ideal power spectra. 

Simulations Performed:

Consider a simple \frac{1}{f} spectrum masked by background noise, through these simulation I will demonstrate how one can extract this noise and go below the background noise upto almost an order of magnitude by avering over sufficient realisations.

1) The Simulated \frac{1}{f} noise is shown below along with the background noise .

 2.) We then use the function cross power spectral density function in MATLAB to estimate the power spectrum of the desired signal. The following plots are estimations for varying number of ensemble averages.

Notice how the fuzzy blue parts below the bcakground noise get smaller and converge to the red trace- which is the actual power spectum and it can be seen that the background noise being at 10^{-2} and after 1000 ensemble averages a decently good \frac{1}{f} spectrum can be seen till 10^{-3} ie one can dig upto one order of magnitude below the electronics noise which is generally the limiting factor. This seems to be a very promising method and can be applied to PD noise as well. This will be implemented on the resistor setup.

Scheme to use it in the PD noise Experiment:

I came up with the following scheme to implement the above cross correlation method.

Attachment 1: both_spec.pdf
Attachment 2: 10avgs.pdf
Attachment 3: 50avgs.pdf
Attachment 4: 100-1avgs.pdf
Attachment 5: 100-1avgs.pdf
Attachment 6: 500-1avgs.pdf
Attachment 7: 1000-1avgs.pdf
Attachment 8: 10avgs.fig
Attachment 9: 50avgs.fig
Attachment 10: 100-1avgs.fig
Attachment 11: 500-1avgs.fig
Attachment 12: 1000-1avgs.fig
Attachment 13: New-Project2_(2).pdf
  1946   Thu Jun 25 15:36:54 2015 ArjunMiscPD noiseUpdates on PD noise setup

Over the past few days we have dismantled the laser gyro setup and started setting up the PD noise measurement setup. We have succesfully mode matched our laser to the cavity and then locked the cavity to our laser by using a 'Homebrew' circuit for implementing the a loose realisation based on the PHD technique.  A brief description is presented on how the mode matching was done and also the circuit which we used to lock the cavity to our laser.

Mode-Matching the Laser to the Cavity:

We used a combination of 2 lenses and two mirrors to mode match the laser and the cavity. The mirrors were used to steer the beam and get rid of alignment mismatch and the positions of lenses was adjsuted to compensate for curvature mismatch.

1) We first placed the lenses and mirrors at approximately at positions where Zach had them for his gryo setup and used a CCD camera to see the shape of the output mode and that verify that we had locked to correct mode, and tried aligning the beam better using the mirrors to see the best we could do. If the best match we achieved was less than 80%(roughly) then we moved on to aligning the lenses. 

2) We iterated the process of finding the maximum transmission by first adjusting the lenses and then adjusting the two mirrors, then finding the maximum mode matching we could achieve. We used a Thor Labs PD to study the output transimission, in conjugation with a CCD camera.

3) One could also study the reflected beam and minimise that to its lowest. After iterating for couple of hours we were finally able to mode match the cavity to the laser with transmission of ~85%.

Mode-Locking the cavity to laser:

 For locking the cavity to the laser we used a 'Homebrewed' locking circut loosely based on the PDH technique, but we have not modulated the laser's phase(this will be our next task- stabilising the laser intensity fluctuations by using a AOM). A dither of 1MHz and amplitude 24dBm was given using a DS345 function generator. This was passed through a power splitter, with half the power going to the RF input of the bias tee and the other half to the LO input of the demodulating mixer after passing it through a 7dB attenuator, this attenuator is necessary as the input power of the mini circuits mixture must have a specific amplitude. The output of the power meter was passed through a DC blocking cicuit as the RF input for the mixer. The IF output of the mixer is then connected to the SRS560-low noise preamplifier. The SRS560 has a programmable input filter which was set to a one pole(roll off of -6dB/octave) low pass filter with a cutoff frequency of 10Hz. This filtered signal was then amplified by a gain of 5000 and this output was the DC input to the Bias Tee. A simple schematic of circuit we used is shown below:

A more detailed schematic is given below with the exact components which  were used:


The image below is a picture of our optical setup and the red line through the image is the laser path.

 The following is a image of the output as the amplifer(SRS560) is turned on and the the output of the PMC gets locked. The two traces are the amplifed error signal and the output of the PMC, as seen on a oscillosope:

PS: Also find attached a image of the settings on the DS345 and SRS560. 

Attachment 1: Cavity-locking-circuit.pdf
Attachment 2: Cavity-locking-circuit_(1).pdf
Attachment 5: IMG_20150624_175321.jpg
Attachment 6: IMG_20150624_175342.jpg
Attachment 7: IMG_20150624_175533.jpg
Attachment 8: IMG_20150624_175234.jpg
  1947   Thu Jun 25 18:31:11 2015 ArjunMiscPD noiseResistor noise measurements using the cross correlation method

I have taken a few sets of measurements using the cross correlation method which I have described in log 1942. In this log I briefly describe my circuit and the results I obtained.


The image below is a snap of my circuit. The circuit is very simple- It has two regulators(7812 and 7912) for +ve and -ve voltage regulation respectively. The instrumentation amplifier used is the standard AD620 with a gain of 100 which corresponds to a gain resistance of 499\Omega. I soldered two identical sets of amplifiers, the voltage regulator circuit and a then mounted my 0.05% resistor bridge setup onto it using berg connectors. I took extra care of ensuring small details like- proper soldering, using twisted pair of cables and using small wires for all connections.

 I then excited the bridge which consists of 1.5k\Omega Metal Film Resistors using the lock-in sine output of peak amplitude 5V. The two output time series were directly fed to the DAQ and stored for analysis. The following analysis is on a 2 hour long time series. Since the total accuracy of the method increases with total measurement time(either by ensemble averging or by recording data for extending periods of time). The following plot is of the spectrum obtained at the two amplifier output

As seen above the amplifiers produce almost identical especially at the drive frequency of 1 kHz suggesting that the amplifiers are behaving identically. We then demodulate this signal on MATLAB using the function amdemod with a sine of 1000 Hz.This fuction comes with a lowpass butterworth filter ,making it very convinient for demodulating signals on MATLAB. The resultant spectrum after demodulation for one of the amplifier is shown below.

The output time series was decimated first( so as to get higher resolution at lower frequencies) and then the cross correlated spectrum was calculated, it was curve fit with a straight line. The corresponding results are shown below:

The numbers in the brackets are the 95% confidence bonds on the slope. 


The above plots still estimate the slope of excess noise spectrum closer to -2 than to -1. Previous estimates are in good agreement with this observation. But, the thing that is bothering me is the reason for this deviation from a -10dB/decade roll off to a -20dB/decade for the amplitude power spectrum which is disagreement with previous studies on this topic. My fear partly is that the error in I am making in the resistor measurements should not carry onto the PD noise setup leading to incorrect results there as well. As stated in a previous log, my suspicion boils down to two places, because, they are only parts I have not played around with. First being a different amplifier, AD620 is the standard instrumentation amplifier and should be the last place to suspect a flaw but it is possible that maybe( however the remote the chance maybe) their could be something wierd in AD620 making it suitable for such noise measurement applications especially when using the heterodyne demodulation technique. It would be interesting to see the results by replacing different amplifiers in place of AD620. Secondly, improper shielding/ external influence. This is more likely I suppose as my shielding attempts were non-exsistent some external source could be causing this deviation. Or another third possible place equally likely could be a flaw in my analysis or my scheme, even though I have done my best to implement the techniques from various sources, as accurately as possible, it is very likely that I may be negligent to some error I am commiting. Any suggestions on the possible sources of error are most welcome.


Attachment 2: IMG_20150624_200105.jpg
Attachment 3: 6.pdf
Attachment 4: 1.pdf
Attachment 5: 7.pdf
Attachment 6: 4.pdf
Attachment 7: 5.pdf
  1953   Mon Jun 29 16:59:47 2015 ArjunMiscPD noisePre-stabilisation of laser

We have managed to pre stabilise the laser using a few SR560's. It is not as stable as the one that would be implemented once the digital system is in place. But it should be good enough for some preliminary data. As described in a previous log we had locked the cavity to track the laser. The PZT actuator in the cavity is driven by a SR560 which has a limited output voltage range from (-4V to +4V) and if due to slow frequency drifts the frequency of the laser drifts beyond the limit to which the PZT can compensate the cavity would unlock itself. Also, currently the intensity noise of laser has not been stabilised, this passes directly through the cavity and will appear at the transmission if its not accounted for. This feedback has been achieved using a AOM. The AOM is driven by a RF function generator and the output power of thr RF function generator can be modulated by this in turn changes the power in the carrier frequency by pushing some power into the first order diffracted beam, thereby stabilising the laser intensity fluctuation by almost 2 orders of magnitude. A brief description of the feed-back loops is given below.

Frequency Stabilisation

The laser was frequency stabilised for its slow drifts, this could lead to SR650 controlling the cavity not being able to compesnsate for it due to its limited output range. The output of cavity stabilisation was low passed and then fed to the frequency control of the laser. But the gain had to be adjusted appropriately, as otherwise the loop could become unstable. This was done by using a simple resistive divider circuit (potentiometer) to first attenuate the control signal for the cavity stabilisation and then low passed and fed back to counter for the slow frequency drifts of the laser.The cutoff freqeucncy of the secondary feedback loop is also important, so as to ensure that at cross-over frequency nyquist stability condition is satisfied. The cutoff for this was kept at 30mHz.

Intensity stabilisation 

Intensity fluctuations from the laser will appear at the output port of the PMC because of the fact that the cavity is locking itself to laser's frequency. These have to be corrected for in the final setup and this has been achieved using an AOM. A AOM splits power in the beam into a diffracted beam and this splitting of power depends on the power that is injected into the AOM and also the direction in whihc the AOM is oriented. When no power is injected in the AOM the carrier beam passes through unaffected and when some non-zero power is injected a part of power goes into another diffracted beam. The AOM we use has a maximum power input of 2W at 80MHz. For  achieving the required functionality, a RF signal from a RF signal generator is amplified using a high power RF amplifier and this drives the AOM. Now to stabilise the intensity fluctuations going into the PMC we can setup a positive feedback by sensing the power at thePMC's output and using that to modulate the signal generated by RF signal generator thereby modulating the power with which the AOM is driven and finally controlling the way power is split between the two output rays. Hence this way the power entering the PMC has been stabilised. In the actual setup this feedback will be provided by the common mode output of the two photodiodes being tested.The entire pre-stabilisation of laser implemented is shown in the schematic below.

The total stabilisation schematic is shown below-this includes the locking of the cavity, feedback to supress the laser frequency drift and supression of intensity fluctuations.

We were also able to get some more optics fixed, since the table will also be used for another experiment, we divided the beam using a halfwave plate and a polarising beam splitter to control the power going into each of the experiments. A image of the setup assembled is attached below, the read trace is the path of the laser.

Attachment 1: Screenshot_2015-06-29-16-20-55.png
Attachment 6: 20150626_202413.jpg
  1957   Wed Jul 1 17:31:16 2015 ArjunMiscPD noiseUpdates on PD noise setup

In the last few days we were able to get almost all the optics set and aligned. The following few images show the complete setup with all optics aligned and a image showing the laser beam entering the box.

The experimental setup with all the mirrors and optics aligned
The experimental setup outside the box












The laser beam entering the box

The next thing to do would be to setup the photodiodes and the readout electronics. Since the digital system the lab is not up yet we decided to hardwire the required filters on the LIGO- generic filter board, becasue using SRS560's was making the space too clumsy.

Attachment 3: IMG_20150630_192049.jpg
  1963   Wed Jul 8 18:32:51 2015 ArjunMiscPD noisePD noise update

The DAQ system in the ATF lab has not been yet setup completely and as mentioned in the previous eLog, we decided to go ahead and build those circuits by hand, as labaorious as they were we were finally able to get a few servo designed, simulated, characterised and running. In this eLog, I describe the servos we built. 


Cavity-Locking Servo

The first servo was the cavity locking servo, as mentioned in a previous post that the SR560 used has very very low output rails(-4V to +4V) and hence can hardly keep the cavity locked in response to the laser drifting for a few minutes. We implemented this as a filter on a solder board, with rails of -15V to +15V but this wasnt enough, to hold the cavity locked for more than 10mins. We needed some very high voltages! So we put to use the piezo driver in the ATF lab.

Our initial servo was a simple one pole active RC filter with a cutoff of ~10Hz and a DC gain of 100. This worked and kept the cavity locked, but it unlocked itself after a 10mins or so. Now, when we implemented the piezo driver, we could keep the cavity locked for much much longer times(~40 mins) but it was only marginally stable and showed some features of instability. This was because the piezo driver itself has a low pass characterstic with a pole of ~8Hz and a gain of 20 and this was making the feedback loop unstable, because now we have {\frac{1}{f^2}} slope after 10Hz and at around unity loop gain frequency ~300Hz the phase margin was very poor ( probably 10's of degrees) this made the loop unstable. 

Solution was simple- add a zero and push up the phase margin! This was done with a zero at around 500Hz and an additional flat gain stage was put in with a gain of 30, this was to further increase the UGF and push it beyond ~3-5 Khz. This was first simulated and the built and tested.

The schematic of the setup is given below along with the components values(note: I have simply modeled the piezo driver's response as non-inverting low pass filter,this does not refer to the PZT actuator's response but just the high voltage piezo driver's response)

The following images are comparison of TF simulated on LISO and measured TF for each of the stages and the combined TF as well. But the TF with the piezo could not be taken yet, so just the LISO result is shown. Some comments on them:-



The measured response is almost a perfect match for the simulated LISO response with the gain differing by 0.02dB.  So the flat gain is working as expected with a wide bandwidth of about ~20kHz.

2) The following is the TF with the second boost stage included:-

 The TF is as expected. With the zero almost exactly at 500Hz. 

3) The full cavity servo simulated TF:-

The features are exactly as predicted, a steep slope from the two poles and a flattening effect by the zero after 500Hz, this combined with a flat gain stage pushes the UGF to almost 7kHz, which is more than sufficient for our purpose and the phase margin has also drastically improved. Also find images of the attached circuit.


 Instensity Control Servo:

The next step was to analyze the and design the intensity servo. For this first the free running laser noise was measured this was around \sim 10^{-4}Vrms/\sqrt{Hz}, which is very bad. We wanted to reduce this common mode noise to shot noise limit, which would require almost 5 orders of supression. Also we would want to setup a stable offset as if its forced to zer we wont have any power from laser at all. We are in the process of designing this and we should be able achieve this in a few days. 

Attachment 1: cavity-servo.pdf
Attachment 2: cav_TF_stage1.pdf
Attachment 3: cav_TF_stage1.pdf
Attachment 4: cav_TF_stage2.pdf
Attachment 5: cav_TF_revised.pdf
Attachment 6: cavity-servo_(1).pdf
  1968   Fri Jul 10 00:59:49 2015 ArjunMiscPD noisePD-noise update

Today we were able to get the servo running which we described in the previous eLog. The servo locked beautifully and the instabilities other wise observed without the zero we specifically added was gone. There was small change that had to be made, the servo had the wrong sign which we very took care of. So, just for the sake of completeness, I am attaching the TF of the inverted and the non- inverted servo. Also, I am attaching the corrected schematic. I coudn't take a snap of how the lock was before and after we introduced the our servo( I will make sure its in the next log).

A image of the servo we built






Addititonally, we assembled koji style beam dumps onto our setup, but the alignment was such that we couldn't place the third beam dump for the reflection coming from the bea splitter. So we may have to realign the beam splitter and then re-align the PDs as well. Its not as tedious as it sounds and we should be done with it tomorrow.


The only thing that would now remain would be to design and implement the servo for intensity suppression, what makes it ore difficult is that we need to include a very quiet reference in the circuit and we are brainstroming over it. Also, find attached a image of the actual servo we built. Also, I have attached the LISO simulation code.

Attachment 1: cav_inv.fig
Attachment 2: cavity_servo.fil
#---------------Frequency tracking control servo---------------

#stage-1- non-inverting gain stage,G=30
r r1 1k n1 gnd
r r2 32.8k n1 n2

op u1 lt1128 nin n1 n2

#stage-2, Low pass filter pole=10Hz, zero=500Hz
r r4 2k n2 n3
... 29 more lines ...
Attachment 3: Screenshot_2015-07-10-00-41-15.png
Attachment 4: IMG_20150708_180717.jpg
Attachment 5: Screenshot_2015-07-10-00-51-01.png
Attachment 6: cavity-servo_(2).pdf
  1969   Sun Jul 12 02:00:15 2015 ArjunMiscPD noisePD noise Update

Today, in the process of designing a intensity control servo I took some noise readings of the PDs and also characterized how the modulation function in the Marconu FG works, as it would this would be essential in the process of desigining a efficient feedback loop. 

Firstly, the alignment of PDs was off, so I had to correct them, which took me more time then it should have, but I finally got it done. Next, I moved on to taking noise readings to measure the 'free running' laser noise, but as it was pointed out to me, this plot in itself is meaningless until the power at which it was taken is specified or in other words the quantity of use would be \frac{\delta P}{P_0} which is also called the Relative Intensity Noise(RIN). What I measured was the Voltage noise in the PDs for a fixed Bias voltage( set using the power control of the laser). I calculated them for different bias voltages and the plot is shown below, the wierd shape is because I tried to splice three dfferent spans and it did ot combine as smoothly as I expected. I took noise at 3 different spans(100Hz, 1kHz and 50kHz) and combined them using splice.m program written by Koji. The change in the voltage noise with the bias voltage can be seen very evidently . We could then use the voltage noise at 1V as a measure of RIN(\frac{\delta V}{V_0}\vert_{V_0=1V}) .  Again the plot looks wierd becasue of the splice function I used, I will post a better plot when I take another set of readings, in my next log. This plot tells us approximately how much of RIN is there and how much suppression would be needed in our servo. Additionally, I also measured the dark noise but they really wierd after splicing, so I will post those as well in my next log. 

Analyzing the AOM and the Marconi FG

The marconi RF function generator I am using has a modulation input which can be used to control the power going into the AOM which would inturn control the power in the main beam, this is my plan in implementing the intensity feedback. So, I studied the AOM and how it responds to imput modulations of different kinds, this is what I learnt:-

  1. If the input modulation has no power in it ( or that its amplitude is 0) the power in the out beam is unchanged. That is zero power at modulation port corresponds to no change in the power.
  2. If the Input voltage as a -ve value( which I set by using the offset function in the function generator generating the modulated signal) then, the power output of the FG decreases.
  3. If the voltage is +ve, the power increases.
  4. There are some other constraints one woud have to consider as well. The max power that can go into the AOM( Model:Gooch & Housego R23080-2W) is 2W( which is 33dBm). A power RF ampifier( mini circuits-ZHL-1-2W) is used to amplify the signal from the FG has a gain of 33dB, which I looked up from the data sheet, so the max power of the  FG must be around 0dBm. But just to be safe, today I just explored its features with a input of -2dBm. I have attached a few photos of me toggling the carrier on/off switch and how it modultes the transmitted power, one coud then send a modulation at a fixed frequency of say 1kHz- this would amplitude modulate the power, which is exactly what we want for our excess noise detection scheme. I have attached a image of this as well. But this whole setup of FG+AOM has to be characterised properly- which is what my next task at hand is.


Attachment 1: fig1.pdf
Attachment 2: IMG_20150711_161731.jpg
Attachment 3: IMG_20150711_161741.jpg
Attachment 4: IMG_20150711_161726.jpg
Attachment 5: IMG_20150711_204122.jpg
  1970   Mon Jul 13 16:47:46 2015 ArjunMiscPD noiseNoise Plots

The plot in the previous eLog was all mixed up, hence I am attaching another plot for the relative voltage noise observed in the photodiodes.I increased the number spans of measurement from 3(100Hz, 1kHz and 50kHz) to 6(100Hz,1kHz,3kHz,10kHz,25kHz and 50kHz). The plot this time is much cleaner then the jaggered one I got yesterday.

First, the relative voltage noise plots for the 2 PDs, the very noise at low frequency(~500mHz) is becuase the spectrum analyzer was set in the DC coupled mode and the Pds were biased at 1V, thats what is being seen there.

noise suppression

I have been thinking on the intensity noise suppression, suggestions from Zach,Rana and Gabrielle were extremely helpful in this regard. What I have in mind is a AC coupled feedback loop, AC coupled because if we have a high Gain at DC, we will have to prodive an additional stable reference to the circuit so the the DC power coming from the laser is not suppressed. Additionally, if we went with DC coupled feedback, we would have to change the reference everytime we change the power going into photodiodes and making a tuneable reference is easy. Rather we could ignore the DC drift of the laser. Our region of interest would be to know the excess noise in photodiodes beyond 10Hz and upto a 1000Hz. So we could provide very high gain in this region and supress these fluctuations in the laser. This is precisely what did today.I used the outputs of one of the PDs and amplified it using 2 SR560's and fed this back to the modulation port of Marconi FG which drove these to supress the intensity flcutuation in the other PD. The schematic is shown is shown below:- 

As mentioned before, I assumed our region of interest to be 1Hz to 1000Hz and owing to the fact that its AC-coupled feedback, I used to bandpassed filter configurations in SR560's. In the first one I used a bandpass with cutoffs 0.3Hz and 1KHz with a roll-off of -6dB/octave and a gain of 20, anymore gain overloads this amplifier. This was cascaded with another SR560, with a bandpass configuration of 1Hz to 100Hz. The gain in this was incresed slowly in steps from 100 to 1000.The total gain in the region of interest being 10,000 Anymore gain and this SR560 gets overloaded. It is very importnant to keep an eye out for large fluctuations, as these immidiately overload the amplifiers, making the whole loop crazy and one has to reduce the gain first and then re-stablize it. Also the power output of the Marconi FG must be kept low as a large amplified feedback signal could harm the AOM which has a input power limit of 2W. In the end I was able to achieve around 2 orders of supression! See associated images below for the configuration that seemed to be stable for quite some time(~1hr and counting).


  The final suppressed noise spectrum is presented below.

My plan next is to simulate and design this loop more efficiently, probably by adding a few boost stages. With -15V to 15V rails in opamps I should be able to give this feedback loop even more gain and maybe supress the fluctuations by another order or so. Once I simulate the circuit I can probably start putting it together, also in my cicuit I will have to give a port for adding the modultaion, this could be done using a summing amplifier. 

Attachment 1: RIN.pdf
Attachment 3: IMG_20150712_154445.jpg
Attachment 4: IMG_20150712_154439.jpg
  1972   Tue Jul 14 20:55:50 2015 ArjunMiscPD noiseSome thoughts on the intensity stabilization servo

The previous eLog I had described, the way I tried to supress intensity noise is a adhoc method. So after a discussion with Rana, I realised the parameters that needed to be considered while designing the sero and it was also pointed that my filter itself was unstable, which I overlooked yesterday.

Things to keep in mind while designing the servo:-

1) Why do we need intensity a servo if finally we calculate the diffrential signal, won't it get rid of the laser noise, as its common mode? What dictates the subtraction efficiency one can have is such a subtraction?

2) What would this efficiency be in the 'ideal' and 'non-ideal' scenarios?

3) This subtraction efficiency would inturn dictate how much gain one needs for the servo.

A simple filter design with a modest gain was suggested by Rana, I am convincing myself how it will help get better subtraction efficiencies. 

  1974   Wed Jul 15 13:40:38 2015 ArjunMiscPD noiseIntensity stabilization servo

I was able to deduce the answers for a few questions I was working on. The factor that will limit how well we ca subtract two signals is limited by the amplifier we use and it Common Mode Reduction Ratio(CMRR). Some calculations to support my point:

Output of any amplifier is given as : V_o=A_d(V_+-V_-)+\frac{1}{2}A_{cm}(V_++V_-), where A_d is the differential gain and A_{cm} is the common mode gain. CMRR is defined as 20\log(\frac{A_d}{A_{cm}}) in dB.

Consider a generic differential amplifier(like AD620) it has a CMRR of 100dB(10^5) for a gain of 100. The common mode noise is at \sim 10^{-5}V/\sqrt{Hz} and the noise we wish to detect is at \sim 10^{-8}V/\sqrt{Hz} ie around the shot noise limit. Lets say the gain of this differential amplifier is 100, then that would mean that the gain associated with the common mode noise would be 10^{-3} as CMRR is 100dB. So the final output of the amplifier would be \sim10^{-3}\times 10^{-5} + 100\times 10^{-8} that is the common mode would be 1% of the total output and this is not very good, by doing a intensity suppression we can improve our ability to subtract by another order or two depending on our servo design. This is the idea behind doing a intensity supression, so that at  our output we have negligible common mode component, which comes from our readout amplifier which has finite CMRR.


  1975   Wed Jul 15 15:45:04 2015 ArjunMiscPD noiseIntensity Servo design

Rana came up with the following intensity suppression servo- A simple first order bandpass filter to supress noise in the band witdth of interest from ~1Hz to 3kHz. The pole and zero are at the same frequency of 300Hz and a gain of 30 at 300Hz. Following are some simulations as to how the closed loop transfer will look. This inturn is also a test for the stability of the closed loop. Additionally, I also measured the TF of the remaining loop( ie the MArconi FG, AOM and photodiode put together) Take a look at the attached images

1) TF of remaining loop(AOM plus Function generator).

2) TF of the designed servo.

3)Magnitude and phase response of the open loop gain.

4) TF of the closed loop will require the responsivity of the photodiode, I am not aware of this value for the photdiode I am using, as soon as find that value I will attach the closed loop reponse as well.

(The sudden phase jmp is because in LISO angles go from -180 to +180 degrees.)



Attachment 1: TF_intensity_servo_remain.png
Attachment 2: filter.pdf
Attachment 3: OLG.pdf
  1977   Wed Jul 15 22:14:19 2015 ArjunMiscPD noiseIntensity Servo design

Today evening I implemented the intensity servo, but I couldnt characterize it properly, which would be my first task tomorrow morning. Also, I have made a small circuit consiting of AD620 to analyze the differential signal and see the amount of common mode supression that I obtain. One thing to note is that, gain of the servo will have to be increased 30 to 300, because the remainig loop(AOM, MArconi FG etc) has a flat magnitude response of ~-20dB, to compensate for that we will have to increase the gain.

Also, another thing is to consider is that, there could be some differential signal due to the fact that the the transimpedence amplifiers for the two photodiodes are not exactly identical and this could cause some additional differential signal. I am currently thinking of a method to characterize and measure that difference.

  1980   Thu Jul 16 18:28:53 2015 ArjunMiscPD noisePD noise update

Today, I implemented the intensity servo and characterised it using the voltage injection method to calculate its loop gain. I also compared it with the simulations for the same that I performed on LISO. I have attached the figures below. I also completed the circuit I made to measure the differential signal of the PDs(its just a voltage regulator setup with a AD620 with a gain of 100), but I could not characterize or take measurements using that. I will do that tomorrow. 

1) RIN with and without feedback( both in-loop and out-of loop). The gain for this loop was set at 200, this was to compensate the -16.5dB loss in the remaining setup(AOM, RF function generator etc).

2) Open loop gain Transfer function- simulated and measured.

I am still stuck on how to measure the difference in the reponse of the two transimpedence amplifiers on the photodiode readout board, I have a few ideas, I am assessing their validity.

Attachment 1: Sim_vs_Obs_OLG.fig
Attachment 2: Sim_vs_Obs_OLG.pdf
Attachment 3: Noise_plots.fig
Attachment 4: Noise_plots.pdf
  1983   Fri Jul 17 18:12:36 2015 ranaMiscPD noisePD noise update

something seems bogus here...the suppression at 1 Hz is much more than the open loop gain. usually this is mathematically impossible...

  1984   Sat Jul 18 22:24:08 2015 ArjunMiscPD noisePD noise update

 I did not note this inconsistency. I will take new measurements. I am wondering what error on my part could have caused this to happen.


something seems bogus here...the suppression at 1 Hz is much more than the open loop gain. usually this is mathematically impossible...


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