The TF looks good. But the noise measurement is obviously limited by the SR785 noise. We need a preamp, which is only for the purpose of the measurement. It has to have the input reffered noise about a factor of a few better than the noise predicted by Zero. At high frequency, probably we will be able to use SR560. With this low noise level, probably we can just use the flat gain of 100 for the SR560 setting. This will give you the input referred noise (of the preamp) of  ~4nV/rtHz at kHz band. Note that the gain needs to be larger than 100 to have low noiseness of SR560.

I think this is a solid measurement.

The circuit has been soldered(refer entries 2399) and the noise for Sallen Key was analyzed

Attachment 1: Circuit Diagram of Sallen Key low pass filter( cut-off= 1Hz)

Attachment 2: Transfer Function of sallen key.  The Frequency response Measurement was done using the Swept Sine group. The input range was -50dBVpk.

Attachment 3: Noise comparison between zero and SR785 measurements. The noise matches the simulated results to a great extent and also it's less noisy so can successfully be used to bias the photodiodes(1V).

Attachment 4: Zip File

[Koji, Nathan, Duo, Shalika]

The dark noise of the Sb3513_A2 photodiode was observed using an SRS785 spectrum analyzer. Different bias voltages were provided to understand the dependency of dark noise on the bias voltage. It was observed that as the bias voltage deeps on decreasing the dark noise decrease too. A transimpedance amplifier with a gain of 5k was used to convert photocurrent into voltage. A Sallen key low pass filter was used in order to provide a low noise bias. 

** Since the gain of the TIA was 5k so the output voltage noise was divided by 5k in order to get dark current noise.

Attachment 1: Dark current noise across Sb3513_A2 500um

Attachment 2: Dark current noise across Sb3513_A2 750um

Attachment 3: Dark current noise across Sb3513_A2 1000um

Attachment 4: Dark Current across Sb3513_A2 500um, 750um, 1000um respectively

Attachment 5: zip file containing all data

[added (Nathan)] Attachment 6: Experiment setup.

Koji set up an experiment measuring the dark current of the photodiodes. A bias voltage is given and the current is converted to voltage via a TIA, where it is measured. Also note that in order to provide a high quality bias voltage, we LP the output of the device with a second order sallen key filter cutoff at 1Hz. 

DB9 switchable breakout box is ready. We are ready to do some PD test now.

Chris and I went to the lab and made some plans about how to use the space around the optics table. Attached a drawing of it. A couple notes about the drawing:

1. Green: underneath. The rough pump is under the table. The connection to the coldhead goes on the floor.

2. Rack: electronics rack.

3. Yellow cabinet: the cabinet that has chemicals in it.

4. Turbo/Rough: pumps.

I found only 1uF Tanatalum Capacitors in the EE shop. Maybe a a higher capacitance of tantalum capacitor will help. 

Another thing, is it good to compare the input referred noise for the different types of regulators? I did a comparison between the output noise only. 

InAsSb PDs were housed in the PD cages. The cages were engraved to indicate the batch (Sb3512 or 3513) and the serials (A1, A2, ...).

The PD legs does not have an indicator for the pin1. So, the tab of the PD case is directed "UP". Also the direction of the tab is marked on the cage. The tab of the short plug was also aligned to Pin1. However, the PD case is too thin and the PDs can rotate in the cases.
So the face photo was also taken so that it indicates how Pin 1 looks like from the PD face. (Attachment 4)

Also made the cable for the LaserComponents PD and the InAsSb PD. Pin n shows up as Pin n of DB9 Male connector.

Once we have the PD test is the bias circuit (with a monitor) and some patch panel kind of preparation, we can start working on the PD test.

The photodiode needs a 1V bias so for a clean bias we have decided to use a sallen key low pass filter with a cut off frequency at 1Hz. The quality factor of the designed sallen key filter is 0.707.

Attachment 1: The Circuit Diagram of Sallen Key filter

The gain of the circuit is 1. 

Attachment 2: The transfer function of the filter

We can see the cut off frequency at 1Hz

Attachment 3: The Input Referred Noise of filter

The input-referred voltage noise was obtained using SR785 and compared with zero simulation. It deviates a lot from the simulated results by a factor of 100.

Attachment 4: Scripts

Find all the data and scripts used for the measurements.

NOTE: Noise plot below had inconsistent units; please ignore.

After reproting phase noise issues with the VCO, here, I have changed the VCO for a high power RF amplifier - a mini circuits ZHL--5W==sma (ZHL-5W-1). It is being driven by thre Moku to provide the 80 MHz signal. Please note the following:

• This is a high power amplifier so you should follow the correct high power amplifier turn on/off procedure.
• The amplifier can output a maximum of 5W. However the AOM can only tolerate a maximum of 0.6 W. Ensure that an appropriately small signal is input into the amplifier, as to not break the AOM.

The new setup is shown in attachment 1. Attachment 2 show the phase noise of the LO arm. This is now a much more acceptable level, as compared to the previous VCO. I am outputting -19.2 dBm, and measuring 5.3 dBm on input 2. This means I am driving the AOM at 24.8 dBm, or 0.302 W.

With the greater drive on the AOM, as compared to previous incorrect useage of the VCO, the signal from the IFO arm is much more robust. Attachment 3 shows the phase time series of the IFO arm. You will notice the large jumps, which spoil the spectrum, not shown here. In attachment 4 I zoom into the largest of these jumps. You can see it is comprised of several linear transitions, which occur over a short time period.

Aidan has suggested that locking the laser is the best way to avoide these non-stationary noise sources. He suggests that these jumps could be mode hops of the laser.

I rebuilt one of our old desktop machines to serve as NFS server for the cymac. It is running Debian 10.0 and assigned IP 10.0.1.169 (hostname qil-nfs). I installed a new 2 TB hard drive dedicated to hosting the LIGO RTS software and frame builder archive, which is shared with all other lab machines via NFS.

I have moved the new machine into the server rack and copied the contents of /opt/rtcds on the cymac into the shared location. Functionality like sitemap and the CDS tools can now be run directly from the QIL workstation (plus any other machine on which we add the NFS mount).

As it was observed that normal voltage supply is noisy and not suitable for our circuit, we plan to use a voltage regulator that will help us provide a clean supply. Referring to previous elog entries the corresponding corrections were made( polarity of electrolytic capacitors, ceramic cap in parallel to electrolytic, 3V difference between input and output of respective regulators).

Attachment 1: The Circuit Diagram of Voltage regulator

The component used                   Input Voltage                              Output Voltage

a. LM7915                                    -18 V                                  -15.1 V

b. LM7815                                    18 V                                    14.86 V

c. LM317                                      18 V                                   14.96 V      

Attachment 2: Output Voltage noise of regulator circuit

The noise observed using SR785 at the output of each regulator is shown. It clearly shows that LM317 manifests less noise in comparison to LM7915 and LM7815. It will be therefore a good idea to use this to provide 15V bias in our circuit.

Attachment 3: The Scripts

Find all the scripts and data used in this measurement.

Following up from my previous post I measured the phase noise of the Mach Zehnder setup with the AOM driven properly with its VCO. The setup is shown in attachment 1, and compressed data in attachment 2 (goto here to get the python library to decompress this data). The value of M for this data shown is +0.85 V, though I have data for other voltages - however it should affect the performance. Using the low frequency preview I was able to see that I would need to coherently subtract the phase measurements of both measurements, which restricted me to a maximum sampling frequency of 15 kHz.

The measured data is shown in attachment 3. Already you can see that the phase noise of the VCO limits the measured phase from the IFO. This also tells me that previously, when I was modulating the VCO, the AM was affecting the measured phase. When I convert this difference into frequency noise the result can be seen in attachment 4. One upside of this setup is that the signal from the IFO is much more robust, so the PLL can stay locked for longer. This is a consquency of increased drive, 24 dB more, on the AOM.

To me this demonstrates that the way forward is to replace the VCO driving the AOM with a RF amplifier, driven by a low noise (or lower noise) signal generator. The Moku is able to drive at 80 MHz. We have a Mini Circuits ZHL-5W-1 (ZHL--5W) amplifier in the laboratory. This has 46.4 dB of gain, and a maximum power output of 37 dBm. The maximum power that can be input into the AOM is 27.8 dBm (0.6 W). Thus with an appropriate setup this miniciruits amplifier should be a viable repalcement for the current VCO.

After reviewing the Brimrose AOM driver mnual, yesterday, it turns out I was previously using it incorrectly. It is a VCO with the frequency port accepting a DC voltage, between 0 and 10 V to control the frequency of the AOM - note that the mapping is not intuitive so one should refer to the manual. The modulation port is used for amplitude modulation, not frequency modulation. This port, modulation, has 50 ohm input impedance and accepts signals between DC and 10 MHz - modulating the power output. Table 1 below shows the operating parameters we should use:

Following this I characterised the phase noise of this VCO. Results are shown in attachment 1, for various powers. The setup is shown in figure 2, with the data for M = +0.85 V provided in attachment 3. These results show that this VCO has poor phase performance. A value of M = +0.02 V gives the same RF power as when a 0.5 V_pp signal @ 80 MHz was input into the modulation port - as I was previously doing.

This has a few implications for previous measurements:

• The reason it was hard for the Mach Zehnder to remain locked for a substantial duration was due to the low input power to the AOM. This probably resulted in little signal in the Mach Zehnder LO arm.
• Previously the amplitude modulation would have provided a 80 MHz signal in the Mach Zehnder LO arm

The script used to create the smoothed ASD can be found here.

1. Heat: Check the polarity of the electrolytic or tantalum caps. 

2. Add 0.1uF high-K ceramic caps in pararel to these electrolytic or tantalum caps.

3. Why does LM317 have only one volt drop? It requires minimum 3V mergin between the input and output voltages. (See the datasheet) 

The 80 MHz brimrose AOM driver, which came with the AOM, can actually drive between 75 MHz and 85 MHz. It has an input port, which accepts between 0V and 10V, for altering the frequency.

Previously, before Friday 2019-08-09, I had set the offset to + 5V, using a signal generator because that was what was available in the QIL lab. The signal generator, for some unclear reason, had then moved this offset to +4.4 V, when plugged into the AOM driver. Attachment 1 shows a power spectrum, from the BB PD, One notices a large peak negatively detuned from the 80 MHz signal frequency. Attachement 2 shows a zoom in, around 70 MHz, revealing this peak to be two peaks, one near 78 MHz, and another smaller one near 79 MHz.

I decided to adjust the voltage, input to the frequency port, to minimise the RF sidebands I observed. The results of this adjustment can be seen in attachments 3, and 4. Notice that there no observable sidebands. This was achieved with an offset of 4.52V.

Attachement 5 is the manual for the AOM, and its driver. (Removed by KA)

Referring to elog entry QIL:2387. I did the correction with the voltage supply and now provided a supply of 18V(+/-) to LM7815 and LM7915. The position of diodes was also corrected for LM7915. The Electrolytic Capacitors(100uf) I am using are getting heated when using with LM7915 only. I didn't find any tantalum capacitors of 100uf in EE shop. Should they be replaced with some other capacitors?

Attachment 1: The Circuit Diagram Voltage regulator

The component used                   Input Voltage                              Output Voltage

a. LM7915                                    -18 V                                  -15.1 V

b. LM7815                                    18 V                                    14.86 V

c. LM317                                      18 V                                    17 V      

Attachment 2: Output Voltage noise of regulator circuit

The noise observed using SR785 at the output of each regulator.

Referring to elog entry 2385. I did the measurements again because mistakingly I had been using the SR560 with it's AC supply on. This time I used the 12V supply with no connection to the AC supply.

Attachment 1: The Circuit Diagram TIA

>>  the TIA with a gain of 5.1k

Attachment 2: Input Referred noise of TIA 

The input-referred current noise across the TIA was measured using SR785 and was compared against the graph obtained from ZERO simulation.

Attachment 3: Differential Circuit

>> gain of 100

Attachment 4: Input Referred noise of Differential Circuit

The input-referred voltage noise measured using SR785 and was compared against the graph obtained from ZERO simulation. 

Attachment 5: Whitening Filter Circuit

>> gain of 10

Attachment 6: Input Referred noise of Whitening Filter Circuit

The input-referred voltage noise was measured using SR785 and was compared against the graph obtained from ZERO simulation.

Attachment 7: The Scripts

All the scripts and data used in the measurement.

** I did notice a reduction in 60Hz harmonics but I still see a deviation from simulated results at higher frequencies and at frequencies below 10Hz.

3. You need to flip the direction of the diode.

1&2 OK, so the circuits were not fucntioning. Use a dual voltage supply (in a proper cascading setting) and give +/-18V.

4. When you use SR560 as a power supply, you need to disconnect the AC power supply. Otherwise, the AC power, which charges the +/-12V lead battery, contaminates the output voltage with the 60Hz lines.

Answers:

1. For LM317 I received the output of 11.2V and for LM7915 -10V and for LM7815 at 10.4V

2. I did a mistake with the supply. Next time I won't use SR560 and will use a voltage supply instead. 

3. The diode is for protection purpose. How should I use the diode for 7915, should I put it in forward bias or not use it at all?

4. I did check the voltage supply provided by SR560 using a multimeter, they were 12V.

Questions:

1) Has the DC output voltages of the regulators checked?

2) What's the target voltages of the regulator circuits? And how the voltages were supplied from the power supply port of the SR560? 7815 is the regulator meant for +15V and 7915 is for -15V. So the input voltages need to have at least 3V larger voltages than the target voltages (like +18V for 7815, -18V for 7915). If the +/-12V are naitvely applied, the regulators don't reach the operating point.
Check "Voltage Drop" descriptions in the data sheets of the regulator chips.

3) What's the purpose of these diodes? I believe they are for the regulator protection against the transient sign flip during power switching etc as well as over voltageprotection. The circuit of the 7915 has the larger potential difference (like -18V) while the output has -15V. This means the diode will always be on. If this is just a typo in the figure, it's not a big deal. If this is the real situation, it is a big problem.

4) Why were there such huge 60Hz lines? Was the SR560 properly operated with its battery?

A comparison between the types of voltage regulators was done in order to know which should be preferred to provide a clean bias supply.

Attachment 1: The Circuit Diagram of the voltage regulator 

Tere are 3 types of voltage regulators that were tested. 

a. LM317

b. LM7915

c. LM7815

Attachment 2: Output Voltage noise of all 3 voltage regulators

A voltage of 12V(+/-) was provided using SR560 to the respective input of the regulator IC and the output noise across each were measured using the SR785.

LM317 will be a better choice to make a voltage regulator for the circuit mentioned in elog entry 2381.

Input Referred noise is calculated for the circuit that is to be used for characterization of photodiodes. For the biasing 12V was used from SR560 as it provides cleaner voltage as compared to other voltage supplies. 

Attachment 1: The Circuit Diagram TIA

>>  the TIA with a gain of 5.1k

Attachment 2: Input Referred noise of TIA 

The input-referred current noise across the TIA was measured using SR785 and was compared against the graph obtained from ZERO simulation.

Attachment 3: Differential Circuit

>> gain of 100

Attachment 4: Input Referred noise of Differential Circuit

The input-referred voltage noise measured using SR785 and was compared against the graph obtained from ZERO simulation. 

Attachment 5: Whitening Filter Circuit

>> gain of 10

Attachment 6: Input Referred noise of Whitening Filter Circuit

The input-referred voltage noise was measured using SR785 and was compared against the graph obtained from ZERO simulation.

Some points that were observed: 

*** I am observing deviation from simulated results at higher frequencies. Presently, I am unable to understand the cause of this deviation. 

*** At low frequencies deviation from simulated results is perhaps caused due to 60Hz harmonics and 1/f noise.

Input Referred noise to be calculated for trans-impedance.

Attachment 1: The Circuit Diagram on paper,

>>  the TIA with a gain of 5.1k

Attachment 2: Noise across TIA 

The input-referred current noise across the TIA was measured using SR785 and was compared against the graph obtained from ZERO simulation.

** This time, I divided the measurements into 7 parts, 0-800, 800-2.4k, 2.4k-5.6k, 5.6k-12k, 12k-24.8k, 24.8k-50.4k, 50.4k-101.6k. The number of points for each was 800. Hanning Window function was used in the template file and the Input channels were grounded. 

Attachment 3: Noise across TIA 

The input-referred current noise across the TIA was measured using SR785 and was compared against the graph obtained from ZERO simulation.

** This time, I divided the measurements into 1 part, 10-6.4k. The number of points for each was 800. Hanning Window function was used in the template file and the Input channels were grounded. 

** To measure the noise the output was measured at the pin6 of the OpAmp.

** A source of 1V was applied to the circuit by keeping a 10k in series with the input of the circuit when Transfer Function was being measured.

** It's difficult to avoid 60Hz harmonics with a circuit kept in open as this one. Lots of its effects are visible in the plot.

Attachment 4: Noise of the power supply used

The power supply was observed to be used to be noisy. 

Here I follow, essentially, the same setup as in attachment 2 of elog 2377. The difference is that I measured the LO path and IFO paths sequentially, not concurrently. This allowed me to measure at 125 kilosamples per second. Note that there is an error in the pymoku conversion tool when recording on channel 2 only. To circumvent this record both paths with channel 1.

Shown in attachment 1 is the frequency noise of the 2um Mach Zehnder interferometer. Attachment 2 shows the measured phase noises of the individual paths. Attachment 3 contains the plotted frequency spectra, and phase to frequency conversion factor. The conversion factor from phase, in radians, to frequency is:

The table below gives the values necessary to calculate this. It is 2.73 x 10⁶ Hz rad^-1.

The smoothed spectra below were generated by this code. The full time series data is too large to upload here, at 272 MB when doubly compressed. Low frequency data is difficult to obtain due to the length of time that there is sufficient SNR for the phasemeter to lock onto the 80 MHz signal. Locking the Mach Zehnder could resolve this problem.

always keep the start frequency at 0 Hz no matter what the span, as I was showing you yesterday in the lab. Otherwise, the bin centers end up in weird places.

Input Referred noise to be calculated for trans-impedance.

Attachment 1: The Setup,

The setup was made on a breadboard, as per the circuit diagram.

Attachment 2: The Circuit Diagram on paper,

>>  the TIA with a gain of 5.1k

Attachment 3: Noise across TIA on SR785 screen

This is the noise as seen on the screen of the SR785

Attachment 4: Noise across TIA captured using the SR785 template file

The input-referred current noise across the TIA was measured using SR785 and was compared against the graph obtained from ZERO simulation.

** There is a difficulty in capturing the exact form of data as displayed on the SR785. Previously, I had captured the full span by keeping the start frequency as 1Hz. This gave only two points between 1Hz and 100Hz, all the rest of 798 points out of 800 were plotted at a higher frequency. This gave a straight line of higher magnitudes at low frequency(1Hz to 100Hz) since there were only 2 points. But the plot on the SR785 screen looks different.

** This time, I divided the measurements into 3 parts, 1Hz to 100 Hz, 100Hz to 10kHz and 10kHz to 100kHz. This left me with a graph which looks clumsier than the previous ones. I guess if there is a way that the template files of SR785 can be modified then it can give a graph which will align properly to the simulated results. For the time being, I can try again to take measurements in 3 sections but this time with fewer(300 instead of 800) points.

** It's difficult to avoid 60Hz harmonics with a circuit kept in open as this one. Lots of its effects are visible in the plot.

Oh no! We've lost all of the low frequency data (which is the whole point of this experiment) and all of the contributing noise sources to the circuit!

Attachment 1: The Circuit Diagram,

>>  the TIA with a gain of 5.1k

>>  Differential Circuit with a gain of 100.

Attachment 2: Noise across TIA

The input-referred current noise across the trans-impedance amplifier was measured using SR785 and was compared against the incoherent sum of input-referred current noise graph obtained from ZERO simulation.

Attachment 3: Noise across Differential circuit

The input-referred current noise across the Differential circuit was measured using SR785 and was compared against the graph obtained from ZERO simulation.

During measurement of transfer function using SR785, a source of 1V was given to a 10k resistor which was connected in series with the circuit taken into consideration. The channels of SR785 were set to AC coupling and input was set to Ground. Apart from that Hanning window function was used for measurements from SR785.

Great! Can you convert this into the laser frequency noise Hz/rtHz? I believe this [rad/rtHz] was still the measured phase noise and was neither the laser phase noise nor frequency noise yet.

Figure 1, attached, shows the phase noise I measure from the 2 um Mach Zehnder interferometer. This phase noise is from the optical path and is affected by:

• Intensity noise.
• Laser frequency noise.
• Differential polarisation shifts through the IFO arms.
• Changes in differential arm length.
• Changes in refractive index.
• Detector noise, in which I am also grouping noise due to the amplifier.

The next step is to determine contributions from each of these sources.

Attachment 2 shows the setup used to measure this noise. Attachment 3 shows the measured phase spectra from both phasemeters of the Moku. Attachment 4 contains the both time series and spectra. The spectra shown here are:

1. Converted to radians, from cycles.
2. Subtracted, in time domain.
3. Converted into power spectra, using Welch's method, 30 averages and Hanning windows.
4. Converted into amplitude spectra, by taking the square root.

Attached are the results from measuring the phase noise of the Moku with a SRS DS345, which was the available signal generator in the QIL lab. Attachment 1 is the estimate of the Moku phase noise, with attachement 2 containing the data. Attachment 3 shows the data this was obtained from, and attachment 4 the experimental setup used to measure this. 15.625 kHz is the maximum frequency that the Moku will let you acquire two channels of data, in the phasemeter setup. In this setup I generated a 20 MHz signal on the DS345. This was then split, via a T piece, and sent through cables of length 0.6414 m to inputs 1 and 2 of the Moku. Concurrently the 10 MHz synchronisation signal was sent, also through a 0.6414 m cable, with an additional elbow piece, from the DS345 output to the Moku 10 MHz input. The phase difference between input 1 and input 2, shown in yellow orange in attachment 3, should be sqrt(2) times the phase noise of the Moku.

Attachment 5 shows the same setup, but with the spectrum straight from the Moku, which can be acquired at a maximum speed of 488 Hz. They are relatively consistent in the overlapping frequencies. Attachment 6 shows what happens when the 10 MHz synchronisation is removed. The low frequency performance of individual channels suffers but there is not a substantial change in the difference, yellow orange.

Attachment 8 shows the phase difference performance at 10 MHz, with attachment 7 showing the setup for this measurment. The cable length for this measurment was 0.9993 m for all paths. Again the difference, yellow orange, is comparable to other measurements.

There were some errors in the previously obtained transfer function of the part where TIA was in series with the Whitening Filter. So, below is the updated transfer function plot.

Attachment 1: The circuit diagram

The usual transimpedance configuration was made using the OP27 IC. There are two TIA as we will be using two PD in our final circuit. The two TIA are connected to Whitening Filter respectively. The Whitening filter has a gain of 10. Apart from that, the output of the two TIA is connected to a differential circuit and whose output is in turn connected to another independent whitening filter.

Attachment 2: The Transfer Function

The transfer function of the transimpedance amplifier, Differential Circuit(at both its input nodes), Whitening Filter was analyzed by using the SR785 spectrum analyzer. A 1V source was applied at the input of a 10k resistor which was in turn connected in series with the respective circuit under consideration. The averaging mode is RMS and the input ground node is floating. The input is AC coupled.

The terminology used in the graph:

TIA- Transfer function of the transimpedance amplifier

Differential at node1- TF of Differential circuit by providing source at node1

Differential at node2- TF of Differential circuit by providing source at node2

Whitening Filter- TF of Whitening filter by providing source at node1

TIA and Whitening filter- TF of transimpedance amplifier in series with the Whitening filter, and the source were at node1 of the TIA.

The transfer function of the circuit was analyzed.

Attachment 1: The circuit diagramThe usual transimpedance configuration was made using the OP27 IC. There are two TIA as we will be using two PD in our final circuit. The two TIA are connected to Whitening Filter respectively. The Whitening filter has a gain of 10. Apart from that, the output of the two TIA is connected to a differential circuit and whose output is in turn connected to another independent whitening filter.

Attachment 2: The Transfer Function

The transfer function of the transimpedance amplifier, Differential Circuit(at both its input nodes), Whitening Filter was analyzed by using the SR785 spectrum analyzer. A 1V source was applied at the input of the respective circuit under consideration. 

**A 1k resistor was kept in series with every node which was given a source of 1V.

The terminology used in the graph:

TIA- Transfer function of the transimpedance amplifier

Differential at node1- TF of Differential circuit by providing source at node1

Differential at node2- TF of Differential circuit by providing source at node2

Whitening Filter- TF of Whitening filter by providing source at node1

TIA and Whitening filter- TF of transimpedance amplifier in series with the Whitening filter, and the source were at node1 of the TIA.

We received the TO-66 sockets for LaserComponents PDs (Andon Electronics F425-1009-01-295V-R27-L14 Qty.10). It is made of FRP. It is very nicely made.

The transimpedance amps, differential circuit and the whitening filter for the 2um Extended InGaAs detectors were made and their noise level was evaluated. The examination of noise was done without the diodes. For every analysis, SR785 spectrum analyzer was used and a simulation using zero in python was also done. The SR785 was controlled using the python program to get the data. The input was AC coupled and Hanning window function was used during the task of getting data.

**The noise of SR785 spectrum analyzer is also mentioned, which was measured by deploying a terminator at one channel. The noise labelled as SR785noise mentions the spectrum analyzer noise alone.

**The noise of the components, with SR785 mentioned along with them, indicates the noise observed using the Spectrum analyzer.

Attachment 1: The circuit diagram

The usual transimpedance configuration was made using the OP27 IC. There are two TIA as we will be using two PD in our final circuit. The two TIA are connected to Whitening Filter respectively. The Whitening filter has a gain of 10. Apart from that, the output of the two TIA is connected to a differential circuit and whose output is in turn connected to another independent whitening filter.

Attachment 2: The amplifier noise Part I

The noise of TIA1 was analyzed using the SR785 spectrum analyzer and also by simulating the circuit using zero in python. 

Attachment 3: The amplifier noise Part II

The noise of TIA2 was analyzed using the SR785 spectrum analyzer and also by simulating the circuit using zero in python. It was observed to be exactly similar to that observed of TIA1.

Attachment 4: The differential circuit noise levels

The noise level of the differential circuit was measured by shorting both the input terminals and observing the output at the pin6 of IC.

Attachment 5: The Whitening Filter noise level.

The noise of the Whitening filter was measured by grounding the input terminal (pin 3) at taking the measurement at the output. The gain of the filter is 10. 

Attachment 6: The TIA and Whitening Filter, connected together, noise level.

The TIA and Whitening filter were connected in series and the noise was observed at the output of the whitening filter.

Attachment 7: The Simulation of Circuit in Zero

The complete circuit was configured using zero. It will help us analyze the noise that we are not expecting. Please open it using Jupyter Notebook.

Two new 27" LED monitors arrived today for the QIL workstation. I've installed them.

Chris and I set up the LIGO RTS environment on the QIL cymac, using code copied from the cryolab cymac. Specifically, the script /opt/rtcds/rtcds-user-env.sh was edited to match the cryolab version and added to the /home/controls/.bashrc file. We also downloaded a copy of the CDS user apps SVN to /opt/rtcds/userapps/release. Tools like dataviewer and ndscope now work on the cymac (fb4: 10.0.1.156).

Our plan is to set up a network drive on a third machine to host the /opt/rtcds directory currently located on the cymac. This way, the directory can be shared with any number of workstations as well as the cymac itself, and the NFS mounts will be unaffected by frequent reboots of the cymac.

I also unsuccessfully attempted to diagnose the race condition that occurs between all the RTS services on boot. Right now the services all start correctly only about 1/3 of the time. I tried setting the order that the services are started and adding a 15-second delay after each service start. However, this did not make things become deterministic.

I finally succeeded getting Debian installed on the new workstation with a working network card. I installed Debian 10.0, which was just released last week and will be supported for five years. After installing the OS, I

• Configured the machine (network interface, user account, etc) following my standard procedure. 
• Installed the cds-workstation superpackage following Jamie's instructions.

The user name is controls as usual and it has the standard W. Bridge password. The lscsoft repo for Buster (Debian 10.0) is still missing many packages, so I installed the cds packages for Stretch (Debian 9.9) instead. They seem to be compatible with 10.0 as far as I can tell. The machine is at the same IP as the one it replaced, 10.0.1.33.

To be able to interface with the cymac, there is still an RTS environment (environment variables and an NFS mount) that needs to be set up. I'm looking into what this involves.

The amplifier sets for the thorlabs 2um PDs were delivered to the lab.

- PD1 and Amp1, PD2 and Amp2 are the proper combination. If a high quality power supply is used, it is not an issue.
- The cables for the external bench supply or the 9V batteries have been made.

I moved a new Dell Precision 3430 onto the lab bench near the door. It's replacing the older unused machine that was in that spot (IP=10.0.1.33). The intention is for the new machine to provide MEDM (sitemap) access to the QIL cymac and to run the full CDS utils suite (awg, NDS, etc.).

There are two operating systems that have CDS package support: Debian 9 and Scientific Linux 7. Unfortunately neither of these operating systems is officially supported for this model computer according to Dell.

I attemped to install base Debian 9.9, which can be done successfully and booted. However, the installer is unable to locate the drivers for the network card, leaving the machine without network capability. This is likely because the network card (Intel i219-LM) is brand new and support hasn't been incorporated into the distributions yet. I next tried installing the latest weekly snapshot (testing build) of Debian with additional commercial firmware included. This time the network card was recognized, and the installation appeared to complete entirely successfully. However, the system then failed to boot the new OS. I tried first just reinstalling the GRUB (boot) loader, then entirely reinstalling the OS, both with the same result. Something in the test build is preventing the hard disk from being recognized as a bootable device.

I have one more idea to try next week: Again install base Debian 9.9 (without network capability), then attempt to manually install the i219-LM drivers provided by Intel.

The transimpedance amps for the 2um (unamped) InGaAs detectors were made and evaluated.

Attachment 1: The circuit diagram

The usual transimpedance configuration. The detector (Thorlabs DET-10D) is an extended InGaAs which is sensitive up to 2.2um. I believe the detector is biased to 1.8V although it is not obvious and the 12V battery is used. The feedback resistor was chosen to be 5kOhm so that the circuit can handle up to ~2mA (~1.7mW). The feedback capacitance pf 100pF for compensation was chosen kind of arbitrary to keep the circuit stable and also the RC cut off to be more than 100kHz. The output resistance is 100Ohm. The selection of the opamp is described below.

Attachment 2: The amplifier noise Part I

The amplifier noise (the first unit called Amp #2)  was evaluated with the opamp swapped with OP27 (BJT), LT1128 (BJT), OPA604 (FET), and LT1792 (FET), chosen from the 40m stock. For the given environment, the FET amps exhibited better performance while the BJT amps suffered from more line noise coupling and the larger 1/f noise. Particularly, LT1792 reached at the level of ~2pA/rtHz, with lower line noises. This looks the best among them. Note that the 5kOhm feedback resistor gives 1.8pA/rtHz current noise. 

Attachment 3: The amplifier noise Part II

Then the second unit (called Amp #1) was made. This unit has more high-frequency noise. It turned out that the noise was coming from the power supply which was the +/-12V from the rear panel of an SR560 which was connected to the AC power. The noise dramatically went away with the battery mode operation of SR560 (by disconnecting the AC power). The floor level was 2.2pA/rtHz and it was slightly higher than the quadratic sum of Johnson noise of 5kOhm and the voltage noise of the amp (4nV/rtHz). This noise level was just sufficient for the purpose of the 2um detector.

Attachment 4: The detector noise levels

Now the detector #1 and #2 were paired with the amp #1 and #2, respectively. In fact the detector 1/f noise was way too large compared to the amplifier noise. There is no hope to detect shot noise level of the mA photocurrent. 

Attachment 5: The detector response

The detector response of each PD+AMP pair was measured using Jenne's laser and Thorlabs PD10A (~150MHz). There was some systematic error of the absolute level calibration, therefore the transfer functions were adjusted so that they have 5kOhm transimpedance at ~1kHz. The phase delay is ~30deg at 100kHz. This partially comes from the combination of 100pF//5kOhm and the ~4MHz bandwidth gain of the opamp. If we want faster response we need to modify these.

I've removed the Moku and its iPad from the lab to see what its up to noise-wise.

With the help from Chub, we opened the vacuum chamber. The procedures are not complicated. Here are some notes, both about the opening procedures and some other things I learned from Chub.

1) Vent the chamber by opening the venting valve slightly. The venting process needs to be slow - we cover it with a wipe, as shown in the photo 1. We stop opening the valve when we hear the air wheezing.

2) The thing with a screen is a gauge, which gives the pressure inside, but it is not quite working - always showing atmosphere pressure. Chub left me a document on it; I should be able to get it working.

3) We removed the screws of the top chamber. Correct hex keys is necessary - I was going to use one that loosely fits but Chub said it is not good.

4) We lifted the cover of the chamber with the crane and put it besides the chamber. We should release the break of the crane first and then break it again when we land the weight.

5) The pump connection is not very good - leaking oil both into the chamber and onto the floor.

6) The surface under the chamber (shown in photo 2) cover can only be cleaned by dry wipe or alcohol. 

Next I am going to set up the optics, electronics and whatever inside before we close it and pump it down.

There was a correction in the script I used to estimate the frequency noise from the inphase and quadrature component. Attachment #1 shows the frequency noise estimated after the correction. 

I have also attached the Matlab script ( I am not able to attach the zip file with data files). I remember, while saving the data, we gave the time duration as 70 s. But while processing the data only I realised that the  time domain data is captured only upto 2 s. Even in this case, I would expect the frequency axis to start from 0.5 Hz, but I don't see that in the FM noise plot. Kinldy let me know whther I am doing anything wrong in data proocessing. 

We installed a new rack in the QIL to help store vacuum equipment and other parts. See attached photos.

Rack has casters at the bottom. They're locked down but if we decide we don't like them, we can sawp them out for feet.

Attachment # 1 show the schematic of the lock in amplifier configuration used in Moku lab. We saved the in phase and quadrature components.

In phase =

quadrature =

where   corresponds to 2 -micron ,  is the delay time in the delay fiber  and .

  is the phase noise of the laser. 

From the inphase and quadrature component value is extracted. So, we are actually extrating the combined effect of phase noise of laser as well the phase noise due to fiber length fluctuations due to environmental fluctuations. ASD of this is converted to frequency noise in Hz/rt Hz. Attachment # 2 shows the frequency noise estimated from two sets of measurements. This curve exhibit a 1/f characteristics from about 240 Hz upto 30 kHz

[Aidan, Chris, Koji]

We went down to the lab to check the situation of the setups for 2um laser measurement and stabilization and the new cryostat.

[2um laser frequency noise measurement]

• Looked at the add-on transimpedance amps: Something was wrong with them. The power bypass caps are attached to the "hot" supply lines in parallel (both sides of the caps are soldered to a same line). And some power supply lines have no voltage. This circuit is not necessary to be bipolar. To be fixed (KA)
• We temporarily connected one of the thorlabs 2um InGaAs biased detectors to an SR560. It showed reasonable output: DC/AC response OK & no nonsense.
• The AOM was bypassed and the homodyne fringe was checked.
  The fringe visibility was low (~10%) and was dependent on the stress applied to the delayline fiber.
  Suspected polarization rotation somewhere -> ToDo: Check the polarization states of the output beams.
• ToDo: Check should be done with each component. how much are the output power, output polarization, dependence/fluctuation of the polarization, etc. 
  We might be able to use the 2um Faraday Isolators (as PBSs) for the measuement.
• Checked the fringing of the fiber delayline Mach Zehnder. We observed one fringe per sec level fluctuation.
• Laser current actuation was checked and it turned out that it is so strong and sufficient to lock the delayline fringe.

[2um AOM]

• The fiber coupled AOM gave us a reasonable amount of DC/AC actuation of the laser intensity.
• The power of the 1st order output has the dependence on the "freq input" of the driver. This is probably because of the matching between the fiber coupling and the deflection angle, which is freq dependent.
• When the freq input is 8.8V_DC, the 1st order output has the maximum efficiency. The efficiency was 96%@990mV_DC input to the modulation in.
• The AOM actuation bandwidth was tested to be ~MHz, at least.
• We are not supposed to give more than 1V to the modulation in while we want to apply 8.8V to the freq input. Incorrect plugging may cause the damage of the modulation input port. The setup needs to be improved with a protection circuits / AOM driver circuit.
• Our understanding is that the modulation input has a 50Ohm input impedance while the freq input has high-Z
• The next step towards the intensity stabilization is low noise photodetector circuits and proper interface to the AOM driver.
• Also we want to set up TECs and other circuits for the LaserComponents PDs.

[Cryostat]

• Cleaning: there are many components are scattered on the table.
• Plan:
  • Move the Zack rack to the next of the optical table (or somewhere)
  • Move the yellow chemical cabinet to the place where the rack was. We can pile up some plastic boxes on it.
  • Remove the delicate optics from the steel table.
  • Place heavy cryostat components on the steel table.
  • Connect the cryo cooler to the cryostat. How do we do that? Fisrt rigidly attach for testing and then move to soft attaching?
  • Replace the optical windows to the 2um ones (2"). The current ones are for 1.5um.
  • We need a 2" 50/50 BS at 2um. Lenses and steering mirrors are in hand.