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ID Date Author Type Category Subject
  1188   Mon Jan 12 13:10:08 2015 DmassNotesCavityCavity Parameters from Dec 2014 Labwork

  1187   Fri Dec 19 21:37:12 2014 ZachLaserSiFiAmplitude modulator characterization

Tonight, I did some characterization of the Photline fiber-coupled amplitude modulators we will use for our experiment (MXAN-LN-10 --- datasheet attached nope google it yourself). These are electro-optic devices that work by using an internal mach-zehnder to convert phase modulation into amplitude modulation.

The test setup for all measurements was the same. I used the exact configuration that I have been using for the beat (see CRYO:1182), but I simply blocked one laser, so that only one beam was hitting the 1811 PD. The amplitude modulators were inserted (one at a time) between the East laser and its output coupler.



Insertion loss

The first thing I did was to investigate the insertion loss of the modulators. We chose the low-loss option, which just meant that the company hand-selected modulators with loss of < 3dB (= 50% power transmission).

I didn't go crazy with precision here, because systematics with fiber coupling can easily prevent a measurement to better than a few percent (an example of this: I installed a 1-meter patch fiber between the laser and the output coupler, instead of the modulator, and I actually saw a slight increase in output power vs. the case with the laser going straight to the output coupler… go figure).

In both cases, I measured very nearly 50% reduction in power (at the top of the MZ fringe---see below) vs. the case with no modulator. So, these things have a loss very close to 3 dB, as advertised. An important thing to point out is that we will need to bias these away from maximum transmission to get a linear PM -> AM coupling, so the real power reduction in our setup will be more than 50%.


DC response

These modulators have an SMA-connectorized "RF" input, as well as two bare pins connected to a separate set of "DC" electrodes (they also have two more pins connected to the cathode and anode of an internal PD, presumably at the other MZ output port, which is kind of cool). As far as I can tell, the RF input is also DC coupled, only it is 50-ohm terminated.

I did a DC sweep of both electrodes from 0-10 V while measuring the output power:


(The RF applied voltage range is lower due to sagging from the 50-ohm load).

Fitting these curves, I determined the following Vpis:

  • S/N 03
    • DC: 6.46 V
    • RF: 4.19 V
  • S/N 17
    • DC: 6.39 V
    • RF: 4.91 V

These are consistent with the numbers listed on the datasheet.


Transfer functions

Next I measured the actuation transfer functions ([RIN/V]) from 1 Hz to 100 MHz, driving the RF input while applying a mid-fringe bias to the DC input, and using

  • Agilent 35670A FFT analyzer and the 1811 DC output for 1 Hz - 50 kHz, and
  • Agilent 4395A RF analyzer and the 1811 AC output for 500 kHz - 100 MHz

Note the dead zone from 50-500 kHz---this was by accident, as I forgot to check the low-frequency resolution of the RF measurement. I will redo this sometime.

Here are the results:



  • The jump from 50 kHz - 500 kHz is from the measurement dead zone and carries no information
  • The lag beginning around 10 kHz is from the stated ~50 kHz bandwidth of the DC output of the 1811. The AC output has a low end at ~25 kHz, so there isn't really a good way to make a measurement in this region with that detector. We could use a DC-coupled version to make a continuous spectrum.
  • The slow rollup at low frequencies is well-sampled and repeatable. I'm not sure what causes it, but it appears to be real. In any case, it's pretty small.
  • The delay at high frequencies is consistent with the optical path length from the modulator to the PD. I calibrated the cables' transfer function out, and what is left is this delay which has a 4.13-m free-space equivalent. There is ~64 cm of free-space travel on the table, plus well over a meter from the output fiber of the modulator.

The response very flat, and roughly what is expected from the DC sweep:

(1/P0) * dP/dV|mid-fringe = pi/Vpi ~ 0.5 ( = -6 dB).

  1186   Fri Dec 19 18:02:33 2014 ChrisLab InfrastructureControl Systemcymac model at 64K rate

I upped the sample rate of the x1cry model to 64K, in the following way:

  1. Change rate=16K to rate=64K in the Simulink model
  2. Recompile, reinstall
  3. Fix the filter coefficients in the Foton file

The only tricky part is the last step. Changing the sample rate requires the filter coefficients to be updated, so they still match the filter designs. But when you open the filter file in Foton, it does the opposite: updates the designs so they match the old, incorrect coefficients. Since x1cry had only a few filters defined, I went through the file and reverted the designs by hand. (Newer versions of Foton would let you automate this step.)

  1185   Thu Dec 18 03:39:32 2014 ZachLaserSiFifree-running laser frequency noise

I spent some time tonight measuring the free-running laser beat noise in various ways. Recall that, as of yesterday, I had tried setting up a couple analog PLLs to no avail and I didn't trust the spectrum I was getting from the Zurich PLL. So, I wanted to measure it another way to see if I could corroborate.

First, eye candy:


Now, an explanation of the various measurements.

I-Q demodulation method


This is a method I have used with some success in measuring the Marconi noise in its quietest state (with no modulation and therefore no means of feedback---see ATF:1877). It is done in the following way:

  1. Split the beat PD output and send it to the RF input of two mixers (I used level-7 ZAD-1-1s), using equal path lengths.
  2. Set Marconi to a frequency close to the beat (~50 MHz in this case) and an amplitude of +10 dBm
  3. Split the Marconi output, send one splitter output to each mixer from (1), but with 90º rotation between them.
  4. The outputs of the mixers are now at the difference frequency between the beat and the Marconi, but maintain their I-Q separation. (This is the reason for using the Marconi rather than beating the lasers at a lower frequency in the first place---the I-Q separation is maintained regardless of the differential laser drift, and it also only requires a short cable length.)
  5. Acquire both I and Q signals and perform the I-Q analysis:
    1. Normalize the signals and atan2(I,Q) to get phi, then unwrap(phi) to get continuous phase evolution vs time
    2. diff(detrend(phi))/diff(t)/2/pi to get instantaneous frequency as a function of time
    3. pwelch

The main complication here is that, as you can see in the plot, the high-frequency RMS of the beat is several tens of kHz, which means you still have to sample at a high rate to get what you need. The best acquisition scheme I could think of was the Zurich box, which can do 460 kS/s. Still, to take meaningful data, I had to very carefully tune the laser beat to the Marconi LO and then quickly engage acquisition before the (wildly fluctuating) IF signals drifted above the Nyquist frequency (around one second of data was used to make this trace).

That said, the result doesn't look that crazy, and in fact it agrees very well with the DFD measurement that was carried out in a completely different way (see below).


Delay-line frequency discriminator (DFD) method


This is the usual scheme where one mixes a signal with a time-delayed version of itself to create dispersion. What I did:

  1. Split the PD signal
  2. Using one splitter output, find the appropriate combination of attenuators and amplifiers needed to obtain a LO-worthy +7-dBm signal (I needed -7 dB and then ~+25 from a ZFL-500-LN) and send it to a mixer LO input via a long (several-meter) cable.
  3. Send the other output to the mixer RF input via a short cable (attenuate if necessary---wasn't in my case).
  4. Verify that the DC level of the IF output varies sinusoidally with the beat frequency
  5. Null the output and measure the frequency resolution. I measured 5.5 nV/Hz.
  6. Amplify with SR560 and measure spectrum on spectrum analyzer
  7. Divide spectrum by SR560 gain and the number in (5) to get frequency noise

This method worked swimmingly and reproduced exactly the result I found using the I-Q scheme. The noise floor (cyan in the plot) was measured by sending a quiet Marconi sine wave of the same amplitude and frequency as the beat through the pipeline.


Zurich PLL method

This method is incredibly straightforward. Simply plug the beat (ensuring it's < 1 Vrms and under 50 MHz) into the Zurich box and lock the internal PLL by pressing "ON" on the screen. Route the PLL control signal to one of the front panel outputs and choose the scale factor in V/Hz. I chose the same number as I measured for the DFD (including the SR560 gain) for ease of comparison on the spectrum analyzer.



  • All methods agree below ~50 Hz 
  • The I-Q and DFD methods agree everywhere, but they are higher than the PLL result by ~2 from 50 Hz to around 10 kHz, above which they re-converge somewhat
  • All traces (save for the PLL in a narrow band from ~50-500 Hz) are higher than those on the spec sheets sent with the laser (shown in black on the plot---note that the West laser is everywhere noisier than the East one).

I'm not sure what to believe. One would think the Zurich PLL is the most trustworthy, but a) I still am bothered by the time-domain behavior I see in the PLL control signal when I adjust the laser beat while watching it, and b) I've generated two nearly identical spectra that differ from it using completely different schemes from measurement to FFT.

All that said, I think the excess noise (and thanks to Dmass for saving me time by pointing this out) is just coming from the ThorLabs drivers, so this should be redone when we have our low-noise ones.


  1184   Wed Dec 17 18:11:38 2014 ZachLaserSiFiLasers mounted, energized, beat set up



If the "locked indicator" light is not green on the Zurich (first tab, under "Reference", then what you get out is junk (e.g. you have unlocked the lock in, and i hasn't re-acquired yet) - you can do this by kicking it too hard with a frequency shift, which would be easy to do if you were slewing laser frequency, as the coefficients of the laser [Hz/mA] is so big. When the lock in loses the signal, you have to manually re-lock it (toggle off and on the button which has the mouseover text: "enable the fixed center frequency mode of the PLL"). You can get  something which sort of looks like a PLL signal which has terrible noise and weird glitchy response when the lock in isn't locked in.

Your instinct to look for slewing at the PLL control point is correct, and a sign that the state of the PLL is healthy/unhealthy


 Yes, I noticed this effect. I'm talking about immediately after acquiring---or re-aquiring---PLL lock. I did this several times at different beat frequencies to see what effect it had on the noise (the spectrum changed considerably, which is another bad sign).

  1183   Wed Dec 17 14:40:15 2014 DmassLaserSiFiLasers mounted, energized, beat set up

  • With Dmass's help, locking a Zurich PLL to the free-running beat. This appeared to work, and we saw a preliminary frequency noise spectrum that looked about right, but I'm skeptical because the control signal doesn't seem to respond to my slewing one laser's frequency
  • Briefly, locking one laser to the other at low frequencies using the Zurich PLL control signal as a frequency discriminator. This didn't work, adding to my suspicion.

If the "locked indicator" light is not green on the Zurich (first tab, under "Reference", then what you get out is junk (e.g. you have unlocked the lock in, and i hasn't re-acquired yet) - you can do this by kicking it too hard with a frequency shift, which would be easy to do if you were slewing laser frequency, as the coefficients of the laser [Hz/mA] is so big. When the lock in loses the signal, you have to manually re-lock it (toggle off and on the button which has the mouseover text: "enable the fixed center frequency mode of the PLL"). You can get  something which sort of looks like a PLL signal which has terrible noise and weird glitchy response when the lock in isn't locked in.

Your instinct to look for slewing at the PLL control point is correct, and a sign that the state of the PLL is healthy/unhealthy

  1182   Wed Dec 17 13:54:19 2014 ZachLaserSiFiLasers mounted, energized, beat set up

On Monday, after I did some inventory of all the parts we have received from various companies, Dmass helped me mount the RIO lasers into their mounts so that I could get started with the optical setup. We cleaned the surfaces with methanol, applied a small layer of silver thermal compound, and then screwed them in.

I then borrowed the following to run the lasers:

  • The (separate) ThorLabs diode driver and temperature controller from Haixing's maglev setup
  • An integrated ThorLabs diode driver / temperature controller from the TCS lab

After finding the right cables, I powered up the lasers and verified the P-I curve for each as listed on the spec sheets.

I then built a quick (temporary) optical beat setup, combining the two beams on an 1811. I had the temperatures (actually, thermistor resistances) set to what was listed as the testing set point on the datasheet, and as soon as I overlapped the beams and focused them onto the PD, there was already a strong ~50 MHz optical beat.

diagram.jpg setup_with_beat.jpg

I have spent some time since then trying to lock various kinds of PLLs, both to interrogate the free-running frequency noise and to get used to controlling the lasers. Some things I've tried:

  • Locking a Marconi to the free-running beat, which I think might be an exercise in futility due to the relatively small range of the Marconi FM
  • Locking one laser to the other directly using a PLL, which I think might be an exercise in futility due to the bandwidth of the current actuation from the ThorLabs driver
  • With Dmass's help, locking a Zurich PLL to the free-running beat. This appeared to work, and we saw a preliminary frequency noise spectrum that looked about right, but I'm skeptical because the control signal doesn't seem to respond to my slewing one laser's frequency.
  • Briefly, locking one laser to the other at low frequencies using the Zurich PLL control signal as a frequency discriminator. This didn't work, adding to my suspicion.

The first two were not helped by the fairly basic loop shaping afforded by attenuators and an SR560.

I think my next step will be to simply use the I-Q demodulation method (like I did to measure the no-FM Marconi noise in ATF:1877) to measure the frequency noise. I'll compare that to what I get with the Zurich PLL.

  1181   Wed Dec 17 13:24:00 2014 ZachLab InfrastructureSiFiCryostat unpacked (x-post from SUS elog)

(I realized that we should probably use the CRYO elog rather than the SUS one, so I've reposted this here).

[Nic, Zach]

Today, we unpacked the IR Labs cryostat that will be the centerpiece of the Cryo SUS experiment. 

Everything was more or less in order, except that the baseplate does not have any outward extensions with which to mount the cryostat to the table. Also, the holes for the screws holding the baseplate to the barrel are not countersunk. So, as of right now, the entire cryostat sits on these screws' caps, which is not ideal. We need to either a.) get a new baseplate made up with some wings on it and countersinking for the screws, or b.) work out another way to hold and mount the cryostat (for example, we might want some soft isolating material there anyway, though it will come at the expense of alignment drift).

I followed the instructions and removed the strange anodized heat shield bottom plate that comes with it during shipping, replacing it with the usual one and then resealing the chamber. As directed, I also pumped out the air again---the charcoal getter is not supposed to be exposed to atmosphere for long periods of time.

  1180   Fri Dec 12 17:15:00 2014 DmassNoise HuntingNoise BudgetWest Photothermal Transfer Function Actually Measured

Photothermal (and other absorption effects) actually measured.


Cavity loss at 300K is 10ppm for west cavity. Fit looks very good, and jives with our understanding of the transfer functions. More to follow.

Attachment 1: W_Ptherm300K.png
  1179   Mon Dec 8 16:12:28 2014 ChrisElectronicsGeneralUsing the prologix gpib adapter

The 4395A network analyzer in the Cryo lab takes ages to save data to floppy. Someone had hooked up a Prologix GPIB-ethernet adapter to it, but it wasn't working on our network. I set it up as follows:

  • Prologix adapter: autoconfigure by DHCP
  • Cryo router's DHCP server: reserve IP address for the Prologix

So, it acts like it has the static address while on the cryo network -- but you can still take it to other networks and use it without monkeying with the settings.

Python scripts copied over from the 40m are installed on gaston, in the directory ~controls/netgpibdata. The AG4395A.py script was tested and works (much faster than the floppy drive).

While testing, Nic and I found and disabled a rogue DHCP server running on the framebuilder.

  1178   Sun Dec 7 12:42:55 2014 ranaNoise HuntingNoise BudgetScatter Noise hump is RAM


 I think that if you see a signal in the demodulated PDH error point with the cavity unlocked, that this must be RF AM on the light.

What mechanism would produce this much AM? It can't be made through common path modulation of the carrier and sidebands. It must be an etalon formed somewhere between the EOM and the input mirror of the suspended cavity. Could be windows / viewports; this can be tested with the ND filter insertion technique we discussed on Wednesday.

  1177   Fri Dec 5 12:50:24 2014 DmassNotesGeneraldBm conversion table

The 4395 saved data in dBm/Hz while displaying Vrms/rtHz, so I had to figure out the conversion factor, and found the following useful table:



  1176   Thu Dec 4 19:52:11 2014 nicolasElectronicsGeneralUsing the Zurich HF2 built in PLL

The Zurich Instruments HF2 has a very nice built in PLL feature. You give it a sine wave up to 50MHz and it will lock a PLL on it, and it will give the control signal at one of the outputs.

It has a nice PLL design interface (first attachment) (the design can be compared to the measured closed loop gain in the second attachment, this doesn’t include the marconi frequency modulation calibration). You tell it the bandwidth you want and it can internally set it’s PID to achieve that, and it will complain if you are asking it for too much. It claims to be able to get a 50kHz UGF, but I was only reliably able to get like 20kHz.

So right off the bat, this won’t be what we want if our goal is a very high BW PLL to suppress whatever nonlinear noise mechanism Rana and Dmass are worried about.

However, if 20kHz BW is enough, then this might be a pretty nice PLL to use. With a 1MHz modulation range, 20kHz PLL BW, and a 20MHz carrier, the noise is given in the last attachment. (below 1mHz/rtHz up to 1kHz, then starts to rise like f until it hits the PLL bandwidth, and it rolls off again.) The red trace is the spectrum analyzer noise.

data and scripts are all here.

Attachment 1: 45.png
Attachment 2: PLLCLG.pdf
Attachment 3: PLLNoise.pdf
  1175   Thu Dec 4 02:45:49 2014 DmassNoise HuntingNoise BudgetNoise budget cleaning up

PDH sensing noise:


New lower sensing noise with addition of transformers between LO and RF in both paths.

Cavity unlocked (detuned with temp) noise shows the scatter bump - unsure if this is interesting or should be totally obvious




LBservo noise:



The LB input noise gets to a minimum of 40 nV/rtHz if we turn the gain 100% up at high freq. This is 4x over what they claim

(shown it is 50 nV/rtHz)


Data is all on svn and will be put into noisebudget update

  1174   Wed Dec 3 17:02:31 2014 DmassNoise HuntingNoise BudgetNoise budget cleaning up

Looked at noisebudget on SVN and some of the measurements are dated / no loner applicable:

Update Noisebudget:

  • PDH sensing noise with transformers in demod chain
  • LB1005 input referred noise with low pass + higher gain setting
  • Current driver output noise (measured across resistive load with Rai's low noise preamp)
  • PDH loops E+W
  • Transmitted RIN E+W (potentially new minima)

Take and compare:

  • Err and control for E+W PDH past 100 kHz (and scope RMS / peak to peak vs integrated RMS of PSD)


  1173   Wed Dec 3 15:43:15 2014 DmassNoise HuntingMeetingsMeeting notes

ISS - what Koji has done with ISS/FSS would be good to copy - sum in DC offset at end of ISS box -

Stefan - input to last stage - change + pin to offset

Check out Nic's old elog after recp't of NEOS drivers - want [ V_rf / V_aud ]

Noise to back of AOM driver (SR785 -> SR560), measure RIN, measure beat, see beat increase


Check out Evan's median subtraction algo (to estimate coupling of RIN) - this takes median of spectra


RIN on transmission - coherent or not?

ND filter before PBS - increase power by X, decrease using ND filter


Isolators good? These are untested-ish - do we want translation stages for these?

Coherent white noise added into error point?

High freq noise at monitor PDs + low PD bandwidth (what PDs are we using / do we see the high RIN on the beat PD)?

Birefringence in coatings leading to multiple peaks with overlapping PDH error signals?


Get FTB-1-1 minicircuits transformers - skip RF grounding for now


New PLL scheme - mix beat down into low band

Zurich PLL?

Other servo boxes if no zurich: Old uPDH box? PDHv1? Misc NIM servo boards?

AD829 xover with SR560 for PLL?


PLOT Look at PLL RMS when beat, make PLL rms go down

  1172   Tue Dec 2 22:51:52 2014 DmassNoise HuntingNoise BudgetBeat spectrum not ltd by

Talked to Rana about the increase in sensing noise due to DC grounding in the PDH demod electronics (we were plugging the Gold PD directly into the mixer- he confirmed that we want transformers here)

He agreed that we want to not make monstrously sized ground loops and should use transformers for DC isolation like so:


I borrowed a pair of "balanced to unbalanced" 1:1 transformers in the PSL/CTN lab and stuck them in the readout chain - the noise went down.

I tried all four permutations of balanced / unbalanced on both the LO and RF inputs of the mixer - I saw no difference in noise between these.

There is no grounding done at RF in these mixers (I opened and looked). I have seen grounding put into the transformers in other demod chains.


Questions I need answered:

  • What kind of transformers do we want to buy for this?

  • What are the consequences of not grounding / grounding the transformer at RF (say with a 4 nF cap across the ground which gives => 1 ohm at 33 MHz)


Slightly downstream of this I (previously) made another change in the setup: here is a picture of what I added between the LB1005 box and the driver:



The LB1005 is a P-I with crossover at 30 kHz (near the cav pole)

The RC filter between the driver and LB acts as 40 dB of low frequency boost

Measurements/plots wanted:

  • Sensing noise for each path with transformers in play
  • LB1005 + atten input ref noise (I believe the atten doesn't change this)
  • *also need: in loop error signal + ctrl sig noise out to high freq
  1171   Tue Dec 2 16:12:18 2014 DmassUpdateRoadmapThe week's measurements

The following measurements are being done this week:


Photothermal transfer function at 120K

Wilcoxen seismometer (accelerometer?) added to setup

Cold seismic noise (boiling noise)

Cold temp noise

Cold beat w/ Weiner subtraction of PDH** / intensity / seismic

**PDH error signal f-domain Weiner subtraction @ low gain regime

  1170   Tue Dec 2 16:05:06 2014 DmassNoise HuntingCryo ISSISS loops almost work

[Dmass, Nic]

We resurrected the intensity loops on Friday. They almost work. 

Why is it difficult?

  • The AOM driver we have is only linear over a narrow range of input voltages, and requires an offset:
  • Approximately 500 mV linear range with offsets of +/- 600 mV (the driver is symmetric and does roughly the same thing with + or - inputs at the back
  • Outside of these regions we see major harmonic distortion / clipping
  • The ISS servo board has a fixed input offset (the voltage on transmission PD which it tries to to via the AOM) - can tune the power / make a voltage divider
  • The ISS servo board does NOT have any way of independently tuning the output offset
  • We resorted to fine tuning the power at transmission via the input power in order to set the offset to exactly what we wanted:
  • We drove the AOM driver in its linear range with the cavities locked, plugged the transmission into the ISS servo box, and looked at the signal coming out of the ISS servo box - we tuned it close to the correct offset for the input of the AOM driver (with a negative sign and lots of gain) by eye/scope
  • We closed the loop and gently fiddled with the input power, and were able to lock the ISS
  • IT WAS EXTREMELY FINNICKY (re: there was a very narrow range of input power that it would lock for, the spectrum was nonstationary/bouncy, and it would break after a minute or so at max)
  • We have ~5% fluctuation in RIN at transmission due to the platform swinging
  • We could re-lock by fine tuning the input power again and get another couple minutes.
  • The boosts worked.

Does it work?

  • Almost / sort of. We could get fast measurements done with it, but it will require constant babysitting and quick spectra collection
  • It does not have enough range to make it into the "healthily working" category.


  1169   Tue Nov 25 23:11:41 2014 DmassNoise HuntingCryo ISSAnomalous intensity noise

We still have the strangely large intensity noise in both paths.


 The lowest transmitted RIN we have measured in lab is:

  • White up to the cavity pole
  • 1/f after the cavity pole
  • few times 1e-5
  • Decades higher than the intensity noise on the beam measured simultaneously right before the crytostat, via a pickoff mirror

A plot of the [RIN/Hz] transfer function as a function of offset from cavity resonance is below -

Maximum possible value is 2.5e-5 RIN/Hz

When we have tuned offsets to try to make this go away (in the past), we have not been crazy offset. I think a reasonable estimate based on this, and transmitted power optimization is < 0.5 RIN/Hz

If we had a monstrous offset (~40 kHz), the frequency noise required to produce this is ~1Hz/rtHz, which is well in excess of our previous best beat

=> the intensity noise measured at transmission is not frequency noise + offset

Nic and I are turning on the ISS and taking a beat tomorrow, to see if Intensity noise -> frequency noise coupling is currently limiting (photothermal)


Musings -

Would there be any way to get white jitter noise which couples into intracavity power but not into the input PDs?

We discovered that we can effect the intensity noise with the half-wave plate right before the cryostat input window - the L/4 plate next to it has no effect.

I would suspect laser feedback**, but measured frequency noise at transmission, and laser output intensity noise are both too low, unless it could somehow be laser feedback induced jitter which looked white up to high frequency?


** because it is so easy to mode hop the lasers even with Faraday's at their output



Attachment 1: CavHz2RIN.png
  1168   Tue Nov 25 22:45:43 2014 DmassNoise HuntingNoise BudgetNew beat spectrum:

I was artificially limiting my PLL range by directly driving the Marconi with the IF output of the mixer - this restricted it to ~500mV range.

The input range of the Marconi is (measured to be) +/- 2V (~ 8x higher than the mixer output I was using. Free-ish lunch here.


We might have to wait until we're cold to have low enough drift to use the Marconi in any truly low noise way, but I believe it's no longer limiting at even room temperatures. Beat noise spectra tomorrow.

  1167   Tue Nov 25 16:06:55 2014 DmassThings to BuyPDHEOM response

I replicated Koji's impedance measurement setup for those-who-cannot-afford-an-impedance-mesaurement-kit (a la this elog), but did not see any large dips in the impedance from mechanical features. I guess this is not so surprising, as the coupling of acoustic modes into impedance isn't necessarily large.

After discussing with Chris, we decided to misalign the polarisation into the EOM and measure the transfer function of Volts -> RIN.

Attached are plots of [RIN/V] through the EOM - the magnitude fluctuates with the polarization misalignment, but the phase remained consistent.

Using this as a proxy for [rad / V], it is unclear which one would be easier to use as a high frequency path - The West EOM has a much higher Q single resonance, and larger phase excursion, but we might be able to correct for it electronically, whereas the East EOM has larger more frequent phase excursions at lower frequency.

REGARDLESS, both of these EOMs are in use for generating sidebands, and if we want to have a high frequency phase correction path, we need to purchase two more EOMs**



** we explored using a bias T at the laser to switch to laser current mod produced sidebands to free up the EOMs and save $$, but this made the laser noise go up and we were unable to track down the culprit over a couple days. We also began to explore making a circuit which had resonant gain and DC coupling so that we could use one EOM for both paths, but we found little enough evidence that this was efficient (but it is interesting), and thus abandoned it.

Attachment 1: EOM_V2RINcompare_full.png
Attachment 2: EOM_V2RINcompare_zoom.png
  1166   Wed Nov 19 14:54:36 2014 nicolasNoise HuntingNoise BudgetISS servo fail?


I tried to implement the ISS servo today for the east path with little to no success

It it not clear to me that the servo was engineered with the actual driver in mind

The AOM driver is only linear over a narrow input range: [156mV to 450 mV]  (can be + or -)

The ISS board seems to go to not have this offset built in, and it is not clear if the excitation inputs would be an appropriate place to add this.

I played around with using terminators on the input and output signals to try to force the DC value at the output of the ISS board to the middle of the linear range, but could not get the loop to do any actual intensity noise suppression for any switching state on the front

I punt to Nic, interim king of ISS servo board, following the passing of Prince Chaz, rightful heir to the throne, to determine what the correct way to sum an offset into the output of the board is.

 It's not pretty, but the servo will create the output offset internally because of the low frequency boost and integrator. When you close the loop, it will push the driver into the linear range.

  1165   Mon Nov 3 20:36:14 2014 DmassNoise HuntingNoise BudgetNew beat spectrum:

Noise floor is marconi noise in PLL (see elog:1049)

Explored anomalous intensity noise with Chris. Were able to get sizable decrease in the noise by tuning the L/2 plates on the input beam.

No change to beat spectrum level.


With both temperature loops engaged, beat drift rate is 400Hz/sec.

RMS frequency noise from PDH loops is ~200Hzrms - mostly from Fourier frequencies 100kHz to 500kHz


Might be able to get away with 2kHz Marconi range if we take care of the drift rate => this problem might disappear @ 120K because of the zero crossing


Would need a 400 Hz marconi range (assuming no other noise pops up) to see coating thermal noise @ 1kHz


What noise/range did the DFD box from the 40m end up hitting?

  1164   Mon Nov 3 20:27:28 2014 DmassNoise HuntingNoise BudgetISS servo fail?

I tried to implement the ISS servo today for the east path with little to no success

It it not clear to me that the servo was engineered with the actual driver in mind

The AOM driver is only linear over a narrow input range: [156mV to 450 mV]  (can be + or -)

The ISS board seems to go to not have this offset built in, and it is not clear if the excitation inputs would be an appropriate place to add this.

I played around with using terminators on the input and output signals to try to force the DC value at the output of the ISS board to the middle of the linear range, but could not get the loop to do any actual intensity noise suppression for any switching state on the front

I punt to Nic, interim king of ISS servo board, following the passing of Prince Chaz, rightful heir to the throne, to determine what the correct way to sum an offset into the output of the board is.

  1163   Mon Nov 3 16:15:21 2014 DmassNoise HuntingNoise BudgetBeat spectrum not ltd by

I looked at the sensing noise (which goes WAY up when the ground of the IF port of the mixer touches the ground of the servo box - from 13 nV to 100 nV)

Sensing noise, input referred servo box noise, and in loop err signal noise are ALL below the current beat limit ( ~ 7e-2 Hz/rtHz around 1kHz ) by ~5x


We should figure out what this noise floor is.

  1162   Mon Nov 3 15:16:28 2014 DmassNoise HuntingNoise BudgetNew beat spectrum:

I set the beat back up and locked both loops with new LB1005 + zero-pole between LB and driver topology

Tuned and measured coupling, RAM offset, and modulation depends for both paths.


  • P_E = 1e-3 W
  • Gamma_E = 2*sqrt(0.100/2.04) = 0.44
  • Coup_E = 51.2/64 = 0.8
  • Z_E = 600 V/W
  • Err Slope = [ 2 * P * Gamma * sqrt(coup) / f_cav] * [Z_E] = [ 2*1e-3*0.44*sqrt(0.8)/40e3 W/Hz ] * [600 V / W] = 1.26e-5 V/Hz


  • P_W = 1e-3 W
  • Gamma_W = 2*sqrt(0.084/1.40) = 0.49
  • Coup_E = 67/91.2 = 73.5
  • Z_E = 400 V/W
  • Err Slope = [ 2 * P * Gamma * sqrt(coup) / f_cav] * [Z_E] = [ 2*1e-3*0.49*sqrt(0.735)/40e3 W/Hz ] * [400 V / W] = 8.4e-6 V/Hz

Locked both paths and tuned gain to minimize error signal RMS

Measured error point of both paths w/ SR785 (see below) - divided by error slope for calibration

Both signals ~100 nVrms/rtHz at floor


Set PLL back up - used level 10 mixer, with amp on beat signal so that beat RF level = +6 dB

Measured beat at input to marconi for a couple different ranges

PLL *not* divided out:

** PLL UGF(s)***

  • Vpp = 672 mV
  • discriminant = [0.336 V / rad]
  • 10 kHz marconi range: 10 kHz/1.41V
  • loop = [ 0.336 V / 1 rad ] * [ 10e3 Hz / 1.41 V ] * [ 1 rad / f Hz] => UGF =  2.38 kHz
  • 20 kHz marconi range: 10 kHz/1.41V
  • loop = [ 0.336 V / 1 rad ] * [ 20e3 Hz / 1.41 V ] * [ 1 rad / f Hz] => UGF =  4.75 kHz


Attachment 1: ErrSigs.png
Attachment 2: BeatNz.png
  1161   Fri Oct 31 16:12:18 2014 DmassNoise HuntingNoise BudgetErr signal noise too high

East in loop error signal too high - coherent with transmitted intensity noise - incoherent with input intensity noise - above measured input referred noise of LB1005.

Control / Error < measured transfer function between these two places

More info as I investigate.

  1160   Fri Oct 31 01:03:25 2014 DmassNoise HuntingNoise BudgetEast loop locked with new gain distribution

Locked with new gain distribution (zero pole pair between output of servo and driver) - UGF of 200kHz achieved (tuned by minimizing RMS of error signal),

error signal RMS = 52 mVrms

Input referred noise of LB1005 measured in this loop state and calibrated

Noise low enough to merit operation in this state ~ 20x lower than previous best ASD measurement in Hz/rtHz

Attachment 1: LB1005PDHOLTF.png
Attachment 2: LB1005inputNoise.png
  1159   Thu Oct 30 23:22:14 2014 DmassNoise HuntingNoise BudgetCurrent Driver Noise Measured

Finally measured the current noise coming out of the drivers using the "Rai Weiss Low Noise preamp"

Measurement limited by johnson noise of load resistor (20 Ohms)

Good measurement up to ~200 kHz - subtracted the noise floor in quadrature (hence noisy above 200 kHz)

Looking at elog:1126, we can see this is smaller than the free running laser noise (which is ~30 Hz/rtHz down to 10 kHz, and goes up as 1/sqrt(f) below that)

The peak around 100 kHz makes the free running noise increase by ~10% (in units of ASD) - it corresponds to ~3 Hz/rtHz laser noise

Attachment 1: WestCurrentNoise.png
  1158   Thu Oct 30 14:31:14 2014 DmassElectronicsGeneralLB1005 noise

LB1005 input noise is naturally 42 nV/rtHz at high freq - to get 10 nV/rtHz at high frequency if we add a x4 preamp, and have the gain knob set to +7.0 - the noise starts going up around 1 kHz



LB1005 testing:

The LB box had excessively nigh input noise compared to its spec (42 nV/rtHz minimum measured vs 10 nV/rtHz minimum spec).

I checked out both the 40m and the OMC lab LB boxes to see if one was broken, and Chris

I tested the input referred noise as measured by the output in a variety of states, including the input offset on/off/at different values and the integrator on/off (had to build a low UGF loop and look at out of band noise for this)

Chris & I opened one up to look at it, and it looks like the 40nV noise is after one of the first couple op-amps in the circuit

As an additional clue, Stefan had complained that the LB1005 box noise was too high (interpretation of this data point left as an exercise to the studious elog reader)

  1157   Wed Oct 29 03:44:21 2014 DmassElectronicsSchematicsCurrent Driver Mod

The current "i_mon" path is totally worthless and only monitors our DC current level

I propose the following change to the current driver board in order to repurpose the current monitor path into an actual buffered control signal monitor (lower left quadrant of schematic)

  • Remove U1
  • R3: 1K -> 0
  • R1: 1k -> 50
  • lift pin 3 of U2 (+ input) and fly small jumper wire to modulation input (center of BNC connector P2)

This gives us a buffered 50 ohm output for monitoring the control signal and taking transfer functions / looking at noise without modifying our loading (now more relevant because of our passive network to modify gain)

The AD8671 should be fine with a 100 ohm load (50+50) - the RMS of the control signal is small at that point (~7mVrms), and the temperature loops should keep the DC value in check

The AD8671 datasheet (here) says it can be shorted indefinitely, so the brief current saturation during lock acquisition should be ok (the DC values during lock are ~5 V, which is 50 mA with a 100 ohm load)

Attachment 1: LD_Driver.pdf
  1156   Tue Oct 28 15:26:24 2014 DmassElectronicsGeneralTTFSS summary

As the options for TTFSS seemed to be:

Ways we could use the TTFSS:

1) Turn down the gain knobs, remove both feedback path poles, shape loop with passive network at output

This is the same route we were going with the LB1005 box - there is no reason to expect the noise will be good if we attenuate and multiply this much (AD8602s?) at the input - would have to change board topology

If we want to use a passive zero-pole at the output of the box, we toss AC range - if we use the AC range requirements set in elog:1150, and make a pole at 40 kHz, we only have 45 dB of loop gain at DC, which is a bit low for comfort


2) Turn down gain knobs, remove *some* feedback path poles, use pockel's cell path, shape accordingly for high frequency x-over

Need to buy 2 more EOMs and add them into path - would still need to rework PZT path -> current driver because shape / gain are so different


3) Find place to add bypass path in parallel like we did in early generation of PDH board (make P-I x-over with a DC coupled bypass path) - do some whitening (gain/range distribution) with passive network between driver and servo board

This is getting into overly hacky territory - not comfortable cutting up the 40m board, and the TTFSS is too far from what we want to justify buying one to mod it


4) Use its phase corrector path while using some other servo for the laser current (this seems very reasonable) - would either use the LB1005 or new generation of PDH boards built in collaboration with W.Z.K.

This seems very reasonable


We need a better understanding of the LB1005 noise / range / AC range requirements of the servo - see elog:1156

  1155   Mon Oct 27 14:00:56 2014 ChrisElectronicsGeneralTTFSS checked out, and shelved for now

Dmass and I borrowed a spare TTFSS and interface chassis from the 40m lab, and got them working in the shop.

We briefly flirted with using the PZT path as part of our frequency servo, but we were soon disenchanted by the circuit's lack of flexibility. For the time being, we still need the option of running a standalone frequency loop, with high bandwidth. The PZT path is ill suited to this. It puts two poles near 40 kHz, where we'd want a zero to compensate the cavity pole. And its low frequency pole is implemented as a passive low-pass on the output, cutting down the range at high frequency. [Dmass can elaborate on the AC range requirements for this loop.]

Those features are there to make the crossover with the EOM path, while coping with a set of issues our laser's frequency actuator doesn't have: the resonances and capacitive load of a PZT. We could hack around them -- but we'd be hacking a board that has no generic or switchable filter stages, a step backward from the status quo (LB1005 or PDH board). We suspect better alternatives are out there.

We did get far enough along to check the PZT path TF against the TTFSS matlab model (available here). The TF shape was predicted well, but the model had an extra 14dB of gain not found in our measurements. We found out why: the model didn't account for the dividers formed by R7 and R34 with the 100 ohm input impedance of the AD602s. A corrected version is attached to this entry.

Attachment 1: getTTFSS.m
% [hOL, hEOM, hPZT] = getTFs(param)
% Example (parameter version 3):
% [hOL, hCross, hEOM, hPZT, info] = getTTFSS(paramTTFSS(3, f), f);
% zplotlog(f, hOL);

function [hOL, hCross, hEOM, hPZT, info] = getTTFSS(param, f)
  % s-domain
  f = f(:);
... 115 more lines ...
Attachment 2: paramTTFSScryo.m
% param = paramTTFSS(ver)

function param = paramTTFSScryo(ver, f)
  if nargin < 1
    ver = 0;
  if nargin > 1
    % s-domain
    f = f(:);
... 167 more lines ...
  1154   Wed Oct 22 20:32:27 2014 DmassElectronicsPDHGold PD Sensing Noise

I couldn't find a plot of post demodulation sensing noise for either of the gold PDH PDs

Here is the sensing noise after the LPF for the east gold PD + mixer chain (output of mixer has a 50 ohm terminator T-ed into it, and uses a minicircuits 5 MHz LPF)

The RMS Voltage noise is somewhat high


If I want to calculate the offset from cavity resonance due to the RMS sensing noise (i.e. the ASD shown below) should I divide out the cavity pole or not? aka: pick algorithm 1 or 2 below:

1) Divide ASD (Vrms/rtHz) by plot 3 in elog:1126, then by plot 2, then calculate the cumulative rms (this seems like it might be wrong because it would give me a high frequency slope of f before I tookt he RMS, and white sensing noise would swamp the rms if we include enough bandwidth)

2) Divide ASD (Vrms/rtHz) by plot 3 in elog:1126, then by the DC value of plot 2, then take the cumulative RMS (these calibrations commute since they are both frequency independent factors, and could just be multiplied by the RMS shown below in red to give the offset in Hz)

Attachment 1: EastSensingNoise.png
  1153   Mon Oct 20 20:16:17 2014 ranaCryostatSchematicsCryostat Windows


This is the drawing I am sending around. The idea would be to get a company to make an adapter so that we can have CVI optics for 1550 nm and Viton in the outside window.

  1152   Fri Oct 17 19:15:35 2014 DmassElectronicsPDHActuation Range Semisolved


Made the above to decrease the PDH board's (currently LB1005) input referred noise without losing so much range that we cannot lock / stay locked

The blue box does: the following when plugged into the current driver:

  • 0 dB DC gain, -20 dB AC gain, pole at 7 Hz, zero at 70 Hz

Swapping the box in and out did exactly what I expect (allowed me to put 20 dB more gain into the LB1005 box)



I didn't get all the gain increase I was expecting - I was only able to increase the gain of the LB by 8 dB; it seems the gain of the loop changed by +10 dB between Tuesday and now. 

  • With a total OLTF gain of +11 dB at 100kHz, the loop did not oscillate (as measured by seeing the noise increase / control signal increase)


  • +10 dB loop gain @ 100kHz => LB gain with Pomona box inserted: knob = 5.56 / AC gain = +3.36 dB
  • +10 dB loop gain @ 100kHz => LB gain with no Pomona box: knob = 4.25 / AC gain = -12.8 dB ** (+1.0 on knob = +12.5 dB)


  • +10 dB loop gain @ 100kHz => LB gain with no Pomona box: knob = 4.93 / AC gain = -4.4 dB

There is +10 dB extra gain somewhere outside the LBbox compared to the gain on Tuesday


  1. Total power: 1.4 mW (higher by 40%, or +3 dB)
  2. Sideband transmission peaks: Vcar = 1.67V // Vsb = 68 mV => gamma = 0.40 - same as Tuesday)
  3. LD Driver transfer function: still 1 mA/V as measured by the SR785 across a 1 Ohm load (e.g. swapping R18 from 500 to 11k did not change its overall transfer function)
  4. Peak to Peak of error signal = 316 Vpp (similar to what was measured on Tuesday
  5. Turned everything off then back on to see if we got into some funny state

This leaves +7 dB unaccounted for, and no known 50 ohm mistakes to solve the mystery

This is mildly disconcerting, but not horrifying. We should periodically check to see if the plant + sensing gain is drifting around. If it stays at this value, I will just assume I made some mistake in loading (maybe an extra 50 ohm terminator T'ed in somehow)

The attenuator idea works.

The Input referred noise of the LB1005 is still pretty high and shouldn't be hard to beat with the TTFSS board - as soon as we prove this is the case we should get 2 TTFSS boards.

  • Input Noise(10 Hz) = 1.35e-9 Vrms/rtHz
  • Input Noise(100 Hz) = 510e-9 Vrms/rtHz
  • Input Noise(1 kHz) = 183e-9 Vrms/rtHz
  • Input Noise(10 kHz) = 115e-9 Vrms/rtHz
  • Input Noise(100 kHz) = 137e-9 Vrms/rtHz


  1151   Thu Oct 16 00:44:25 2014 DmassElectronicsPDHActuation Range Needed

After talking with Rana:

We should be able to lower the AC gain without lowering the DC gain (and thus sacrificing our range) by:

changing R18 in the laser driver to 47 ohms

adding a pomona box between the LB box and the driver with a 450 ohm resistor in series and a 47 uF capacitor in parallel.

I have some nice 47 uF caps in hand.

  1150   Wed Oct 15 23:05:30 2014 DmassElectronicsPDHActuation Range Needed

Summary of East Actuation Range Requirements:

  • Control signal range = +/- 7V
  • Control signal RMS = 7mVrms
  • Control signal full peak to peak for rare noise spikes = 47 mVpp
  • Control signal drift rate = 20 mV/min
  • Control signal locking transients huge - take up large fraction of range - need to address this before we redistribute any gain - see attached


We didn't actually know how much range we needed for the PDH boards

Changed driver back to 1mA/V

Realigned east path 00 mode

Tuned input power (as measured at Refl PD) to 1mW

Tuned waveplates and got 0.78 coupling

Swept cavity via driver input (10 Vpp = 10 mA now that I reverted to old resistor values)

  • Checked sidebands via transmission peak - meaured gamma = 0.49 initially, then down to 0.40 once I turned down total power and remeasured - using gamma = 0.4 as I expect it's a more realistic representation of our steady state operating point

Checked demod phase by symmetry of sideband error signals - was nearly correct

Plant = 2/fcav*sqrt(coupling)*gamma*Pin = 1.8e-8 W/Hz

RFPD = 600 V/W (trusting elog:1126)

  • => error signal slope = 1e-6 V/Hz

I used the LB1005 to lock the loop (t was very easy) - gain knob = 4.25

The error signal RMS was less than 1/3 the peak to peak of the swept out discriminant

I looked at the control signal at various time scales on a scope and monitored its drift rate and RMS:

  • Vrms of control signal at scope = 6.7mVrms
  • The largest excursion at any time scale I checked on the scope was 47mVpp
  • I measured the drift rate of the control signal - it was ~20 mV per minute over 7 minutes (\approx 6 MHz/min according to elog:1126)

I quickly checked the loop gain with the SR785 to make sure we weren't in some abnormally low gain state - the gain @100kHz was 10dB, and it was falling smoothly as 1/f up to that point


I then checked the range of the control signal, using the error signal spectrum as measured on a SR785 and the transmitted power to diagnose the relative health of the loop as the control signal drifted

  • The noise started to increase at +6.16V and -7.25V
  • Lock broke at 8V and -7.5V
  • We can thus treat the output range of the LB1005 as -7 to +6 Volts

I then tried to measure the locking transients (see attached pictures)

The large smooth part of the control signal transient is presumably due to clunkiness in the knob on the front of the ITC510 (with it's absurdly large Kelvin/Radians coefficient)

New drivers or a cymac based temperature knob via the temp tune input on the back of the ITC510 could solve this. This needs to be addressed if we want to unload ANY gain from the current driver, or we won't be able to lock via the temperature knob, since these control signal swings take up most of the actuation range measured above

I don't know what determines the amplitude of the discontinuous jump at the moment of lock acquisition - is this inherent to the system in some way, or can we make this smaller? They are sizable compared to the range, which bodes poorly for unloading actuator strength

If we can crush the transients somehow we can unload gain from the laser driver and win in the noise budget

Attachment 1: EastTransients.png
  1149   Tue Oct 14 21:38:51 2014 DmassElectronicsPDHPDHv2 upgrade mark 1

Driver input range much larger than output range of servo board in any condition

LB1005 noise too high to be useful 

Simulated current driver in LTspice - it has ridiculous input range because of its topology - (we would need to drive HV at the input to blow it)

We can keep turning down the gain by increasing the resistors all the way up to 100k (10 uA/V) and only lose by the output range of the PDH servo board times the gain of the driver

I unloaded this gain (the 10x) into the PDH servo board did a quick check to answer the "how much can we get out of the LB1005 box?" question

LB1005 settings:

  • gain knob = 4.91 (-6 dB at high freq)
  • P-I corner at 30 kHz

Looking at elog:1126, the generous error signal slope is 8.3e4 Hz/V


input noise [Vrms/rtHz] input noise [Hz/rtHz]
10 4.4e-6 0.37
100 1.3e-6 0.11
1e3 4e-7 0.03
1e4 2.8e-7 0.023
1e5 4e-7 0.03


This is about 20x the predicted coating thermal noise level

I believe this is coming from the output, and we could win more by unloading more gain into the servo board, but past labwork tells me we need ~1MHz range, so that's out.

  1148   Tue Oct 7 19:34:51 2014 ranaElectronicsSchematicsOscillator Box concept for making PDH LO and RF
mind mapping software
  1147   Mon Oct 6 20:40:25 2014 DmassElectronicsPDHPDHv2 upgrade mark 1

current driver coefficient changed from 1 mA/V to 0.084 mA/V for both paths


I tried to modify the current driver to turn down its [mA/V] coefficient so that we didn't have such an absurd Hz/V actuator coefficient

I changed R14,15,16,17,19,20,21,22,23 from 1K to 11K on both paths (based on a previous conversation with Rich)

I put a 1 Ohm load across the output of the driver, and measured the response through the driver (both with a 4395 and a Scope)

As expected, the loading (50 Ohms or Hi Z) made little difference since it is in parallel with 1 Ohm

As desired, the response got much lower

Old response: 1mA/V

New response: 0.084 mA/V

I am unsure exactly if/how I have modified the range of the current driver with this change, so I emailed Rich and asked for the spice model we have used for this previously rather than repeat that work

DCC link to current driver for the curious/forgetful/human (here)

  1146   Wed Oct 1 02:53:44 2014 ranaUpdateSuspensionNew Suspension Investigation


 I think it should be fine as shown. The first couple of modes are the actual translational ones.

The Mode 8 shape, to me, looks like one where the little mass is moving by itself, but the space and support are not. As long as this is not coincident with the spacer body modes it shouldn't matter too much.

You might try looking into using some SS, but I don't imagine there's too much difference. If the shop has one of the kinds of Spring Steel in stock, that's a good option for flexures (since we're not too worried about UHV specs):


  1145   Tue Sep 30 12:15:03 2014 DmassUpdateSuspensionNew Suspension Investigation

SUS thoughts:

COMSOL model might be incorrectly estimating frequencies because dummy load acted as inertial mass, should just model cavity on supports.

Do we use gravity? Things which adding gravtity to COMSOL gives us:

  • Gravitational dilution (easy to order of magnitude calculate)
  • Nonlinearity in Young's Mod from stress
  • Contact area based on gravity + contact mechanics (see elog:1143) - DOES COMSOL DO THIS CORRECTLY?


Simulating the cavity on supports and looking at the ratio of strain energy density in the cavity vs the supports answers:

"If we are isolating the cavity well enough so that most of the damping at the body mode comes from the support and the cavity itself, then the strain energy density ratio times the mechanical loss factors will predict the correct Q"

However, if we are not isolating the cavity sufficiently well, and some of the energy/cycle is leaking out into the outlying structure (or the mechanical screw joints between structures), then COMSOL will predict an incorrect Q. We want to answer in some quantifiable fashion:

"How well are we isolating the body mode and supporting a high Q with our support scheme?"

We could just add mode layers of support structure to the model, but we don't believe we can correctly model the screw joints. I proposed the following recipe to do it all in 1 simulation



  • Make cavity on supports in COMSOL (with no phononic bandgap action), with supports constrained at bottom
  • Eigenfrequency analysis
  • X(f) = Volume integral of strain in body of support at frequency of interesting body mode, under where we want to make the cut for bandgap
  • Make cuts in supports (separate model?), otherwise identical to above model
  • Eigenfrequency analysis
  • Y(f) = Volume integral of strain in body of support at frequency of interesting body mode, at same place as above
  • H(f) = Y(f)/X(f) = the transfer function which tells us how much isolation we have bought with our support scheme, and has a very simple intuitive interpretation
  • Strain energy density ratio predicts Q more correctly as H(f = body mode) => 0



  1144   Tue Sep 30 04:58:32 2014 DmassUpdateSuspensionNew Suspension Investigation

I investigated the following geometry in COMSOL:


The square load the top is an ultra dense near infinite young's modulus material (user specified) meant to simulate how the cavity interacts with the support node

This is what it looks like under load (as usual, COMSOL's deformation is not-to-scale):


I changed around the width of the material left, the thickness of the cut, and the height of the cut in MATLAB, to generate the attached plots

The mode shapes are as follows:


If this accurately simulates the eigenfrequency of the cavity under load, it seems we will be in trouble because of mode 8 - none of the recomended cuts pushed this to an eigenfrequency which was lower than the body mode, which seems to imply DC translational force coupling.

If we believe the simulation, we can:

Choose different material (want high density, lowish young's modulus, high yield strain)

Choose different geometry (likely something for a new generation of the experiment/student)

Attachment 3: efreqwitdh.png
Attachment 4: efreqthick.png
Attachment 5: efreqheight.png
  1143   Tue Sep 23 02:39:52 2014 DmassUpdateSuspensionSus update - cavity body mode

The cavity drum mode on the PEEK supports looked like this:


I constrained the bottom half of the support points, and found the eigenfrequencies'

The drum mode (pictured above) is at f = 37190 Hz

N.B. I didnt have the windows or vent hole in the model, but the frequency agreed fairly well with experiment and evans previous (support free) result.

I don't have a good number for loss of the PEEK parts, so I am unsure if this is wholly responsible for our low Q. It would be interesting to plug in the loss numbers (by exporting the 3D shape to MATLAB and following the strain energy density ratio times loss formula)


Contact Area:

Another thing we didn't previously have good numbers for was contact area.

These are analytically solved problems (see Wikipedia's contact mechanics page)

I calculated the relevant contact areas to be:

  • For the old PEEK supports, the contact area is 2.2e-5 m^2
  • For the new Al (optimally constrained) supports, the contact area is 5.0e-5 m^2


Next up: Silicon cavity with new Al 6061 supports COMSOL model


Area of most confusion: the Russians commented that the best way to support a high Q mode was to put something very soft (e.g. gold foil) between the cavity and the support node. I am unsure how to properly model this is COMSOL / properly say what it will do.

The question: if I take a rigid body (Si cavity), place it on supports which have little/no mechanical response below the mode we're interested in (drum mode), and slip a thin layer of gold between those two, what happens to the transfer function between the cavity and the support?

  1142   Mon Sep 22 19:27:25 2014 DmassUpdateSuspensionSus update

We were holding the cavities in a very dumb way. We will now hold them in a less dumb way.

The original cavity supports look like this:


The axis of the support is parallel to the axis of the cavity, which greatly overconstrained how we were holding the cavity (4 line contacts)

The new cavity support looks like this


and is optimally constrained (4 point contacts on the bottom of a cylinder)

Since this is what the cavity sits on, it would be bad if there were accidental resonances at the cavity body mode frequency.

Here is a plot of the eigenfrequencies of each support:


The modes around 0 Hz are all artifacts of the COMSOL simulation (translational body modes from not being constrained)

The PEEK support has a mode relatively close to the cavity body mode:


 The equivalent mode in the 6061 aluminum support is:


 For reference, the lowest frequency mode in the new standoffs is:


And seems very decoupled from longitudinal motion of the cavity.


Relevant material properties:

  • Young's Modulus:
    • PEEK (300K) = 3.6 GPa (http://en.wikipedia.org/wiki/PEEK)
    • Al 6061 (300K) = 69 GPa (http://cryogenics.nist.gov/MPropsMAY/6061%20Aluminum/AL-6061-T6_Plots/AL_6061-T6_Youngs.JPG)
    • Al 6061 (120K) = 76 GPa (http://cryogenics.nist.gov/MPropsMAY/6061%20Aluminum/AL-6061-T6_Plots/AL_6061-T6_Youngs.JPG)
  • Poisson's Ratio:
    • PEEK (300K) = 0.38 (http://www.azom.com/properties.aspx?ArticleID=1882)
    • Al 6061 (300K) = 0.33 (http://asm.matweb.com/search/SpecificMaterial.asp?bassnum=MA6061t6)
  • Density
    • PEEK (300K) = 1320 kg/m^3 (http://en.wikipedia.org/wiki/PEEK)
    • Al 6061 (2700 kg/m^3 (http://en.wikipedia.org/wiki/6061_aluminium_alloy)
  • Quality factor:
    • Al 6061 (300K) = 2e5 (Bar detector paper: http://www.sciencedirect.com/science/article/pii/S0011227502000218 - figure 4)
    • Al 6061 (120K) = 3-4e5 (http://www.sciencedirect.com/science/article/pii/S0011227502000218 - figure 4)
    • PEEK (300K) = NEED NUMBER/SOURCE - no luck finding yet
    • PEEK (120K) = NEED NUMBER/SOURCE - no luck finding yet
  • Loss factor (tan\delta - this might be only describing the dielectric loss of the PEEK - see http://en.wikipedia.org/wiki/Dissipation_factor)
    • dry PEEK (300K) = 4e-3 (Polymers at Cryogenic Temperatures - shortened link: http://tinyurl.com/p849o9b)
    • dry PEEK (125K) = 7.5e-3 (http://tinyurl.com/p849o9b)
    • *** a couple weeks back I thought this was related to mechanical loss, which would be consistent with our lower Q @ cryogenic temperatures result for the body mode - that idea goes back to "unproven hypothesis" status for now.


  1141   Tue Aug 19 13:59:13 2014 ranaCryostatSchematicsCryostat Windows

 We want to replace the cryostat windows with fused silica (IR grade) windows with a Viton seal on both sides. We will get the windows coated by CVI (or someone) for 1550 AR on both sides. Need the existing hole pattern in order to continue.

Some useful links:

  1.  http://www.kimballphysics.com/mcf275-mtgflg-c1vp
  2. http://www.mdcvacuum.com/
  3.  http://www.janis.com/products/AccessoriesandAncillaryEquipment/WindowTransmissionCurves.aspx 


From MDC Webpage:

Viewport Sealing Methods

  1140   Tue Aug 19 11:10:48 2014 DmassDailyProgressLab WorkLaser current sidebands

Can't use current modulation based sidebands in "as-is" setup - there were HUGE (5%) line harmonic (60, 120, 180) dips in transmission which got worse as we started gain peaking.

When I switched to using the Pockels cell for PM sidebands, here were no visible dips in transmission visible on the scope when triggered on the AC line.

This means:

If we want an actuator for the phase corrector path, we either need to hunt for and find/mitigate the source of this ground loop, buy another Pockels cell, or add a DC summing path into the resonant sideband circuit


  1139   Mon Aug 18 13:10:45 2014 ChrisElectronicsPlotsTTFSS HV board TF

I borrowed a TTFSS from the 40m lab to see if its HV stage may be useful for the cryo phase correction path.  Attached is the transfer function of the HV board, measured from TP1 to TP4 -- looks similar to what we previously implemented on the PDH board.  (TP4 should be -66dB below the HV output path in this measurement, due to the voltage divider formed by R51 and the input impedance of the analyzer.)

Attachment 1: IMG_20140815_191728.jpg
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