I measured the calibration df/dV for VCO to be ~48 kHz/V. Then I convert the Vnoise fro VCO to frequency noise.

I injected the signal at 90 Hz and 1 kHz and measure the corresponding peaks from beat noise and VCO.

Then I rescale both data to match the peaks to get the conversion factor.

                        f=90 Hz, pk/floor               f =1kHz pk/floor

VCO                   1.43mV/20uV                   2.95 mV/20 uV

BEAT                 2.18mV/ 25uV                     4.11 mV / 25uV

conversion fac  

 beat =  VCO x K  =  1.53                                   1.37

let's use 1.4 for average

Since df/dV for beat is 71kHz/ V, df/dV for VCO is 71/1.4 kHz/V =  51kHz/ V.

The plot below shows RC noise as measured by VCO and beat note.

What about the frequency response. Why did we see the different shapes between the spectra even with the coherence ~1.
You can measure the transfer function between those two VCO feedback and should try to explain it.

Is there any transfer function in the AOM VCO, for example?
If I can believe D980401-B.pdf, the VCO freq control path seems to have pole-zero pair at 1.5Hz and 41Hz.

The Voltage vs Frequency conversion for VCO is here.

It's not linear over the range. I have to check what is the average value of the VCO during our measurement.

Then, if it's small enough, linear approximation can be used to convert the data from VCO to Hz/rt Hz.

I think It's better to use linear fit because the psd of the voltage does not contain information about the sign of the signal.

• Even after the removal of the Al tip, the scattered light noise looks still exist.
• Particularly I still could see apparent fringe wrapping when I shook the table or touch the foam construction.
   
• I still could not reject the possibility of the backscattering towards the PMC.
• We had the fringe wrapping up to ~200Hz. This corresponds to the motion of the scattered body or the optical
  path for the scattered light by ~100um. Is that possible?
• The VCO feedback and the beat note PLL feedback seemed to have the same information so far.
• The lollipops at the trans mission ports are terrible. They are mechanically incorrect.
• What is the correct conversion between the VCO feedback and the beat note PLL feedback??
  Both are VCO feedback signals but the slope looks different. Need precise investigation.

Today we improved some frequency noise by reducing the scattering light at the ACAV's foam cap.

We are searching for the cause of noise above the noise budget. 

By shaking the table and the foam box, the frequency noise seems to increase a lot at 100Hz to 500 Hz.

We are looking at both beat signal and VCO signal.

First, we added an ND filter just behind the EOM before the beam is split in to two paths to see if we can

reduce the power thus, reduce the scattering light. There is no significant change.

Next, by tapping the optics to ACAV path the noise increases significantly, so we checked and found

that the beam into the ACAV is clipped on the opening of the foam cap and causes some scattering, see [elog entry].

We tear the aluminum tape on the cap of a bit and tried to make the hole bigger, until we could not see the scattering.

The noise is lower by 1 order of magnitude around 100 Hz from beat and VCO measurements,

see fig 1 and fig 2.  We check the RCAV, which has smaller opening, but there seems to be no clipping.

We will check the scattering problem on RCAV again, by slightly remove the cap and see if there is improvement.

Fig 3, shows the comparison between the beat note freq and the VCO signal. A lot of mechanical peaks appears

on the beat note, but not on the VCO signal. We will want to fix the optics for beat note measurement, maybe bringing

the beam back to 3" height. Also, the beam splitter that add both beams together are very susceptible to seismic.

Aside from the scattering at the opening, we haven't found any major component contributing to the noise yet.

Unit on fig 1  (beat) is converted to Hz/ rt Hz.

Unit on fig 2 (VCO) is not converted yet, I have to look up the df/dV calibration for VCO.

After getting converting factor (RIN-> freq noise) = 1.01 MHz/W. I plot the result below.

 The contribution to frequency noise in beat note from RCAV and ACAV due to RIN is calculated by,

fnoise = RIN x power input x df/dp factor. 

The beat note freq noise is plot on the noise budget.

Our beat frequency noise is not limited by RIN. I'll have to think what would be the current limiting source.

I'll check Megan's report about VCO and Marconi phase noise.

I modulate the laser power @ 100Hz and measure RIN from PMC,RCAV,ACAV trans and beat note at 100 Hz, t

and then find The conversion dp/df for RIN to frequency noise to be 1.01 MHz/W. I also measure the TF from the power adj to PMC TRANSPD.

High coherence between RCAV and ACAV trans seems suspicious, and it  changes with FSS gain setting.

My gain setup might not be optimized. I'll check it tomorrow.

I use a function generator to send sine wave signal at 100 Hz, 0.2 Vpk-pk to modulate the laser power.

The peaks at100 Hz for RIN from PMC/RCAV/ACAV  and fbeat are

                Vrm/rtHz at 100 Hz [V]          DC [V]             RIN 100Hz (Vrms/DC) [1/rt Hz]

 PMC          12.49 e-3                             1.5                  8.327 e-3

ACAV          3.27 e-3                              0.313            10.45 e-3

RCAV        11.57 e-3                              0.59              19.61 e-3 

------------------------------------------------------------------------------------------

 fbeat         984 e-3      with 70.8 kHz/V covertion ->     70Hz/rtHz 

The coherence between beat note and RCAV/ACAV rin is ~1. So if we assume that the mechanism

that converts RIN to noise in beat note are the same in both cavities, we have

fnoise ^2 = (RIN_AC x Pin_AC x conversion)^2 + (RIN_RC x Pin_RC x conversion)^2 .

(This might not be accurate, since the peak from f beat is not that relatively high)

Pin for Acav and Rcav are 2.3 and 3.3 mW.

This gives conversion = 1.01 MHZ / V @ 100HZ

This conversion will be applied to RIN level to see how much it would be convert to fbeat noise.

 (assuming the conversion is flat at all freq, we will find this conversion at different frequencies later)

 TF from  power mod input to PMC trans readout is plotted below. The magnitude at 100 Hz is -17dB

 The source from SR785 is split by a T, one goes to ref channel A, another is sent to laser power modulation.

Response B is

-------------------------------

NOTE: (from the yesterday)

 DC level for

ACAV_RCTRANSPD  = 285 mV 

RCAV_RCTRANSPD = 612 mV

PMC_RCTRANSPD = 1.48 V

We need to see a plot of the coherence between the PMC trans and Rcav trans and the Acav trans. Also the RIN of all 3 plotted on the same plot.

Then, inject a small line into the current adjust at 100 Hz and measure the height of the peak on all three trans and the beat signal. Use this peak to scale them all and put them on the same plot in physical units.

Then measure the TF from the current adjust actuator to the PMC trans and close an AC coupled feedback loop on it.

 I realigned the beam to PMC, ACAV, RCAV, optimized gain, and find a significant coherence between ACAV_RCTRASPD and RCAV_TRANSPD.

I haven't re-aligned the beam to each cavities for awhile, and the alignment was quite bad.

PMC_RCTRANSPD: 9.4 - > 10.9 V

RCAV_RCTRANSPD: 1.7 ->2.07 V

ACAV_RCTRANSPD: 0.8 -> 1.3 V

So I need to optimize the gain setup again, see detail below. 

I measured beat signal and coherence before and after realigning. 

For coherence, I see nothing significant except <beat | ACAV>, see fig1, so I did not save the rest of the measurement.

After I realigned the beam, there is a big coherence between ACAV andRCAV see fig 2.

the coherence between PMC and RCAV follow the same trait as that of PMC and ACAV, but slightly less, so I show only PMC and ACAV.

However,  the beat note before and after realigning the beam are still the same, see fig3.

I'll add RIN from  RCAV/ACAV/PMC .

-----------------

gain setup

--------------

PMC: Gain 16

         RF_ADJ 5.7 V

FSS: Common gain 21.5

         FAST gain 21.2

        RF_ADJ  10

These values allow maximum Transmission power and stable lock (I checked this by unlock and lock the cavity and see if the signal is stable after loss lock)

I measured the beat note signal from two different setup (f modulation) and plot the result below

We want to see where our beat signal is, and compare it to the noise budget, and improve the sensitivity.

I'm using the same noise budget for now, because

my noise model from RIN has not been finished yet. I'll try to finish it soon.

For beat measurement, I measured the feedback signal of the PLL loop,

since the UGF of the loop, which is 53 kHz, covers the region of our interest.

I used two frequency span on marconi, 20 kHz and 100 kHz.

I checked the calibration for each frequency spans which are 14.31 kHz/V and 70.8 kHz/V respectively.

The results are the same at low f to ~ 200 Hz, at higher f, 20 kHz span(brown line) has better sensitivity.

*b100 and b20 in the data.mat files are the result of beat at fmod 100kHz and 20 kHz in the format of

frequency, V/rtHz , f / rtHz

--------------

Setting

PMC

power input: 16mW

V_RF: 5.7 V

Phase ADj: 2.23 V

Gain: 13 dB

RCAV

power input: 2.86 mW

V_RF: 10

Phase ADJ: 4.2 V

Common gain: 18.9 dB

Fast Gain: 20 dB

ACAV

power input: 2.0 mW

Three newly repaired SR560s have arrived. I put two in EE lab, and one in PSL.

The current plot for PMC's OLG TF is plotted below. The RF V is 6V, Gain slider is 14 dB.

The UGF is 820 Hz with phase margin (PM) = 180 - 53 = 127 degree.

At higher gain slider setup, the system starts to oscillate. One possible cause is the peak near 10^4 Hz which

might be the PZT's resonance frequency.

Without the notch the total gain we can increase will be limited by the peak.

I'll make a notch to damp it down.

The current spec will be ~20dB notch at 12.5 kHz, FWHM ~1kHz

From current setup, the optical TF should be + 16.5 dB flat, and the gain added by the gain slider is +14 dB.

previous setup we have opt TF = -5 dB and +30 dB from gain slider.

So we have improved the overall gain by ~5 dB and to UGF increased from 530 to 830 Hz.

 I checked that the optical gain in PMC loop increases as the power in the sideband increases. The result is 10.7 dB/V.

This measurement is for checking how much gain (in optical path) will we get from changing power in the side bands.

The excitation is sent to EXT DC channel on PMC. Reference signal is at HV mon, response is picked up at Mix mon.

This TF includes PZT and OPT paths, PZT TF should remain the same independent from the side band power.

I vary the RF voltage, and adjust the gain slider for maximum stability.  The gain setup should not matter

in the TF part we are measuring as long as the loop is stable.

I measured the gain at 3 different frequencies, 290.8 Hz, 1.035 kHz, 5.09 kHz where the TF look reasonable and smooth.

(The loop UGF is ~ 500-900 Hz, Thus the data at 1k and 5 kHz are nicer than that of 290 Hz)

 the slopes from each fit are

The results are fairly linear in our region (RF between 4.8 to 5.9 V). The gain slider for this voltage range is between 13 - 20 dB.

At higher RF voltage, PMC_RCTRANSPD starts to drop significantly.

At lower RF voltage, the gain is too low.

This means we can increase the gain in OPT TF up to 10 dB by adjusting RF voltage (increase side band power)

I measure the OLG TF of the PMC with 3 different RF and gain slider settings. I plot the OLG TF of each setup and identify their UGF.

 I increase the RF as a first step to optimize PMC loop w/o modifying the circuit. This will increase the TF of the optical path.

The setting are

First I  adjust the RF power to reach where I can adjust the stability by changing the gain slider.

RF V above (6 or 7) the gain is too large, even with smallest gain slider, the signal is not stable and the PMC_RCTRANSPD drops from maximum.

So RF V ends up around 5.5 V. this makes the gain slider sit around 10 -15 where I obtain maximum stability. 

The gain slider should not to be set too low because of stability problem. The voltage supply for the opamp should be > +10V.

The gain slider setups are chosen to obtain the maximum stability and maximum power out put (PMC_RCTRANSPD.)

\

c and d have the same RF V, I change the gain to see if there would be any significant change in the performance, and

the data will be used for RF V calibration (how much gain we got from RF V adj).

Now we have some room to increase the gain once we lower the power, but

I have to understand why increasing the gain slider makes signal unstable.

The phase margin seems to be ok.  It might be the slope of the TF at UGF that causes instability.

I plot the TF from each stage in the PMC loop and plot below.

1)  Servo (+ gain slider)[V/V], From the mixer output to the output of PA85. The amplitude can be added upto 30.5 dB by the gain slider setup

2) PZT [V/V]. From PA85 V output to V at PZT. This includes the last R in the servo (R44 = 64.3k Ohm) and C_pzt (0.23 uF). 

 3) Opt [V/V]. This includes the PMC and the frequency discriminator part up to the signal to the mixer.

The PMC converts V -> Hz [1.36 MHz/V]. The PMC pole is 1.9 MHz, so I

assume that it is flat at the region of interest (1-100kHz). The frequency discriminator convert Hz->V, and assuming flat response for now.

Thus the total unit of this part is [V/V] too. I'll separate this part into PMC and+ RFPD later.

I measured the slope of the error signal vs RF voltage, the result is plotted below

 To increase the overall gain of the PMC loop, one thing we can do is changing the slope of the error signal.

This will increase the gain on the optical path of the loop.

So I measured the slope of the error signal, this information will allow me to know how much

gain I would get from each RF setting. The slope increases as the RF voltage increases, until V_RF ~ 8 V.

The error signal does not change at all when V_RF on the slider is between 8 to 10 [Max] V, and

there is no saturation in the signal.

Note: I use the oscilloscope to measure the slope around the center of the error signal, by

measureing dt and dV to get dV/dt around the center, (this can be converted to dV/dHz by the sideband)

but the result has large deviation, so I measure the pk-pk in stead, and divide that by the

cavity FWHM = 3.8 MHz which corresponds to the peak-peak of the signal. to get the average slope.

It will be lower than the actual value

but I'll keep it for now.

I got the calibration from [here]

1) DC ext channel on PMC servo: 32.82 MHz/ V

The DC gain between DC ext channel and the voltage at PZT is 27.65 dB (x24.13),

so the Actuator gain will be 32.82/24.13 = 1.36 MHz/ V;

 The plot on fig1 is the Transfer function of the PZT actuator in MHz/ Volt.

The liso plot, [fig1] offset by 30.5 dB, match the result from the measurement.

This means that the gain from AD602 is 30.5 dB, even though the gain slider says 30dB.

Assuming that from DC to 100kHz, the TF from optic is flat.

The OLG TF measurement must equal The TF from servo(From LISO) + gain slider(30.5 dB) + PZT(LISO) + optics(flat offset)

The offset in the plot is 25.5 dB. With the 30.5 dB from gain slider, TF from optics is -5dB flat, with 180 degree phase shift see fig2. [add calibration from Hz -> V] [plot2]

The result from previous entry which gives the optic's TF to be flat at 1 dB is wrong because I did not use the whole TF from the servo

when I compare the model and the measurement, so I missed -6 dB from AD797.

What are the units of the vert axes?

Separate the open loop gain into three part:

- Optical Gain, Unit [V/m] or [V/Hz], usually flat or simple low path shape

- Servo Filter Gain, Unit [V/V], various shape

- Actuator Gain, Unit [m/V] or [Hz/V], flat or low path filter like up to kHz~100kHz (depending on the time constant of the RC filter),
mechanical resonances above that freq region, which usually determin the highest UGF.

You can change the servo gain by modifying the circuit.

You can change the optical gain by changing the amount of the light in the cavity / on the PD as well as changing the cavity finesse etc.

You can change the actuator gain by replacing the actuator.

Sorry for the confusion, PZT actuator is included in the optical TF. 

The plot on fig2 below shows the TF of PZT part, offset by 1 dB to match the misnomer optical path TF.

Thus, the real optical TF is rather flat with magnitude~ 1 dB, the phase shift is 180 degree,

 and the modifiable TF (LISO model) is plot on fig1. This plot has not taken the gain from the slider into account yet.

Incomprehensible.

Why is the optical TF not (kinda) flat?

Why does the PZT actuator completely ignored?

You need to talk to me tomorrow afternoon when I am in ATF.

I determined the OLG TF of the whole PMC loop, and TFs from servo paths and optical path.

We want to modify the PMC servo to optimize the PMC loop, so we have to know what are the TFs from part where we can modify,

and where we can't (optical path).

The whole TF is measured before, but I remeasured again just to make sure that there won't be any problem from the laser.

How I measure the whole TF is [here].

 I measured the OLG TF from the PMC servo.

The results agree well with the LISO model, see fig 1.

The pole (in LISO model)around 100kHz comes from non ideal behavior of PA85.

When I switch to ideal opamp model, the response is flat.

Then,

 Optical TF = Whole TF - Servo TF.

The Optical TF won't be modified. It will be used to compute the whole TF after the PMC servo modification. 

The measurement at low frequency does not look nice because the signal was suppressed by the gain.

But the TF around UGF still looks fine to work with.

Frank opened up the laser to find any burnt mark, but found nothing and put it back, and now the laser is working.

We don't know for sure yet, what's wrong with the laser. But I'll use this opportunity to work on modification of PMC servo.

The insulation work is done. 

I calculate the spotsize on mirrors of different cavity length.

The setup for FSS experiment can be modified to measure coating thermal noise, (providing we can push down to coating noise limit).

The possible setup will have two cavities with different length. The short one will be more sensitive to coating than the long one.

(I'm not sure yet why should we have two short cavities.)

  I used R=0.5 m mirrors in the calculation, since that what we have. the length starts from 1 cm to 20 cm.

Results from shorter length might be added later.

The blue plot shows the spotsize on the mirror for symmetric cavity,

the red and green plots show the spotsize on flat and curve mirror for curve-flat cavity.

From [ref1], the minimum spot size that the adiabatic approximation still holds (with 1% accuracy) is

R_heat/ w ~0.01

R_heat is sqrt(k / Cf ), so for SiO2, k = 1.38 W/mK, C = 1.6 J / Km^3 R_heat is ~10 um at 10Hz,

Thus, the minimum spotsize is ~ 1 mm. The cavity length can go down to 5 mm.

The pipes are installed. The insulation for the pipes will be installed on Nov 2, Tuesday 8:30am.

The work area will be the same, they just wrap insulation around the pipe, there should not be a lot of dust.

 At 8:30 am, tomorrow, a workman will come in and install two pipes in the lab.

The pipes will be brazed, so no smoke or dust. 

The working area will only above the fume hood near the entrance.

I'll be in the lab during the installing process.

The current 1064nm, 100 mW laser in our setup, NPRO lightwave 126 is broken. We are looking for a new one to replace it.

The laser stopped working when I tried to lock the cavity and saw that RCTRANSPD fluctuated a bit even after I adjust the gain setup.

So I turned the HEPA filter above the table off to see if the signal would be more stable, it was not. When I turned it back on

the laser was off. I don't think the laser and the HEPA filter are associated, but that's what happened.

The power output, as indicated on the laser driver is 8 mW. When I turned the laser off for 5 mins and turned it on.

The temperature of  the crystal started from ~40 C and there was power out, then, in ~10 seconds, the temperature went up to  94 C, and the power dropped to 8 mW again.

The voltage supply for TEC went up to 4V which is the maximum V for cooling.

I switched to the 10W laser driver, the same symptom happened again, so the problem might be the head, not the driver.

 I checked the PSL setup today, and the PMC gain setup has to be changed from 30 dB (maximum on gain slider) to 22.5 for best transmission signal.

From the previous  setup, see quote below, the maximum gain of 30 dB on the PMC gain slider was not high enough,

This means that even though the gain is set to maximum, the signal from the transmitted light does not oscillate.

But today it did oscillate, and I had to reduce the gain to 22.5 dB. When I checked the PMC gain, I turned off the FSS loop

to make sure that the FSS loop won't actuate on the NPRO, and the signal are purely from PMC loop.


There are no other changes of parameters. The power input is 30.7 mW, RF power, phase shift are the same

as before.


After adjusting the PMC's gain, I also roughly adjust FSS's loop gain. I haven't optimized it yet, just determined it by looking

at C3:PSL-RCAV_RCTRANSPD signal on the oscilloscope.

common gain is changed from 5 to 8 dB

Fast gain is changed from 8.5 to 13 dB.


I saved change the values for PMC gain, common gain and fast gain in STARTUP.CMD file.

I don't know what causes the gain changes here, I will check the TF of PMC loop again. If it is real, it means

we might not have to modify the PDH servo card.

NOTE: the slowDC for locking both cavities is ~ -0.103 V

Summary: The PMC's FWHM was measured to be 3.8 MHz.

- Motivation

We are characterizing the PMC loop TF. We needed to measure the cavity's FWHM in order to know the cutoff frequency of the cavity.
This number will be used in the Simulink model to simulate the TF of PMC.

- Method

The length of the PZT was scanned, while the PDH error signal was also recorded.

The time span between the maximum and minimum peaks of the error signal was measured in order to obtain the width of the cavity resonance. The time-to-frequency conversion [second to Hz] will give us the FWHM. The conversion between the time and the frequency was obtained by looking at the zero crossings of the error signal which are separated by the modulation frequency.

- Measurement setup

• A function generator provides a triangular waveform at 200 Hz, 10V pk-pk, which is split by a T connector. One goes to an oscilloscope for trigger, another goes to EXT DC channel on PMC card.
   
• The signal going to EXT DC ch is used for scanning the PMC.
   
• The sideband is 21.5 MHz away from the carrier.
   
• Another ch on the oscilloscope is connected to MIX OUT ch on PMC card.
  This measures the error signal after the mixer.

- Result

• The time span between the carrier and the sideband: 768 us.

• The sideband frequency: 21.5 MHz

• ==> the conversion factor: 21.5 MHz/ 768 us = 28 GHz/s

• The time span between pk-pk of the error signal at the carrier resonance: 136 us
   
• ==> The FWHM is 136 us x 27.5 GHz/sec = 3.8 MHz

- Discussion

1) The cavity pole frequency is obtained from the measured FWHM. It is FWHM/2 = 2 MHz.
FSR is  c/Lroundtrip = c/(0.42m) = 714 MHz.
Thus we can compute the finesse F = FSR/FWHM = 188.

2) Another way to measure the FWHM is by measuring the transmission peak of the transmitted light while scanning the cavity. I’ll try this and see if two results agree.

Many thanks to Koji for useful discussion on the measurement and how to improve elog quality. 

Hoo! So beautiful fit!

I measured PMC's TF between MIXER OUT and PCMON, the measurement agrees well with LISO model.

The source is 10mV swept sine. It is sent through FP2. MIXER OUT and PCMON are connected to chA and chB respectively.

The measurement was done twice, with loop closed (PMC is locked to the laser )and loop opened (PMC's enable switch is off.)

The data from LISO model is added by 30dB to match the gain slider.

The result from closed loop looks weird at low frequency, I'm not sure why so I'm reading Application note 243 to find out about coherence.

I did this to make sure that the schematic and the board are matching, all parts in the schematic are labeled correctly.

Please ignore fig1, I did not have phase plot in it.

I fitted the TF of PMC servo card simulated from LISO, and found poles at 2.01 Hz, 59.7 Hz, 13.4 MHz, and zero at 479.4 Hz.

 From last week, I used LISO to model the TF but it did not provide the values of poles and zeroes.

Thanks Koji who taught me how to use LISO fitting feature. Now I can find poles and zeroes of the simulation.

The frequency range spans from 1 Hz -  1 MHz, covering out region of interest (~1 Hz up to a few hundred kHz).

  For higher frequency, I couldn't make it work yet.

pole 2.0139158941  ### fitted (name = pole0)
zero 479.4339553158  ### fitted (name = zero0)
pole 59.7091341574k  ### fitted (name = pole1)

factor 9.5703858437

pole 13.4116105660M 99.7575256877m  

The plot shows simulated data and fit data, we have a nice fit that we can hardly see red and green plots.

 The code for simulating TF is pmc.fil, which can be found on previous entry, the code for fitting, tffit.fil, is attached below.

I measured the capacitance of PMC's PZT to be 0.23 uF.

The PZT actuator attached to PMC middle mirror is used to change the total length of the PMC cavity.

It has internal capacitance which effects the TF of the system. The value is used in LISO model to calculate the PMC loop TF.

To measure that, I removed high voltage output from PMC card from the PZT connector on the table,

discharged the PZT with 50 ohm resistor for a minute. Then I used an L-C meter to measure the capacitance.

I started work on a LISO model of the PMC servo - it does not yet agree with reality.

Yesterday, I measured the open loop gain (OLG) of the PMC loop.

It consists of two parts, which are the PMC servo's OLG and the rest, i.e. OLG of photodiode, PMC, PZT actuator.

Knowing each part's OLG is useful for modification.

Since we are going to do most modification on the PMC's servo, we want to know what is its TF. 

This is where LISO comes in. I use it to simulate the TF of the PMC servo.

I don't know how to model AD602, because it is not actually an opamp and therefore not in the LISO opamp library.

The datasheet says it has -3db at 35MHz. Thus, for our region of interest, it probably has a flat response. It is just an

adjustable gain amplifier.

I'm not sure how to use LISO to calculate poles and zeros of my model yet. I'm reading the manual.

This simulation will be compared with the measurement.

Once we verify that all the parts behave the way they should, we can think about the modification.

We want to modify the TF because even though we maximize the gain slider, the system is still stable

( no sign of oscillaltion from too much gain.) It means we can still optimize our TF for better stability.

 Lastly, knowing the servo's OLG and the whole loop OLG,

we can compute what is the OLG of the rest of the system by simple subtraction.

I hope the quality of this elog entry is improved, however slightly it might be. It has motivation why I do what I do, details, and people who want to reproduce my work should be able to follow it.

Thanks Koji for a useful discussion on how to elog properly.

The attached plot shows modeled transfer function of the PMC Servo card + PZT capacitance.

Components' names in LISO code are taken from the schematic

I'm working on Simulink model to calculate PMC's open loop gain. For now, all the parameters for frequency discriminator are copied from linfss6.m.

I'll work on correcting those parameters later.

The simulink model will allow us to learn how external noise will look like in our system.

The bode plot for PMC's TF is plotted in fig1.

I'm not sure why the measurement on both magnitude and phase looks bad at low frequency (f<10 Hz).

I think the model (fig2) is not correct because of the wrong units ( angular frequency,w, to frequency, f.) I'll check  this again.

The phase part seem to be completely wrong.

The parameters, (for example, cavitiy pole, frequency discriminator) for  the TF are not confirmed yet. I'll elog them once I verify them.

 *******************************

Yesterday, I had a chance to test U6(OPA 27) in the schematic.The measurement agrees very well with the computed result, see fig 3.

The signal from source out was sent through FP2 test. TP2 and TP4 were connected to channel A and B on SR785 respectively.

SR785 was set to swept sine mode, frequency span from 1Hz to 10^5 Hz.

So we know that at least one part of the servo works properly.

When I showed Frank that the card did not work, I mentioned that I got electric shock when I connected high voltage input to the card.

Frank realized that it happens only when the connector is not grounded, and it was not. The ground connection from HVin to the card

was broken, It's hard to see unless I touched the wire that connected the board to the female BNC on the panel. I replaced that rigid wire by a flexible wire.

Now the PMC card is working fine, and the original PA85 might not be broken at all.

I checked PMC circuit to make sure that the schematic matches the actual board.

This is important because we want to make sure that it works exactly the way it should.

This is also useful when we want to modify it.

I check R and C on the board, everything seems fine except R2.

It says 9091 F which matches the schematic (9.09 k), but a multimeter reads 4.5 K, I might measure it wrong.

I also measured the TF between TP2 and TP4, I'll attach the result.

When I measured TF between TP3 and TP8,  I did not fully push the card into the socket, and this killed PA85 when I applied high voltage to the card.

Koji found me an unused Mach-Zender card at 40m. I replaced its PA85 for PMC card.

I tried it. This time, I made sure that the card was properly in its place, I screwed it down to the crate.

Then I connected PD cable, LO cable, HVout then HVin.

However, it sitll does not work , no high Voltage coming out. Only low voltage ~0.5 V which changes with the value on the PMC DC control slider.

I don't know if I accidentally killed it during the transplant operation.

It's a very bad day to be PA85.

C3:PSL-NPRO_PWRMON channel, and npro.db file are added

for monitoring power output of the NPRO (reflected beam from Faraday Isolator)

I haven't reset the crate yet, so the channel may not appear in DAQ yet.

It's connected to VMIVME-3113 at #C0 S60.

The photo diode is Thorlab PDA55 with RG1000 filter.

Calibration for power is 1.81 mW/V, now it reads 5.56V.

Today I measured open loop transfer function of PMC loop.

The measurement has two parts.

First the swept sine signal is sent to FP2 test point, TP4 is connected to A, TP3 is connected to B,

the magnitude, B/A, gives us [C][D][E] .

For the second part, the swept sine is sent to ext DC channel,

TP3 is connected to A, and TP4 is connected to B.

this is the TF of [F][A][B]

======================================================================

 [A]--<FP2>-----[B]-----{TP4} ------[C]-----[D]---<Ext DC> ----[E] ---- {TP3}----[F]-----> back to [A]

======================================================================

The magnitudes and phase from both measurements are added up

to get the whole open loop TF of PMC loop [A][B][C][D][E][F].

UGF is ~1k Hz.

PMC setup

Gain 30dB (mzx)

LO PWR 0.585

Power input 30.9 mW

PMC_PHCON 2.5 + 180 flip

PMC_RFADJ 4.0

I'll verify that the schematic matches up with the real circuit we are using.

After I increased the pk-pk of the error signal for RCAV system (by increasing the RF power to maximum), 

 C3:PSL-FSS_FAST is still railing once it goes below ~ -2 V kicking the system out of lock.

I'll check again if the laser is lock to the center of the error signal or not.

The FSS servo is not working. When SLOWDC which controls the NPRO temperature is brought to near resonance,

it falls off the lock when we enable the loop. So it's a debugging day.

First, I aligned the beam on RFPD, making sure that the beam is on the center, and align the beam to the cavity.

The signal from transmitted beam oscillates a lot. The gain was too high

(This is surprising, we haven't changed any power but the gain is too high already.) 

the set up is changed as follow

common gain, 16  --> 7

Fast gain,   15 --> 14

RF power  7.0 --> 7.2

Phase Adj  4.5 --> 3.92

Then I checked the error signal from MIXER OUT channel on FSS card. The signal looks fine.

The DC level is ~ 17 mW, peak to peak level of the carrier is 160 mV, and 39 mV for sidebands.

[the setup is:

RF power 7.0 V
Phase adj 4.5 V +180 flip]

The transfer function between In2 and fast mon seems fine.

---------------------------------

Now the loop can be locked, but when FSS_FAST gets lower than -2 V. It runs away and lose lock.

This does not happen when it goes to plus sign. it can go up to 5 or 6 V before losing lock.

So we increased the RF power to maximum (10 V) which increases the error signal pk-pk to 394 mV.

FSS MON seems to stop railing, I'll adjust the gain setting again to minimize fluctuation in mixer out. 

 A photodiode for measuring the laser power reflecting from the Faraday isolator's in port is added.

The cable is prepared and connected from the PD to DAQ.

The channel name will be C3:PSL-NPRO_PWRMON. 

all channels are working now

rebooted the PSL crate to see if it fixes the problem with some of the channels inaccessible from external computers

workers finished the piping stuff for today but have to come back to connect it to the stuff one floor above. It's not the sprinkler stuff, it's heating water.
So some other guys will show up in the future to drill holes for the sprinkler pipes and installation of the sprinklers.

turned the laser back on

Plot from yesterday measurement. The new result is in red.

The data is measured from the feedback signal in PLL loop. Its UGF is about 53 KHz.

The data is calibrated to Hz/ rt Hz unit by a factor of 71.3 kHz/ volt.

    After PBS is rotated to the right position (yesterday I made a mistake by minimizing the split beam.

The split beam is minimized. This guarantees that the beam passing through is highly linearly polarized.

The reflection back to PMC is reduced from 1.9 mW to ~1.1 mW.

The cross correlation between the beat measurement and RIN is measured before and after the reduction of back reflection.

The plot shows certain correlation between RCAV's RIN and beat measurement when there is back reflection to PMC.