ACAV's PD for transmitted beam is connected to PSL1

RCAV's PD for transmitted beam is connected to PSL2

And now RCAV_RCTRANSPD dropped from ~5.3 to ~5.05

               ACAV_RCTRANSPD dropped from  ~2.5 to ~2.4

I reflow soldered 3 channels to an AD590 temperature board for a differential measurement test. I expect to run the experiment soon to see the noise of the sensor. 

show us what you got for the LISO or ZERO noise budget of this circuit before testing; also sketch of the experiment setup

AD590 Zero Noise Budget Analysis

• I modeled the noise of the ADR4550 from its datasheet (https://www.analog.com/media/en/technical-documentation/data-sheets/ADR4520_4525_4530_4533_4540_4550.pdf Figure 78 Page 28), with the modeled noise slightly more than described. 
• I used a flat value of 40pA/root(Hz) for the noise of the AD590 as described in Table 1 on https://www.analog.com/media/en/technical-documentation/data-sheets/AD590.pdf

Because of the previous data comparing the open box vs the closed box was varying unexpectedly, I re-ran the experiment with both sets of data taken on the same day. 

• The closed box data was taken from 8 am to 1 pm on 8/9/2019
• The open box data was taken from 1:45 pm to  6:40 pm

Now both sets of data show very similar curves, with the box open data slightly above the closed data in most areas. Maybe the last data was different due to other systematic conditions on that day, but it does seem like now the difference is gone. Both sets follow the original closed box data measurement from before.

in order to gain more s/n ratio i modified the existing AD590 readout-box a little bit. I assumed that we wanna operate the cavity at 35C (which is not too high but well above RT or the temp of an additional temp stabilized box around both cavities) The required range for shifting the cavities is ~ 1 FSR, better would be a little bit more for each cavity as we can shift both independent.
As

df~156MHz /K    and     1 FSR~740MHz

this corresponds to ~4.75K/FSR we have to shift.

For testing purposes it might be helpful to have more than that as e.g. if we limit the total range to lets say 6K we might end up at the end of the range and run into trouble as soon some disturbance from outside (e.g we remove part of the insulation, lets say an end cap) might shift the whole thing at the end of the range. As soon as this happens the servo would go crazy.

So i think we should go for 10K range, centered around 35C, so from 30C to 40C. I modified the box for that, so the transimpedance resistors have now a value of 29.4K, which gives us ~9.21V for 40C at the output of this stage.

In order to supply it from an independed power supply to reduce our current ground loops, i've chosen a WM071, the same as we use for the PDH boxes. As they come only in +/-15V, i had to change the voltage regulators in the box to +/-12V instead of =/-15V.
This results in a maximum output voltage of the LT1125 of a couple of 100mV more than 10V, depending on the current they have to source/sink. So 9.2V is still well below the max.

I added a filtered 5V reference, (AD586, 4.7uF filter cap) for the dc offset @35C. The corresponding resistor for the summing amp is 1379.76 which can be implemented almost exact using 2k05 and 4k22 in parallel (1379.7) or 1k54 and 13k3 (1380.2). The feedback resistor of the last stage can then be calculated to be 170k45 in order to match 30C to 40C to -10V to 10V. Paralleling can be used here as well to get an almost exact value.

The matching is not that critical as we don't wanna measure absolut temp, but if can do it that easy why not.

---  new schematic following soon   ---

Summary

The AEOM has been installed in the South path replacing the EOM 21MHz used for the PMC. There is a high noise that I clearly see at the photodiode in transmission. 

When I have placed the AEOM in the path I have decided to take the alignment of the previous EOM as reference. Not ideal because the reference should be the incoming beam. The beam is not parallel to the table and it was decided to be as less as possible invasive. The mode matching and the alignment gave at that time 20% of visibility (at each polarization). After the installation parameters where unchanged. Later I have improved the alignment bringing the visibility at 30% for both the polarizations. After that, when everything was in place I have easily locked the cavity but the power in transmition was showing a very high noise. I have spent all the day trying to twick the alignment because and servo loop gain, but we need to solve this before going further. My back does not allow me to proceed for today.

NOTE:

I have also noted that the South Laser which is labeed 2W laser has the lambda/4 and the lambda/2 rotated in a way that at the output of FI we had few mm. I am not sure if damping the power at the FI is a good thing.

I made a simulation for PMC body mode, and found out that for Al PMC, the first body mode is 1kHz. And 780 Hz for stainless steel pmc.

November 27, 2012

 It is desirable for the first body mode of the PMC to be at or above 1000 Hz in order to provide consistent length for the cavity.

 ANSYS Summary

1. Geometry
   • For any given test, all parts of the assembly were changed to be the same material.  Materials tested include fused silica, aluminum, and stainless steel, and material will be specified per test.
2. Connections > Contacts
   • Most contacts determined by ANSYS were sufficient.  However, the contacts between the press-fit slots or cones and the ball bearings were adjusted for accuracy.  The "Pinball Region" was changed from "Program Controlled" to "Radius", and the radius set to between 0.1 and 0.3 meter.  This is to ensure that ANSYS recognizes these two objects are resting on each other.
3. Analysis Settings - Supports
   • The PMC assembly was modified so that the constraints could be more accurately modeled.  Split lines were added to the bottom of the PMC base to model the force applied by the dog clamps, and these small 0.5"x0.5" squares were defined as "Fixed Supports" in the ANSYS model.
   • The entire base was labeled as a frictionless support, because it is sitting on a table. 
4. Results
   • The first body mode when the PMC is made entirely out of stainless steel is 865Hz
   • The first body mode when the PMC is made entirely out of aluminum is 881Hz

Above you can see the first mode shape of the PMC.  The colors represent the displacement - deep blue indicates no motion, while red indicates the greatest amount of motion.  The animation of this mode shape shows the PMC spacer rocking transversely on the PMC base.  The PMC base does not move at all.

5. Changes to geometry (December 3rd, 2012)
   
   • Holes were cut through the PMC spacer to try to increase the first body mode frequency.  New geometry is shown in the figure below
   • 
   • There was no increase in the first body mode frequency - when made out of stainless steel, ANSYS reported the first body mode of this to be at 855 Hz

 One question that came up is whether ANSYS is importing the geometry file at the correct size.  According to the scale on the screen, it is the right size.  However, when the material is changed to resemble fused silica, the lowest body mode is 998 Hz, which is about an order of magnitude lower than expected.  This indicates some other error, possibly in importing the structure into ANSYS.

 /more to come

 This does not look like a longitudinal mode. Do you have the frequency for the first longitudinal mode(along cavity length)? the first longitudinal mode should look like this ( this model has no fixed boundary condition, just a block in space).

Kristen and Norna came to ATF for impact-hammering of the metal PMC in the gyro setup

LINK

 As Frank suggested, I edited the drawing, so that the adaptor plate can accommodate both types of AOM.

I made drawings for aom block and adaptor plate. The assembly is for 3" beam height.

The assembly is consisted of two parts. The bottom aluminum part is for mounting new focus 9071 4 axis stage on the table.

The top plastic part, is for mounting an AOM to the 4-axis stage.

The bottom part actually is designed for a standard EOM, so the height is  3". With a plastic adaptor plate, it can be used for an AOM as well, so I'll order a few of the alumnium parts.

There are 2 designs for AOM adaptor parts, because we have two AOM models. They have different screw size and mounting positions, I 'll order a couple for each design. 

 New setup for fiber phase noise cancellation with one AOM

 

 

seems like a bad AOM and an impedance matching problem together. exchanging the semi-rigid cable by other cables produces a lot of different results in power modulation / pointing vs frequency. right now i found a combination where for frequencies 80MHz and higher it is almost flat, but below it drops a lot. the pointing is related to the absorbed power in the AOM. - you can really feel the heat of the crystal, no joke. will measure the temp on tomorrow for different frequencies where the pointing is worst case to see if it is a macroscopic change in temp. Aidan has this nice thermal image camera to do this. i think we should try the crystal technology AOM we have. i don't think that aligning helps a lot here. tried this the whole afternoon :-(  the lowest power modulation over the entire frequency range is about 20%_pp

There appears to be a factor of 5/3 difference between the ASD output of the SR785 and the CRIME lab's cymac3 ADC.  To be clear, .  I am not sure why at this point.
Earlier today, Gabriele helped me calibrate cymac3, telling me the voltage range (+- 10V) and ADC bit number (16, so 2^16 = 65k counts), giving a calibration of .  When I apply this calibration to the median-averaging ASD code Gautam provided me, I appear to underestimate the true ASD, where the "true ASD" is as reported by my SR785. 

The first plot is the SR785's response to a 1 Vrms sine wave at 512 Hz, with 1 Hz bin widths.  The ASD peak at 512 Hz reaches 0.7086 Vrms/rtHz.

The second plot is the calibrated cymac3 ADC response with 1 Hz bin widths.  At 512 Hz it reaches only 0.4252 Vrms/rtHz.

In the end, I decided to just apply the mystery gain of 5/3 to the ADC.  This is the final plot.  Seems close enough.

This ratio of 5/3 is suspicious and makes me think the calibration of the ADC is wrong.  I am having trouble locating the ADC datasheet.  I checked around at a couple of injection frequencies and things seem consistent with the correction factor applied.  I will look more into the ADC in the morning, but I don't want to spend very much more time on this.  The whole point is to have a quick, constant check on our beatnote ASD which is continually being saved and compared to the ASD from five minutes ago, ten minutes ago, an hour, a day, to easily see how our final product, the beatnote ASD, is moving all the time.  If we kill a bunch of noise, we want to know when and why.

EDIT: I checked the time series signal and found that the ADC was reporting half of what I was putting in.  This is because the cymac3 is expecting differential inputs, and our beatnote BNC is grounding the low input.  I didn't understand differential and single-sided ADCs until today, after reading this.  Turns out the fully-differential is able to double its input range sensitivity with the same voltage range input by splitting the input 50/50 and flipping the low input's phase by 180 degrees.  If you ground the low input, you halve your signal.  I corrected for this by multiplying my calibration by a factor of two to account for my signal being halved.

So now my factor of 5/3 becomes a factor of 5/6. 

But the problem still remains, why is the median-averaged ASD from the ADC and RMS ASD from my SR786 for my 1 Vrms sine wave injection giving different final results?
To explain this correctly, I have to go through the median ASD estimation process in Gautam's python code. 
First, we use scipy.signal.spectrogram to estimate a bunch of PSDs from our input time series.  We take the square root to get ASDs.  Then, we collapse along the time dimension by taking the median value for each frequency bin, call this the median ASD.  Finally, we divide by the bias factor of to get from a median ASD to an RMS ASD, the industry standard.  This shown in Evan's thesis, Appendix B.

However, I misunderstood some of Evan's assumptions in this thesis.  Importantly, the Rayleigh distribution for magnitude is only good for zero-mean Gaussian noise.  If you inject a signal, your mean and your median are gonna be about the same at that frequency, because the signal completely drowns out the noise.  You have to first eliminate your signal, apply the bias to go from a median to RMS ASD, then add the signal back in.

If you ignorantly multiply your entire ASD, including signal, by , like I did, you will find that your ADC calibration results seem to be overestimated by a factor of about 5/6, because .

To test this, Gautam and I used the RIGOL function generator to generate 1 Vrms of noise.  We found that (1) the two channels of the RIGOL produce very different actual levels of noise when 1 Vrms is requested, and (2) our median-averaged ASD python code produced the same noise levels as the RMS ASD taken by diaggui, which we are sure is right.

This is encouraging.  We now trust that the median averaged ASD estimation code is doing the right thing, particularly since it matches up with diaggui's output.  But the python code has the advantage of (1) being in python, (2) being median-averaged, and therefore robust to glitches, which our PLL has every time we change the Marconi carrier frequency, and (3) being easily callable and controllable whenever we want to take a permanent spectra. 

I plan to make a crontab tonight which runs the dailyNoisebudgetPlotter.py every minute or so.  I also have some javascript code from Max Isi which updates a webpage client-side to show whatever new plot has been produced.
I won't save every plot we make permanently, because each plot is ~200K, and there are 1440 minutes in a day, so 288M in plots would quickly become ridiculous.  But I think saving one from every thirty minutes is good, ~1M of plots each day is doable.

[Rana, Tara, Evan, Eric, Nic] 

We are designing a PMC, to do that we should be able to answer some fundamental questions about a PMC.

Why do we want a PMC?

• Intensity stabilization(is it?)
• Filtering of beam profile
• Alignment reference
• Reference for waist size and position
• Jitter suppression
• Polarization filtering

What should we consider in the design of the cavity?

• High finesse eventually causes the transmission to drop
• Cavity g should be chosen so that higher-order PMC modes are well outside the bandwidth of the TEM00 mode
• Finesse should be chosen so that laser intensity noise is suppressed below the shot noise limit at frequencies of interest
• Number of mirrors - determines whether or not we get polarization filtering
  • In a 3-mirror cavity, when one polarization is resonant, the other will be antiresonant. In a 4-mirror cavity, both will resonate simultaneously (with small differences in transmission intensity and phase).

What should we consider in the design of the spacer?

• Acoustic pickup (probably dominates over seismic pickup at frequencies we are interested in)
• Steel or silica:
  • Longitudinal mode frequencies are comparable
  • Silica is harder to machine
• Vacuum can?
• Dimensions:
  • A shorter PMC will have higher vibrational modes
  • A longer PMC has more filtering
  • A spacer that is too thin will have easily excited bending modes

Other:

• 15 MHz is too low for PDH modulation, because the bandwidth of the resulting loop is too low
• For a resonant PD, we want a low Q (a few)
• 3 mirror or 4 mirror design?

Considerations for PMC design:

1. Stiffness(Acoustic susceptibility) & heavy material: With heavier material, the pmc motion on the support becomes smaller.(RXA: please quantify with a formula)
2. Filtering factor (Finesse/FSR/Cavity pole), g-factor: Filter out intensity noise around 10 MHz (RXA: please quantify with a formula)
3. Design for thermal expansion cancellation between the spacer and the end cap: So that the PMC is less sensitive to ambient temperature
4.  3 or 4 mirrors?  3 is polarization selective. For general lab use with power less than 1 W,  3 mirror design should be good. (RXA: I don't follow this logic at all)

RXA: In general, all of these considerations need some sort of quantitative detail. Make a DeBra Matrix so that we can evaluate. 

Measured the noise floor of the accelerometer to compare it with the seismic noise we measure with it (and the TFs i want to measure). Suspended it from the table using a very soft spring to make it quiet enough. Resonance frequency of suspension was ~1Hz (small peak visible in spectrum, damping of the thing took forever).

Seismic noise of the table is hitting the noise floor of the accelerometer at about 10Hz. Couldn't find the raw data from the seismometer measurements. Will update this later.

found the old data from the seismometer measurements we did some time ago. Measurements agree more or less. Seismic noise floor measured some time ago was higher above 30Hz, but that was measured at the time where we had the construction going on and not during vacation time.

I set up an Acromag slow controls based on the procedure that Keith wrote in T1400200. It's really pretty easy. It took an hour and 15 minutes from installing Ubuntu on a machine to having a functioning ADC channel from the Acromag unit. I haven't yet set up a DAC unit - this will require some tweaking of some of the EPICS parameters. Once I've done that I'll upload a complete procedure to the Wiki.

This is relatively promising for supporting/replacing VME slow channels.

Aidan.

Success!

I configured the Acromag XT1541 DAC to run with EPICS. This was a touch trickier than the ADC as there is a subtlety with the channel configuration in the EPICS database. The bottom line is that now I can change the value in an EPICS channel and a multimeter attached to the unit will show a corresponding change in voltage.

The attached files (ioctest2.cmd and IOCTEST2.db) are used to access the first output channel, OUT00, on the unit. Now that I've got the thing working I can debug the calibration. Once that's sorted I'll summarize the set-up procedure on a Wiki page with glorious detail for future reference.

The following command line is used to open the modbus EPICS server.

${EPICS_MODULES}/modbus/bin/${EPICS_HOST_ARCH}/modbusApp ioctest2.cmd

The ioctest1 files are for the ADC unit.

https://nodus.ligo.caltech.edu:30889/ATFWiki/doku.php?id=main:resources:computing:acromag

I managed to figure out the modbusDrv configuration settings to get the binary output of the Acromag working. I've updated the Wiki page to reflect this. I've wired the XT1541 DAC, BIO Acromag unit to the T1EN and T2EN channels on the TTFSS box but I still can't get remote control of it yet for some reason. When the PDH loop is closed and I switch the TTFSS box to REMOTE, the loop stays closed regardless of what I do to the binary outputs in EPICS.

https://nodus.ligo.caltech.edu:30889/ATFWiki/doku.php?id=main:resources:computing:acromag

I had hooked up the a channel to monitor the tank temperature. However, now I am seeing random noise and also zero readouts on every other sample.  

I had something similar over the last week when I was trying to set up autolocker and PID slow controls for the laser. The fast monitor signal reported by epics was very noise. There I thought it was just poor performance of the FSS locking loops and that I would need to get in and check alignments, polarisations etc.  Now I'm thinking there is something wrong with the chain from the ADC to what caget spits out.  

A calibrated voltage input of +2 V was put into the C3:PSL-VACTEMP_MON1 channel (acromag channel regiser 7 (out of 0-7).  A quick grab of the ouput is attached below, it doesn't look great.  Might have to look tomorrow to see if this is something to do with modbus dropping measurments or if there is some other issue.

I was double checking the UGF of the PLL and was getting the strange results of 8 kHz UGF with the standard settings of 10 kHz/V and SR560 set to 23 dB gain (Marconi was set to +13 dBm RF power level). In the past we got 30 kHz for this configuration.

Since then an acromag channel was connected to the same 600 Ω output of the SR560 as the Marconi signal generator. This was to monitor the state of the PLL loop so that the Marconi center frequency could be adjusted (through GPIB and a python script) to always have the PLL locked.

It turns out that the acromag channel monitoring the BN DC level is overloading/lowpass filtering the 600 Ω input. Attached below is a transfer function of the SR560 with 2x100 gain setting (no filtering): here the BN has been disconnected this is just the preamp itself. With the acromag connected it appears to add a pole at 3 kHz. This will have changed the PLL loop response in recent measurments.  

---

I have now moved the acromag to the 50 Ω buffered output of the SR560, reported voltages should be the same (acromag is high impedance) but the loop now has the expected 30 kHz as before.  Calibrations of previous week should be taken with a grain of salt.

Measured the acromag noise (input and output).

For the input noise, a function generator was set up to input a sin signal of 0.1Vpp at 0.1Hz and connected to an input channel. The time series was recorded for 4 hours (sampled at around 10Hz, but not exactly due to differences in how long each loop took to run), and then the Lomb-Scargle algorithm (Fourier transform with uneven sampling) was applied to obtain a frequency spectrum. The result was then run through Andrew's rms.m script (attached). Both are plotted below:

For the output noise, a script was written to continuously write a 0.1Vpp at 0.1Hz sin signal to an output channel that was connected to the signal analyzer, which was used to take the following data.

I showed Andrew how to set up the slow controls. We got the previous 3 ADC channels back up.

• C3:PSL-TRANS_ACAV_ISS_DC
• C3:PSL-TRANS_RCAV_ISS_DC
• C3:PSL-TRANS_RF_DC

Also:

• rewired a few BNC feedthroughs up through the front panel of the Acromag chassis
• bolted the front panel to the chassis independently of the screws that mount the chassis to the rack

The computer acromag2 has inexplicably died.  It turns up some error messages recovering jornal on /dev/sda1 that doesn't bode well.  All computers in the lab are backed up to somewhere remote so nothing is lost.  Also we version control everthing that matters on the computers anyway.

Rather than debugging a potentially faulty hard drive issue we will just replace the HD with one from the draw (Larry gave me a box of small/midsized hard drives about a month ago).  

Today I installed the Faraday isolator after the PMC. Tara and I then spent some time trying to figure out why the PDH error signal suddenly had a huge DC offset (it was because I accidentally knocked the angle control on one of the HWP mounts while installing the FI beam dump). Before installing the FI, we had observed that the loop oscillates noticeably at about 100 kHz and had hoped it was caused by back-reflection into the laser (which the FI would fix). Installing the FI seems to have no effect on the oscillations. After installing the FI I adjusted the HWP immediately following and retuned the phasing of the PDH loop by adding some extra cable to the PD SMA input. I've attached a picture showing the sweeps of the cavity refl response and PDH error signal, and a picture showing the oscillations when the loop is engaged.

I tried minimizing the rejected light out of the FI to optimize the angle of the QWP directly in front of the cavity, but this light appears to be dominated by reflections other than those off of the cavity. The rejected light consists of two distinct spots which can be seen with an IR card. I think one of them is a reflection from the lens immediately following the FI, and the other is a reflection from the 14.75 MHz EOM.

Edit Thu Apr 12 22:09:21 2018 (awade): WRONG, THE ISSSUE IS THAT ACROMAG NEED MIN 6V TO OPERATE BINARY INPUTS

When Craig restarted the acromag IOC yesterday the North path FSS loop engage binary channel went into a permanent latch off mode.  This is a recurring problem that can be fixed by plugging the 5 V power in line to the acromag binary channels in with the FSS control boxes unplugged. Sometimes you need to plug and unplug a few times.  

It could be an issue with the way we have used 820 Ω resistors to bring the pull up 10 kΩ down to 758 Ω. It probably should be buffered somehow.  For now its good enough to get it working, once it's powered up its fine.

As an intermediate fix I soldered a 1000 µF electrolitic cap in line with the 5V supply to give it juice when first powered up.  This seems to make the latching go away most of the time (90%) when first powering up the units. So... slight improment.

Last night I soldered together four BNCs and attached them to one of the acromags with some free channels.  Turns out this acromag is named vader (10.0.1.50), and was being used for temperature sensors.
I just added all four channels into /home/controls/modbus/db/LaserSlowControlsAndMonitors.db , and named the first two additions C3:PSL-SCAV_REFL_DC and C3:PSL-NCAV_REFL_DC

All of this was done so I could tell if the "breathing" we see in the TRANS_DC is really representative of power in the cavities or just some crap happening on our tiny ISS board in transmission.  Seems like it's real power in the cavity, but we'll see after like an hour.
If so, we need to control the temperature of our EAOMs.  Thermal hats, peltiers, I don't care, needs to happen ASAP.
 

I was finding myself changing thresholds every time I change power levels. Also, the fastmon rms monitors were not working with the docker-compose. I'm listing all script and medm changes since the sad departure of awade:

• rmsMonitor.py is standalone now and does not depent on gaincycle.py.
• rmsMonitor writes rms values in a channel and reads thresold value from a channel. There is also a switch to disable gain cycle functionality.
• In FSS slow controls, I made channels for showing light power in mW (after calibration) and a channel that calculates mode matching. These values are of course not very accurate but will give us an idea.
• I made thresholds into channels in the autolocker script. Autlocker script also had a logical bug with dealing with required states. I have fixed that now.
• I have added threshold channels for PMC autolocker as well. Now, the threshold value can be changed by medm screens directly.
• There is a new functionality in autlocker called tracking. When engaged, the autolocker top and bottom rails follow the changes due to slow pid so that a rough position of resonance is always in memory.

I'm attaching screenshots of modified medm screens.

Today, I added a new out-of-loop transmission PD (Thorlabs PDA10CS) for the north path. This will be helpful in future measurements of RIN coupling to beatnote noise. This PD is added at (8, 42) using the dumped light. The optical layout would be updated in a few days. I've also connected Acromag channel for North Transmission DC to this photodiode, so the transmitted power channels and the mode matching percentage channel of North Cavity are meaningful again.

ISS Gain for the Northside has been increased to 2x10000 since half of the light is now being used by the OOL PD.

Today, I added a new out-of-loop transmission PD (Thorlabs PDA10CS) for the south path. This will be helpful in future measurements of RIN coupling to beatnote noise. This PD is added at (1, 40) using the dumped light. The optical layout would be updated in a few days. I've confirmed that this photodiode is reading the same RIN as read earlier in CTN:2555. I've also connected Acromag channel for South Transmission DC to this photodiode, so the transmitted power channels and the mode matching percentage channel of South Cavity are meaningful again.

I measured dark noise of the beatnote detector reaching moku and its effect on measured beatnote frequency noise.

Measurement

• I used Andrew's low noise preamplifier to measure this noise.
  
• I first took the transfer function through the preamplifier so that everything later can be referenced back to its input.
• I measured measurement setup noise by shorting the input to the pre-amplifier.
• Then I measured dark noise after the 20 dB coupler (the cable that I connect to moku) when the detector was completely blocked with a beam dump.
• The noise was converted to precieved beatntoe frequency noise by using peak to peak voltage of beatnote signal. (Noise*freq*2*pi/Vsigpkpk)
• The noise came a little less than Tara's measurement. I'm updating my noise budget model with this now.

Data

With comparison with old estimates:

CTN_Latest_BN_Spec_With_OldEst.pdf

CTN_Daily_BN_Spec_With_OldEst.pdf

Usual noise budget links:

CTN_Latest_BN_Spec.pdf

CTN_Daily_BN_Spec.pdf

Adding more specifics:

Discrepancy #1

Following points are in relation to previously used noisebudget.ipynb file.

• One can see the two different values of effective coating coefficient of thermal expansion (CTE) at the outputs of cell 9 and cell 42.
• For thermo-optic noise calculation, this variable is named as coatCTE and calculated using Evans et al. Eq. (A1) and Eq. (A2) and comes out to (1.96 +/- 0.25)*1e-5 1/K.
• For the photothermal noise calculation, this variable is named as coatEffCTE and is simply the weighted average of CTE of all layers (not their effective CTE due to the presence of substrate). This comes out to (5.6 +/- 0.4)*1e-6 1/K.
• The photothermal transfer function plot which has been used widely so far uses this second definition. The cancellation of photothermal TF due to coating TE and TR relies on this modified definition of effective coating CTE.

Following points are in relation to the new code at https://git.ligo.org/cit-ctnlab/ctn_noisebudget/tree/master/noisebudget/ObjectOriented.

• In my new code, I used the same definition everywhere which was the Evans et al. Eq. (A1) and Eq. (A2). So the direct noise contribution of coating thermo-optic noise matches but the photothermal TF do not.
• To move on, I'll for now locally change the definition of effective coating CTE for the photothermal TF calculation to match with previous calculations. This is because the thermo-optic cancellation was "experimentally verified" as told to me by Rana.
• The changes are done in noiseBudgetModule.py in calculatePhotoThermalNoise() function definition at line 590 at the time of writing this post.
• Resolved this discrepancy for now.

Today I ran the two codes with the same parameter values to check if the effective reflectivities calculated during the calculation of thermo-optic nose matches. They do match exactly actually. Attached is an over plot.

Coating effective coefficient of thermo-refractive effect comes out to be:

Old Code: 8.61(46)e-05 K**-1

New Code: (8.59+/-0.21) e-05 K**-1

So this discrepancy was not really there. I was just comparing apples with oranges earlier.

Also, I found that the effective reflectivities from the top surface of layers calculated in Evans et a. PRD 78, 102003 (2008) were different from reflectivities calculated by matrix method in Hong et al. PRD 87, 082001 (2013). It turned out that the sign before phase in Eq. B3 and B4 in Evans et a. PRD 78, 102003 (2008) were opposite to what comes from the matrix method. After discussions with Gabriele, I came to the conclusion that this sign only creates a difference of giving complex conjugates of effective reflectivity. But to have consistency, I have corrected the sign of phase in the new code. Attached is a comparison of old effective reflectivity and the new one. One can see that the imaginary values are opposite in sign but everything else matches perfectly. Also, this has no effect on the effective coating coefficient of thermo-refraction.

My goal was to investigate the effect of placing a 1k ohm resistor as the input of the DC port of the Bias Tee. The expectation was it would decrease the bandwidth around DC, and this is supported by LTspice simulation. According to simulation, at around 300 Hz there becomes a few dB difference between having the resistor and not having it. From the paper 'W. Zhang, M. J. Martin, C. Benko, J. L. Hall, J. Ye, C. Hagemann, T. Legero, U. Sterr, F. Riehle, G. D. Cole, and M. Aspelmeyer, "Reduction of residual amplitude modulation to 1 × 10^-6 for frequency modulation and laser stabilization," Opt. Lett. 39, 1980-1983 (2014)'(https://www.osapublishing.org/ol/abstract.cfm?URI=ol-39-7-1980) in Figure 2B we see the frequency noise without servos active. The noise falls off after around 50 Hz, which indicates the change of bandwidth to 300 Hz should not be an issue. I also showed the effect for 500 ohms to see how the transfer function changes intermediately between 0 and 1k ohm resistances.

Going back the original issue of scattering, it appears that there is light being back reflected from somewhere in the post PMC path but before the reference cavities.  

Reducing number of optics after the North PMC 

I had installed a bunch of polarization optics before the north 14.75 MHz EOM in an effort to reduce RFAM (see attachement 1).  It looks like stuffing so many optics in such a small space is a bad idea.  You can see weak retro reflected beams from the wave plates and, probably, the PBS as well.  The short propagation distance makes it difficult to angle optics enough to be able to separate them from the main beam laterally to dump.  The EOM can't really be moved because the mode matching solution is a little tight for the available space.

After talking with rana and Craig yesterday it seems like the Pre Mode Cleaner (PMC) should be filtering polarization well enough when locked that the PBS and quarter-wave plate (QWP) are unnecessary. I removed all but the half-wave plate (HWP) and checked the residual polarization on transmission with a diagnostic PBS in place. I found 2 µW of power out of 1.2 mW was remaining when tuned all the way to s-pol.: this is a 1:600 extinction ratio which is about what we would expect from such a beam cube.  This measurement may be biased by the lower limit of the power meter, PBS should be giving 1:1000.  

I moved the PBS to before the PMC to clean up light out of the 21.5 MHz PMC phase modulator. The only optics in the post PMC-> EOM path are now a lens, a steering mirror and a half-wave plate (see attachment #2).  After realigning the PMC cavity and the north refcav I was able to reduce the RFAM to -55 dBm, which is good enough for now.  These slight changes in RFAM level mean that the FSS offset will need some adjustment.  I was unable to see any improvement in the beat spectrum as the beat note had drifted down to 2 MHz.  I turned the heating down a small amount and left it overnight to settle.

I didn't angle the HWP or lens by that much, this shouldn't be necessary because the PMC is a traveling wave cavity.  The elements should be pretty close to normal.  The glass beam dump should be checked to ensure it is not clipping any retro-reflected beams on the rough edge of the glass.

Clamping down the PMC

I never clamped down the PMC. It is just sitting on the ball baring points. This isn't great.

When I realized the tapped holes on the side of the base I went looking for clamps.  They are pictured in attachment #3 but they do not fit.  It turns out there were some issues with the choice of ball bearings on which the PMC sits.  The ball barrings sit over holes so that the PMC when placed will realigned exactly with its previous position on the base.  Antonio had found that the holes drilled for the ball barrings were spec'ed a little too big.  For standard increments of bearings size the closest size fits nicely over the hole but under force they actually slip down into the hole and are almost impossible to get out.  He bought the next ball bearing size up. However, this means that the clamps no longer reach the full height PMC assembly.  The assumed tolerances were made too tight on all these components, the next edition of drawings should allow for some wiggle room.

The drawings should be updated with at least 1-3 mm of range on slot cut side pieces for the clamps so there is room for changes in height due to ball bearing size.  Possibly even more, if future people want to put Viton or Sorbothane dampening into the clamping. The non-tapped holes should also be changed to through-all. Or at least drill with a narrower diameter through-all. This will help future users poke out objects that get stuck in the holes.  

For that matter the design of the clamps seems wrong.  There is a bar that goes over the top that is fixed with a slot-cut piece affixed to each side. This is intuitively wrong as the bolts all go in horizontally when the clamping force needs to be applied is downwards!  It means that the clamps are locking a vertically applied force from the sides; to bolt the PMC down you need to apply force to the bar and tighten the bolts at the same time for two different clamping bars.  The screws should have at least one vertical pair on each clamp so that tension can be applied in the same direction as the clamping force.  

PMC documents on the DCC

For future reference, here is a list of all the PMC documents on the DCC:

• LIGO-E1400332 (PMC body and end cap drawings)

• LIGO-D1600172 (PMC clamp side piece clamp Stud)

• LIGO-D1600171 (PMC bar clamp)

• LIGO-D1600170 (PMC base) 

• LIGO-E1400241 (PMC flat beam splitter specification)

• LIGO-E1400241 (PMC curved mirror specification)

Evan's technical note for PMC design considerations: LIGO-T1600071.

I can't find assembly procedures on the DCC.  There was a report from one of Kate Dooley's summer students, LIGO-T1600503, that shows a jig for gluing the PZT. 

Today I made a standalone Adjustable TTL Trigger generator box. Following are some features:

• The rising edge of output can be used to trigger all TTL compliant external trigger enabled equipment (oscilloscopes, HP4395A, SR785 etc.)
• Uses AD620 at very high gain (G=5k) to create a comparator. Positive input is the signal and negative input is controlled with a 100 kOhm potentiometer.
• The threshold can be set anywhere between +18V and -18V.
• Powered internally with four 9V Alkaline batteries and has a switch to turn ON/OFF. Should last really long.
• The output is through a 4.7 V Zener Diode which activates in reverse bias when the signal becomes 140 uV higher than the set threshold.
• Input impedance equivalent to AD620.
• The trigger is not latched and output goes to -0.7 as soon as the signal falls below the threshold.
• Has a window port for checking the threshold.
• The practical way of operating would be to see the signal and output of the box together in an oscilloscope first and fine tune the potentiometer to see triggers at the right point.

The common error signal on the TTFSS has a 5 mV offset, which was causing the loop to catch on the edge of the error signal, near the sideband. I've adjusted the offset pot on the TTFSS interface board from 502 to 960 to remove this offset, and the loop now catches only on the carrier.

Also, I've taken the DC path from the south cavity RFPD and plugged it into an SR560 with gain 10 and then into C3:PSL-RCAV_FMON. This is temporary, and I've done it so that I can remotely lock the south cavity more easily for the gyro beat measurement. With the gain of the SR560, refl on resonance is about 2 V at minimum.

I'm thinking about the spec for AlAs/GaAs coatings. Here is the list of what I have:

• coating on concave side of the mirror for 0.5m x6 (I'm not sure if they can do the transfer on 0.5m mirror now) for 1.0m x6 for flat mirror x3 -
• for circularly polarized light, normal incidence
• Transmission @1064 = 100ppm +/- 10ppm. 10% error is still within the acceptable value for 10ppm loss (T ~ 67-73%), see T1200057v11 -
• Absorption + scatter loss < 10ppm, this is what Garrett told us. -
• coatings diameter = 8mm (The number is from Garrett), the loss around the edge for our beam with diameter=364 um is less than 10^-10 ppm. -
• Max scratch surface and point defects are not determined yet. I can look up the specs from our current SiO2/Ta2O5 mirror since they are ok for us. -
• I think we are aiming for the thermo-optic optimized coatings. The layer structure can be found in T1200003-v1.

==Coating diamter for 0.5m ROC mirror==

About the coatings diameter, Garrett said it depends on the aperture size/ coating diameter. So I made a plot to estimate the loss due to the finite size coating vs Coating diameter for our spot radius of 182 um. The loss is simply calculated by the ratio of the power not falling on the coating = Ploss/Pin = (exp(-2*r0.^2./w0.^2))*1e6*26000/pi   

where r0 = coating radius, w0 = spot radius, a factor of 1e6 for showing the result in ppm, 26000/pi is the total loss due to the light bouncing in the cavity.

fig1: Loss vs coating diameter (in meter)

It seems we can go to 2mm coating diameter, and the loss is still much less than 1ppm (the expected loss from absorption and scatter is ~ 10ppm). However, we have to consider about how well they can center the film, how well we can assemble the cavity. So larger coating diameter is always better. If we assume that 1mm error is limiting us, coating diameter of 4-5 mm should be ok for us.

 ==for mirror with 1m ROC==

If the ROC is 1.0m, the coating diameter can be 8mm. For the cavity with 1.45" long, the spot radius on the mirror will be 215um (182um with 0.5m mirror). This changes the noise budget of the setup a little bit. The total noise level is lower by a factor of ~ 1.2. (see below figure) at 100 Hz.

fig2: Noise budget comparison between setup with 0.5 m and 1.0m RoC mirrors, plotted on top of each other. Noises that change with spotsize are coating brownian, substrate brownian, thermoelastic in substrate, and thermo-optic.

==What do we choose? 0.5m or 1.0m==

For both 0.5 and 1m, the cavity will be stable (see T1200057-v11, fig11). So either choice is fine

if we use 1.0 m,

• we loss the signal level a bit,
• but we are more certain that the coating will work. 
• The procurement should be faster (as promised by Garrett)
• have large area coating up to 8mm diamter
• need to check if we can mode match or not (I'm positive that we can, but I'll check or let Evan check)

So at this point, I'm thinking about going with 1.0 m mirror.

We should be able to mode match into a cavity with 1.0 m ROC mirrors using only the optics we already have on the table. 

Current mirrors: 0.5 m ROC (has -1114 mm FL)

• 370 um PMC waist at z = 0 m = 0 in
• 229.1 mm FL lens at z = 0.203 m = 8.0 in
• 209 um intermediate waist
• 85.8 mm FL lens at z = 0.923 m = 36.3 in
• 41 um intermediate waist
• 143.2 mm FL lens at z = 1.201 m = 47.3 in
• 182 um cavity waist at z = 1.944 m = 76.5 in
• Mode overlap 1.000

Proposed mirrors: 1 m ROC (has -2227 mm FL)

• 370 um PMC waist at z = 0 m = 0 in
• 229.1 mm FL lens at z = 0.203 m = 8.0 in
• 209 um intermediate waist
• 85.8 mm FL lens at z = 0.889 m = 35.0 in
• 45 um intermediate waist
• 143.2 mm FL lens at z = 1.166 m = 45.9 in
• 210 um cavity waist at z = 1.952 m = 76.8 in
• Mode overlap 0.999

The various waists for the proposed mode matching are equal to or larger than the waists for the current mode matching, so I don't think we should be any more worried about sensitivity than we already are.

I'm computing  coating Brownian and thermo optic noise (TO)  in AlGaAs coatings using GWINC code to compare it with the result reported by G Cole etal. Brownian noise from my result is similar to theirs, but TO noise is still not correct.  I'm working on it.

==Background==

We have talked about what kind optimizations should we go for AlGaAs coatings in order to minimize TO noise. There are two choices for us to consider

1. using the layer structure as proposed in T1200003, or
2.  adding a half wavelength cap on top of the quarter wave stack coatings as suggested by the authors.

Since the second option is more desirable in terms of manufacturing because of its simplicity, I decided to check if it really can  bring TO noise below coating Brownian noise. If it is true, we can use it for our mirrors.

 ==calculation==

• I use GWINC code for TO noise and brownian noise calculations to verify the result if they are agree or not.
• Materials parameters used in the calculation are taken from the paper. But most of the coatings material properties of an individual layer of AlxGa1-xAs are not provided. There are only the average values of thermal expansion, heat capacity, thermal conductivity, dn/dT.  There are refractive indices (nh/nl = 3.48/2.977) and layer structure (81 layers, starts with nh, ends with nh). So, as a start, the values for high index material and low index material are the same as the averaged values.

==result==

• My Coating thermal noise level is 8.4*10^-35 m^2/Hz while their result is 9.8 *10^-35 m^2/Hz, @ 1Hz. This is not very bad, since there are some differences in the formulas between GWINC and their calcualtion.
• However, my TO is off the roof, almost 2 orders of magnitude above their result. I'm checking if it is because of the code is wrong(typo in the parameters) or the fact that I used all the averaged values.

(I'll add more details about the calculations later)

[matt, tara] Got Al_xGa_1-xAs material parameters from Matt Abernathy. I plug the numbers (all in SI) in GWINC, but the result is still not quite similar to that in Cole etal paper.

 ioffe has materials parameters for TO noise calculation.

Specific heat: 0.33+0.12x J/gK
 Mass density rho = 5.3165-1.5875x g/cm^3
Thermal conductivity,kappa: 0.55-2.12x+2.48x^2 W/cmK  (There is also thermal diffusivity = kapp/(rho*specific heat) [m^2/s]. The results are the same)
Thermal Expansion: (5.73-0.53x)·10-6/K

dn/dT: 3.66-2.03x *10^-4/K
This is from a paper, "Thermal dependence of the refractive index of GaAs and AlAs measured using semiconductor multilayer optical cavities", by Talghader and Smith. Keep in mind that this paper has an important Erratum if you want use values from it.
Unfortunately, this paper measures dn/dT at a max wavelength of 1030nm, so it's not quite accurate, but probably good enough.

Note:

One of the variables in GWINC code is ThermalDiffusivity. But the numbers used in previous TO plot is thermal conductivity of materials. I'll check the TO calculation codes and see if it is just a naming error, or the calculation is actually wrong.

I used GWINC code to calculate TO noise in AlGaAs coatings, with some modifications to the code I can get the result that is comparable to Cole etal's result. However, there seems to be some minor details that I have to check. The half wave cap solutions for TO cancellation is not verified by the current calculation yet.

 What I modified and checked in the code:

• The variable called thermaldiffusivity in the code is actually treated as thermal conductivity in the calculation, so all calculations in the past are still correct.
• The layer structure in the code was originally for Ta2O5/SiO2. The first layer started with SiO2 (low index material,nL), and ended with Ta2O5(high index material,nH) at the substrate surface. However, the AlGaAs coatings start with nH and ends with nH. I changed the calculation for effective thermal expansion accordingly. With the correct layer structure and materials parameters from Matt, the TO nosie is closer to JILA's result. However, the shape is still not the same, what reported in JILA is almost flat across 1-100 Hz. The calculated transmission from the layers is 1.8 ppm, but the paper says 4ppm. I'm looking into this.

Above figures: top plot is the result from GWINC. Its title should be Al0.92Ga0.08As coatings, not SiO2/Ta2O5, bottom picture is taken from Cole, etal. TO noise crosses coating brownian noise around 3 Hz for both plots, however the slope is very different. NOTE: the y axes are in Hz^2 / Hz.

As a quick check for the proposed half wavelength cap solution to reduce TO noise, I modified the layer structure and computed TO noise. Since they did not mention what kind of material for the cap I tried:

1. 81 layers, starts with nH, ends with nH, the first layer is 0.5 lambda thick. This is not working.
2. 82 layers, starts wit half wave nL, followed by the original 81 layers. This also does not work. Both cases have comparable TO noise, but transmissions are different.

I'll check their formula and GWINC to see where the differences are.