I made a more advanced model of the circuit using Zero and compared them to the experimental results.
I made the following changes to SN009:
I repeated this measurement to compare later after changes to RFPDs.
Data and Analysis
For the convenience of others, I'm summarizing the open questions I asked on elog in December. Comments on the posts, advice or answers to my questions would be nice.
Edited on Mon Dec 23 15:21:50 2019 .
I took some TF measurements, but I'm not sure if I used the right method to do this. All the resutls look essentially the same, so maybe I'm just measuring instrument noise and nothing else. Regardless, I' posting the results.
For both measurements, I sent the source signal, first directly to PZT or EOM and second with a 50 Ohm termination in parallel. In case of EOM, I also did another measurement with low power to see if I can uncover any saturation effects my high powered source might be causing.
The output of two photodiodes, Thorlabs PDA10CS at (110,44) at dumped end of a PBS before PMC and at (64,28) at dumbed end of INput PBS of Faraday Isolator after PMC was measured and I took DC level of the outputs averaged over 10 s right after taking TF measurement. This was divided by the measured TF to get units of 1/V i.e. from PZT/EOM to RIN.
The parameter configuration files for the measurements are in the data directory.
What might have gone wrong:
I suspect that maybe AG4395A is unable to drive the capacitive load of PZT and EOM after a certain frequency. I need to find a better way to actuate the PZT and EOM in a known fashion, possibly through the FSS box or some buffer driver.
In the case of EOM, I'm also uncertain if the amplitude of actuation would be enough to do anything whatsover. Maybe the transfer function needs to be taken with high voltage driver.
Any comments on my measurement techniques are welcome as I'm surely not doing this right.
Rana's question about PMC:
I concluded that PMC is causing the 433 kHz peak because the only time I do not see it in CTN/:2502 is behind PMC when FSS is OFF. I couldn't think of a way to check if PMC Servo is causing it on its own. Can I do that without closing the loop somehow?
Could be that either
I suggest measuring the RIN/PZT and RIN/EOM TFs, both before and after the PMC.
Is the PMC servo making the 430 kHz peak?
Today I inserted two photodiodes, Thorlabs PDA10CS at (110,44) at dumped end of a PBS before PMC and at (64,28) at dumbed end of INput PBS of Faraday Isolator after PMC. These photodiodes were set at 20dB gain which according to the manufacturer gives us a bandwidth of 1.9 MHz.
Then, I just took Noise Spectrums using AG4395A with the parameter file attached in the data directory, from 0 to 1 MHz at 1 kHz ResBW and 50 averages. The output was divided by the DC level of the photodiode which was measured using TDS 3052B averaged over 10s. This measurement was done with FSS ON or OFF.
North Laser is hereby acquitted.
After the last calibration, South Laser power at Laser head determined by its internal photodiode and show as setting PWR today (noticed today) dropped to 80 mW with none of the other settings changed. The diode current is still the same, so this could be one of the following two things:
1) The internal circuit to measure laser power at head went wrong.
2) Something is wrong with the laser crystal.
The laser power as seen by the reflection photodiode at South PMC captured a glitch on Dec 17th, 2019 8:30 am (might be 7:30 am in PST). The glitch shows that PMC went out of lock due to it but then returned to a laser power lower than before. The voltage level decreased from 2.72 V to 2.3 V after lock, which is equivalent to a drop of ~15.4%. The laser head power level according to PWR monitor dropped from 101 mW to 80 mW, which is equivalent to ~20.8%. While these don't match, PMC reflection is also not a true measure of power level. But I am sure this power has actually reduced and even if the laser head power meter is off a little bit, we have witnessed a big drop in power.
Restarting the laser (Put it on Standby and switch it back on) didn't change the power level.
Restarting the laser with turning the key off and on also didn't change the power level.
Even though 80 mW is enough for our experiment, this sudden decline in power shows there is something happening with the laser that we do not understand.
If we are going to keep the same laser with busted Noise Eater, should be go in the direction of implementing the ISS?
In Oct, 2018, I along with Johannes developed ISS boards and photodiode transimpedance amplifier boards. These were characterized by Johannes at Cryo_Lab:2180 and Cryo_Lab:2181. However, Johannes was using AOM while we have EOAM here. There are slight differences which should be addressable given we have an offset port also in the ISS board. But the expected open loop gain needs to be worked out along with right choice of transimpedance and cavity pole neutralizer stage in ISS.
Johannes gave me his M2ISS photodiode mounts before leaving, so we have them too. This looks like a 1-week project from start to finish. So with the thumb rule of "nothing goes as expected" and me being a grad student, this should take 2 weeks. Is it worth it? Should I start working on this? Or can I just get a laser with working NE?
Also, there is no real documentation of why we stopped using the existing ISS in CTN? Last mention of ISS is at CTN:2132 and after that I came and was working on ISS boards which we never installed in CTN since or FSS, RFPDs and thermal controls were priority then.
As suggested few group meetings ago, I took time series data at the input and output of boost stage opamp U7 (at TP15 and TP16) using TDS 3034C. I know trusting oscilloscope for synchronous measurement of two channels is not a great idea, but this is a good zeroth order approximation for this approach. 500 MHz 10MOhm impedance probes were used for the measurement of the signals. The data is taken at two different acquisition rates and I used basic math to calculate the expected ideal output. Following formula was used:
Ofcourse the offset point above wouldn't work. Also, I have not included the notch filter that is also present at this stage.
The results of this analysis are attached. I also, calculated the required slew rate of the AD847 at this position. At the 100 MSa/s sampling rate, we saw that maximum required slew rate for ideal signal was around 110 V/us, much less than rated 300 V/us. I don't see a bad change in shape of the waveform. At 1 GSa/s sampling rate, we see that required slew rate reached about 220 V/us at indeed oscillations at high frequencies are limited even though this is below the rated value.
Looking from data taken at CTN:2474 present in this directory, these fast oscillations that we are seeing on the oscilloscope are around the modulation frequency 36 MHz. These leftover downconverted 2-Omega from teh RFPD might be saturating slew rate limtis of many opamps in the TTFSS box. I think I should look back into Andrew's implimentation of the elliptical filter right after the mixer in the RF board. He did say that we need to make the filter lossy to ensure the reflected signal at 36 MHz gets absorbed at a 50 Ohm resistor. For this we need to add a resistor to gnd at node rfbn2. Seeking permission to make modifications to board.
In talks with Anchal, we have adapted our model and the transfer function fits rather well. These measurements were taken with the AGmeasure and SRmeasure script on the larger instruments in the CTN lab (I can't remember the exact make of them). Two were used for different frequency bands: one up to 100 kHz and the other one above. In order to have continuous measurements, the high-frequency measurement was shifted to be continuous with the lower frequency measurement. This was not a significant shift and does not impact the overall interpretation. It is merely a correction for different calibrations in the Instruments used to measure the transfer functions.
The Magnitude of the transfer functions matches well and Anchal agrees that they also look good. So the circuit can be used in its current form. It isn't currently in a box but that is an easy fix. I am still making full documentation on it and that should be done over break. I am also looking to make it as a single PCB to eliminate any points of failure that come from attaching multiple PCBs together. It is not entirely necessary because as is seen in the graphs, the circuit works well but designing a PCB is a good task for over break because it does not really require any in lab work and will be highly beneficial down the road. I am still talking to Anchal about the circuit and its implementation.
I rechecked the 20dB coupled RF output of SN009 (RFPD on north Cavity reflection) and SN010 (RFPD on south Cavity refleciton). The following are the mean values over 10s taken at DC coupling with 50 Ohm input impedance, +/- 40 mV input range with two different oscilloscopes.
Note that these are 20dB coupled values, so the actual offset reaching the FSS board is about 70 mV. The PD input is AC coupled through a transformer so it shouldn't be reaching further than there but ideally MAX4107 at the RFPD is supposed to have a maximum input offset voltage of 3 mV which at gain 10 should look like 30 mV. So what we are seeing (7.0 mV) is more than twice the rated maximum input offset. I'm not sure if this means our MAX4017 is busted or something is wrong in the loop. Help needed in understanding this result.
I found the latest available photos for the Cavity Reflection RFPD circuits.
There were minor changes made after taking these photos which are logged at: https://nodus.ligo.caltech.edu:30889/ATFWiki/doku.php?id=main:experiments:psl:rfpd
I'll update these above photos whenever the next time I get a chance. They would be present at:
I took beatnote spectrums in the current modulation index (was set to around 0.3 earlier). Then I took spectrum after attenuating the modulation signal power to both EOMs on North and South Path by 3 dB and 6 dB. This should reduce the modulation depth by and 1/2. After every change, a small displacement happens in the thermal control of the cavities, so I had to wait for some time to let it settle. The gains of the FSS loops were kept constant to make sure only the modulation depth is the parameter that is changing. Gain values were 24 dB and 16 dB for South COM and FAST gains, and 11 dB and 10 dB for North COM and FAST gain.
please overlay all the plots and also include the ideal TFs - its hard to understand anything without the comparisons
Testing the circuit using SRmeasure and AGmeasure, I got more accurate readings of the transfer functions. The PZT path TF looks good. The EOM path looks good up until 105 Hz, then it doesn't drop like it is supposed to. My next task is to fix the high frequency EOM path. These graphs are a work in progress. I will include plots with the ideal TF at a later time.
I put laser settings on both North and South Cavities back to default. From this point onwards, all settings about the lasers would be known and kept track of. The red values are the settings that were changed.
While turning the nominal diode current of south laser all the way clockwise, I found that the laser power peaks before the maximum diode current is reached. This diode current is about 1.9 A. This is unexpected. Any explanations on this would be helpful.
First, off I forgot a resistor on the previous diagram so here is the fixed version. Also, Anchal said that the EOM noise is probably caused by instrument noise at the low end. The phase response should be flat on both and on the PZT path the non-flat response is caused by some high pass filter caused by AC coupling or something of the sort.
I am still trying to figure out how to get a flat response on the phase. Anchal did some math that looks at the possibility of it and I am going through it.
I'm trying to think hard with my small brain how the distortion would affect the PDH functioning and inject noise in the frequency of the laser. I have a line of reasoning which starts with a question.:
Of course, all this depends on the RF sidebands interfering constructively upon reflection. I remember (I don't know from where) that it is the opposite. Either there is a fault in my calculations or this is indeed what is happening. I need to understand this properly to go further. Need help.
I tested the circuit and got the attached results. The PZT transfer function looks reasonable. some adjustments to the values of the resistors in the circuit should have the match up. The phase is still out of alignment for both. The transfer function of the EOM looks far from the model. I suspect that this is due to a bad solder point or poor choice of components rather than the circuit's design.
when the NPRO crystal is oscillatin
no tellin what is relaxin
I took a spectrum of PMC error signal when the FSS loop is not closed. This should provide a rough estimate of the free running laser noise. We had earlier seen a peak at 435 kHz in the Northside, hence I wanted to take this data with some references. First of all, this peak is very similar in the description of relaxation-oscillation peaks of these NPRO lasers mentioned on page 52 of this manual. The "Noise Eater (NE)" is supposed to suppress this peak significantly. However, in the spectrum of the PMC error signal, there is no difference when noise eater was ON or OFF.
I took a spectrum of Southside as well, just to see if I could see action of Noise eater there. For south laser, the noise eater suppressed noise only till 100 kHz or so and probably this side also has a similar relaxation-oscillation peak problem but is shadowed by a large feature at 30 kHz. Not, the absolute value of the spectrum between North and south are vastly different due to different amount of light, different transimpedances od the PDs and different gain values in the feedback circuit.
However, the noise eater is supposed to reduce relative intensity noise only. And the error signals of PMC should really be telling us noise in the frequency of the laser. So maybe I'm connecting two dots in different Hilbert spaces. But Rana suggested that a busted Noise Eater could be the reason for the 435 kHz peak, I just do not understand how RIN would cause frequency noise so badly. I thought photothermal transfer functions from RIN to frequency noise were very small.
This has happened few times now that acromag channel for the can heater driver stopped updating according to the PID script and the can gets heated to a very high temperature. This pushes the temperature out of the ranges of the current AD590 temperature sensor board. I have changed the range of channel 2 (this was being used for out-of-loop) to ensure we can still see some meaningful temperature value when such incidents happen. I have replaced R18 from 100k to 27k. The updated table is:
The new circuit design has been built and is shown below. I tested the transfer functions and the PZT path looks good by eye but the EOM path looks like there is a solder point that is bad. I will show the graphs when I have them done.
With discussions with Anchal and some reading
It seems impossible to create a casual circuit with a zero phase shift. (See this for more)
If we have a circuit with an impulse response h(t) and transfer function H(f)=F[h(t)] where H(-f)=H*(f). For the filter to cause no phase shift then ∠H(f)=0 for a complex exponential input for all f. It is also impossible to have a constant phase shift unless that phase shift is zero.
Therefore "filter does not change the phase at all, then H(f) is a real-valued function, and because of the conjugacy constraint, it is also an even function of f. But then its Fourier transform h(t) is an even function of time, and thus the filter cannot be causal (except in trivial cases): if its impulse response is nonzero for any particular t>0, then it is also nonzero for −t (where −t<0)"
Because this can't be done casually, it should be done using a Field Programmable Gate Array. Unfortunately, I don't think we have access to one. I am reading up on the Moku FIR Filter builder to find out if we can program it to do what we want.
I have designed a passive circuit that seems to match the ideal transfer functions in shape. Scaling should just be a game of playing with the values of the resistors and capacitors. The phase still seems to be an issue. There is an unwanted phase shift from 0 -> -90.
The next step is trying to finalize the values for the resistors and caps. Possibly model in zero if I have time. Then build and test. Also fix the phase.
I have updated the plant model to contain the cavity pole also. Cavity pole is a pair of positive and negative real poles, so it is hard (or maybe impossible) to imitate it exactly with an electronic circuit. Or maybe, my analysis is wrong.
Nevertheless, I have for now made this circuit which has a second-order pole, so it correctly matches the magnitude of the model transfer function up to 1 MHz for both PZT and EOM paths. Note that the elliptical filter is not included in this as we can connect the circuit to Test port 1 which injects just before the filter in LIGO-D0901894. Also, for the gains in EOM path, I had to add some factors to make it the same as the model transfer function. All components are calculated for E12 series resistors and capacitors.
Attached is a pdf of the notebook which contains all the mathematics in latex and a zip file with all files to recreate and further work on this. Ian can use these as support to learn zero further.
I doubt it.
When feedback loops be noisy and busted,
the swept sine TFs cannot be trusted
So, we can say with some confidence that the 435 kHz signal seen in the oscilloscope in CTN:2482 at TP1 is actually due to some non-linear effect most probably
I quickly took a high-frequency Open Loop Gain measurement of NFSS loop at 10 dB COM Gain and 10 dB FAST gain, using the same measurement method as in CTN:2443. The UGF has not changed much but there is a dip at 435 kHz. This was there before too, I was just not paying enough attention to this part of OLTF before. So, we can say with some confidence that the 435 kHz signal seen in the oscilloscope in CTN:2482 at TP1 is actually due to some non-linear effect most probably and does not get suppressed at all. The phase margin near UGF looks about 135 degrees so there is no solid reason to believe this could be due to loop oscillation.
So I got to think of what combination of RF frequencies might be mixing down to create this oscillation and where. This oscillation is also visible in Plot 6 and 7 of NFSS_RFPD_Output_Oscilloscope.pdf of the measurements done in CTN:2470.
I took time-series data at TP1 on NFSS. This is just after the elliptical filter which is after the demodulation on board.
As also seen in the spectrum measurement at this testpoint in CTN:2474, there is a lot of power at around 435 kHz. But this is not noise!
As seen on the oscilloscope, this is a near-perfect sinusoid. So this must be either of the following:
This measurement was taken with a 500 MHz 10x Probe with a 300MHz TDS 3034C oscilloscope at 0.5 GSa/s sampling rate.
Interestingly, there is no such oscillation or peak on the South side. However, the south sides COM Gain is 24 dB and Fast Gain is 14 dB. So it could be because it is just suppressing this non-linear effect properly or just has a very high UGF.
I did some theoretical calculations using the datasheet value of second harmonic SFDR from MAX4107 and the transfer function I measured from Test IN ports of RFPDs (using 100 kOhm series resistance).
Edited Wed Nov 20 14:43:07 2019: Corrected an error in code.
cool - how bout use this new no-how to estimate the excess in FSS?
In line with industrial practices, I did two tone third order intermodulation test today on the FSS RFPDs. This test was inspired by procedure described in this technical note by MiniCircuits and this paper at IEEE.
Datasheet for MAX4107
I calculated these values by integrating in the 8 MHz neighborhood around the marked harmonic peak, the power spectral density using the frequency at the point as the lower edge of the bin. Slew rate is calculated by multiplying the rms voltage level with the frequency and the fraction is calculated against the datasheet value for Max 4107.
SFSS RFPD Output Slew Rate Usage (MAX 4107, SR: 500 V/us)
These calculations at least show that MAX 4107 should be much far away from reaching its slew rate limit in both RFPDs.
But what now?
I've wrote this script, nonlinTF.py which controls a Marconi 2023A and SR785 together. Marconi is used to providing a carrier frequency which is mixed with the Source Out signal from SR785 before feeding into the TEST2 input port on D040105 of TTFSS boxes. Then OUT1 port on D040105 of TTFSS box is used to read back at channel two of SR785 (channel one being fed with a copy of the Source Out signal). So SR785 is effectively measuring any downconversion in the loop (due to some nonlinearity) from micing of CF-IF, CF and CF+IF probe signals injected into the loop. The effectively closed-loop transfer function between TEST2 and OUT1 should be G/(1+G), so this injected signal should not suffer any suppression, nor should it affect the locks. The locks were maintained without any problems during the whole measurement. The CF frequency was stepped by 100 kHz from 100kHz to 10 MHz and then by 1 Mhz upto 100 MHz.
Mixer ZX05-1LHW (level +13 dBm) was used for the mixing and IF peak voltage was set to 30 mV. The configuration of the measurement for the transfer function is present in the configuration file in the folder.
I've borrowed a Marconi VCO, a mini-circuits LPF and mixer from the CTN lab for use at the WB EEshop.
AG Mon Dec 30 10:24:28 2019 : Has been returned to CTN now.
South Common Gain: 24 dB! , Fast Gain: 14 dB
North Common Gain: 10 dB, Fast Gain: 10 dB
I raised the North TTFSS box by 6 inches to make way for working on South box and to reduce the congestion of connectors in front of the two boxes. I have also clamped the boxes to a fixed position now, so they can't move. This would ensure the cables are not hitting the end of the platform and face any severe strain.
The next step towards improving lab cable hygiene and layout is to replace all RF cables with RG-405 Belden-N 1671J cables. However, the effects of this change would be less significant then fixing the sick FSS. So I'll first focus on that.
Rana told me that in 40m, the PMCs are autolocked by reducing the gain of the loop and changing the ramp until the lock is acquired. Then the gain is increased back to operation point. I tried this method with our South PMC as the usual method being used of 'changing Blanking state' wasn't working anymore. However, even with the gain set all the way to -10 dB, the loop was not locking exactly at the center of the TEM00 mode. And was unable to skip higher-order modes. There is a header H1 which changes the input stage gain. Removing this header pin, reducing the input stage gain by a factor of 100. Even after doing this, I was unable to robustly acquire the lock by this method. Besides, this reduced gain was the case earlier (CTN/2427) and it was too low as the VCA U5A AD602 had to be kept at maximum 30dB gain. So I did not want to reduce this first stage gain.
Somewhat similar to our FSS loops, I find it much cleaner to just not close the loop until we have reached near the lock point. This could be done fairly easily with the existing code. I just had to change the loopStateEnable variable from Engage (which changes the Blanking pin on U5A AD602) to input switch (FP1TEST for South and FP2TEST for North). So now, when finding a lock point, the input is changed to terminated inputs and the loop is closed when lock point is found. This works very nicely, just like the FSS autolocks.
This has finally fixed any problems with PMC autolocks.
First step in FSS Diagnostics was to see RF output from the RFPDs in FSS when they are locked. I ran some extensive measurements to cover all the information about this signal. The RF out is sent to the FSS box through ZFDC-20-5-S+ 19.5 dB directional coupler. The coupled output's spectrum is measured at different frequency ranges using both AG4395A and SR785. The measurement configuration files are included with data for metadata of the measurement. The signal is also analyzed in time series with measurement upto 1 GSa/s with TDS 3034C and one measurement at 5 GSa/s with TDS 3052B. Both measurements were done manually setting minimum possible voltage resolution and using DC coupling with 50 Ohm impedance. All data is attached raw here for now. More interpretation and analysis to come soon.
It turns out I did not have a full understanding of the problem and it was not really a problem. The blanking (pin 4) on U5A AD602 doesn't shut down the channel, it just reduced the gain by 100. So if the gain in previous stage is large enough, the lock can still be acquired. And that's what was happening.
Ideally, we need to keep the AD602 on all the time and lock by scanning the offset with low gain. The loop will catch the lock (the exact same thing I thought was a problem) and once that has happened, we can just increase the loop gain the set value.
Presently, the gain behind the U5A AD602 is 101, which is kind of high. I just need to check if the above-mentioned locking method would work robustly without wrongly getting locked to any higher-order modes with the gain slider set to some threshold value within -10 dB to 30 dB. If that can't be done, I might have to reduce the gain in the first stage. For now, the cavities are locked and beatnote is traveling towards set point.
I brought a new AD602AR from 40m and replaced the U5A AD602 which from the previous post seemed like the culprit, but it wasn't.
I'll think of some new way of figuring out the point of the problem. It would be nice if someone can help me with this. All the history of the issue is on this thread starting at CTN:2451.
Edit Thu Oct 31 10:26:50 2019
Issue fixed. See CTN:2469.
I reduced the power falling on the PMC to ensure the high signal level isn't causing this problem. It was not. The problem still persisted.
Then, I did this reproducible step (quoted below)again, but this time I had a small 10 mV signal from SR785 going into FP2TEST and I was taking transfer function to TP2. If the U5A AD602 is switched off by the Blanking pin, the transfer function should remain null. This gave me a way of checking if the AD602 is wrongly getting switched on on its own.
This is good evidence in my opinion that the AD602 at U5A is faulty. I need comments on this conclusion. If I don't hear otherwise by tomorrow noon, I'll start working on replacing it.
Today, this problem happened again (check history for details). I have done the following investigative steps:
I'll take another inspection with a fresh mind next time. This problem needs to be resolved as we can't leave some unexplained phenomenon to keep happening in the lab.
The same thing happened again. This time, not just with the SPMC actuation voltage, but the South Laser slow voltage control was also unresponsive. However, I am not very sure about the latter. This was resolved once the restarted the whole lab. This narrows down the problem to following possibilities:
These still don't explain CTN:2456. Again, since this is an irreproducible error, I will just have to wait for it to happen to gather more clues. Right now, everything is fine and beatnote is traveling towards set frequency.
I ran the code for aLIGO coating structure and tried to reproduce fig5 and fig7 of Hong et al. paper. It turned out that the derivatives of the complex reflectivity were not matching with the paper. I rewrote the code, with a fresh mind without looking at previous code and voila, after increasing computation time slightly due to more brute force calculations, I was able to reproduce the figures. These figures are attached. Since I do not have access to the data of the figures in the paper (I tried to email the authors but got no replies), I could only try to plot it on the same scale and limits and check the values by eyes. The values seem to match. So I am more confident now to declare that this code completely follows the paper's calculations.
However, this does not change the coating Brownian noise. I have updated the noise budgets at the Daily and Latest plots.
Latest BN Spectrum: CTN_Latest_BN_Spec.pdf
Daily BN Spectrum: CTN_Daily_BN_Spec.pdf