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ID Date Author Type Category Subject
  2061   Thu Jan 25 19:05:38 2018 CraigDailyProgressFSSHigher Order Mode Power and Modulation Depth Estimate

NCAV total incident power P_0^\text{NCAV} = 1.102 \, \text{mW}

SCAV total incident power P_0^\text{SCAV} = 0.887 \, \text{mW}

NCAV carrier transmission power P_c^\text{NCAV} = 717 \, \mu\text{W}

SCAV carrier transmission power P_c^\text{SCAV} = 433 \, \mu\text{W}

NCAV estimated sideband power P_s^\text{NCAV} = 12.1 \, \mu\text{W}

SCAV estimated sideband power P_s^\text{SCAV} = 11.4 \, \mu\text{W}

NCAV estimated higher order mode power P_\text{HOM}^\text{NCAV} = 361 \, \mu\text{W}

SCAV estimated higher order mode power P_\text{HOM}^\text{SCAV} = 431 \, \mu\text{W}

NCAV modulation depth $\Gamma_\text{NCAV} = 0.257$

SCAV modulation depth $\Gamma_\text{SCAV} = 0.321$


 P_0 = P_c + 2 P_s + P_\text{HOM}, where P_0 is total laser power incident on the cavity, P_c is the power in the carrier, P_s is the power in the sidebands, and P_\text{HOM} is the power in the higher order modes, aka junk light.  I want to estimate P_\text{HOM} for our current mode matching to see how the PMCs will help us reduce PDH shot noise.


To get the above numbers, I measured the peak transmission power of the carrier resonance and the sideband resonance while scanning each cavity about its resonance frequency.  The trans power voltage is measured by each cavity's DCPD, which were realigned just prior to this measurement.  I then measured the carrier transmission power with the cavity locked using the Thorlabs power meter.

NCAV Peak Carrier DCPD Volts: 2.85 V
SCAV Peak Carrier DCPD Volts: = 4.16 V
NCAV Peak Sideband DCPD Volts: = 0.048 V
SCAV Peak Sideband DCPD Volts: = 0.110 V

NCAV power ratio  P_s /P_c = 0.0168
SCAV power ratio P_s /P_c = 0.0264

NCAV carrier transmission power P_c = 717 µW
SCAV carrier transmission power P_c = 433 µW

From the DCPD power ratios and the carrier trans, I estimate the power in the sidebands, assuming a lossless, critically-coupled cavity and all resonant power is transmitted and nothing else (i.e. no 'polluting' light like HOMs):

NCAV sideband transmission power P_s = 12.1 µW
SCAV sideband transmission power P_s = 11.4 µW

This gives power in the higher order modes, assuming P_\text{HOM} = P_0 - P_c - 2P_s :

NCAV higher order mode power P_\text{HOM} = 361 µW
SCAV higher order mode power P_\text{HOM} = 431 µW

Finally, I estimate the modulation depth \Gamma using the transmission power ratios by solving the equation below numerically in Mathematica (zipped attachment):

\dfrac{P_s}{P_c}=\dfrac{J_1(\Gamma)^2}{J_1(\Gamma)^2}


From the above results,

NCAV Percentage of light in HOMs: 32.8%
SCAV Percentage of light in HOMs: 48.6%

This is not great, we can surely do better with the mode matching to avoid unnecessarily scattering light into HOMs off the cavities.

Attachment 1: ModulationDepthCalc.nb.gz
Attachment 2: ModulationDepthCalc.pdf
ModulationDepthCalc.pdf ModulationDepthCalc.pdf
  2060   Thu Jan 25 15:34:29 2018 awadeDailyProgressFSSTesting and installing PMC electronics mk1

Some trouble shooting and unhacking of PMC servo board. 

Adding resistor pair to match 6V acromag binary channel values to PMC servo board

I added resistor pair to the acromag input as per option D in PSL:2058 for SW1 and SW2 binary inputs.  Values were 1.79 kΩ parallel to acromag 10 kΩ and 359 Ω for the series resistances.

For a 6 V supply I see 5.06 V in the on state and 0.696 V in the off state.  I will buy or find some low profile 6 V plug pack supplies to run the binary inputs directly. for the FSS and PMC boxes.

Hooking up the 'Blanking' Pin

There is a pin labeled 'Blanking' in the schematic.  I wasn't seeing any gain after the AD602, it turns out this is a on/off gain and needs to be drawn to ground to switch the amplifier on.  Two options are to manually resolder the pull up resistor to ground instead of +5V or add another channel.  I opted to add another channel in case it becomes useful in the future to kill the loop while still being able to see error signal after U2.

In the case of the blanking pin, the pull up resistor is 1 kΩ, not clear why this is the case, it may be needed to ensure the AD602 is held in the correct state: there could also be no reason. For the 1 kΩ pullup the input series resistance should be set to 33 Ω and resistance parallel to the acromag pull down should be set at 157 Ω.  This results in an on current of 40 mA through the 157 Ω and off current of 5 mA through the 1 kΩ.  The acromag XT1541 is capable of driving up to 250 mA through each binary channel (see page 12 of manual). However, there is on order of 0.23 W of power dissipated through the 154 Ω.  I don't have a 1 W resistor handy so I'll parallel a few 1/4Wers.

Also I attached the schematics below for future reference.  There are multiple versions of the board and are two board are of different vintage.  There are slight differences in the numbering conventions of components.  North path board is revision A-C, south will be revision D.

Unhacking a previous board hack 

With all the channels connected I did some initial testing of the north path board.  I injected a ~+7 dB signal at 21.5 MHz at stepped through a couple of phases of the PD input with a -13 dBm signal.  I found that the chain of signals was correct up to TP4.  After this point at U9 (the PA85 HV op amp) the signal was fixed at a 1/5th of the HV (I used single sided 50V).  I double checked to make sure none of the earlier stages were railed and that inputs into U9 were tided to ground (0V in the case of PMCRAMP) apart from the signal being varied.

After looking around the board and realizing that some resistors had been removed (R23, R25) I searched back through the elog and found that Frank had modified to board to uses it as a HV PZT amp (see PSL:529).  C25 is also missing. The nominal value on some versions of the schematic is 0.09 µF, that would place a pole at 12 Hz.  In later versions this is removed with the assumption that series resistance with the op amp output (R44 = 63.4 kΩ) and the capacitance of the PZT+cabling (estimated 227 nF) would produce a similar passive pole at about 10 Hz. Frank modified the board by adding a 1 kΩ in series with the HV output. We're going to need a 10 Hz pole for loop stability but I think I'll add a 10 Hz BNC connector Pomona box LPF in line with the PZT.  I don't have a characterization of the PZT capacitance or cabling.  Leaving it at 1 kΩ gives us some future flexibility. 

Also looking at the connections on the board it looks like for this D980352-A edition the rail splitting buffer of U13C is actually connected to R26. In later versions this is marked as diagnostic only. The value of summing input resistor R26 is actually 14.89 kΩ which explains the 0.2 x HV supply offset mentioned above (0.020 gain from HV rail through the rail splitter and gain of 10 at the summing junction at U9.  For now it makes more sense to center the HV amp at V_HV/2 so I changed R26 from 14.89 kΩ to 6.1 kΩ

I've included changes made in the third attachment. Its the The D980352-A version of the board but the schematic is from a later version of the board.  Marked values are correct.

Attachment 1: D980352-A.pdf
D980352-A.pdf
Attachment 2: D980352-D.pdf
D980352-D.pdf
Attachment 3: D980352-A_AWade20180128Mods.pdf
D980352-A_AWade20180128Mods.pdf
  2059   Wed Jan 24 23:05:28 2018 CraigDailyProgressFSSPDH Reflected Power Measurement

awade is thinking about adding in the pre-mode cleaners to reduce scatter and clean up the higher order modes entering our main cavities. 
This made me curious as to how much laser power currently resides in our higher order modes with our current mode matching.

Unlocked REFL Voltage, as measured by the RFPD:
NCAV 1.55 V
SCAV 1.12 V

Unlocked REFL Power, as measured by the Thorlabs IR Power meter:
NCAV 979 µW
SCAV 761 µW

Locked REFL Voltage
NCAV 1.00 V
SCAV 0.404 V

Locked REFL Power (inferred linearly)
NCAV 632 µW
SCAV 275 µW

Percentage of Laser Light Reflected while cavity is locked (i.e. % of light in sidebands or higher order modes):
NCAV 64.6%
SCAV 36.1%

I am not sure what our modulation depth is, so I am not sure how much light must be in our sidebands.  Evan measured the modulation depths five years ago to be around 0.2.  This should be redone.

Will be useful to have these numbers when we install/mode match with the PMCs installed, to see if we can do better.  Since we are not PDH shot noise limited yet, this is not directly affecting our noise measurements, but this will surely limit us from measuring coatings thermal noise at some point in the future.

  2058   Wed Jan 24 18:29:13 2018 awadeDailyProgressFSSFixing the Acromag latching issues and adding resistors to match the logic levels to FSS and PMC interface boards.

In troubleshooting binary engage interfacing between acromag XT1541 and the PSL servo board for PMC locking (see context PSL:2056) I think I've located the origin of our latching issues with the acromag cards: we're not using enough excitation voltage for the acromag's internal FET buffers to clear their on-voltage thresholds.

Acromag Binary Chanels REQUIRE 6-32 V Excitation

The XT1541 manual gives a recommended voltage range of 6-32 V for the digital excitation driving voltage.  I had been running it at 5.1V because it actually worked (most of the time) and it is a bad idea to drive a 5V level logic chip at an excess of 5 V. However, often, on a power cycle, the excitation voltage of the binary outputs isn't quite enough to push over the minimum threshold and the binary output of the cards latches off. I tried a few power cycles of the cards at binary excitation voltage at 4V, 5V, 6V etc and found that 5 V was just at the threshold where the binary outputs were responsive to modbus IOC channel changes.  6V> guarantees working.

Dealing with logic with pull up resistors

Most of the inputs for binary engage in LIGO electronics have some kind of pull up resistor on their inputs.  For the FSS it is 4.99 kΩ pull up to +5V.  In the pre-modecleaner servo boards (LIGO-D980352-D) there is a 10 kΩ pull up to +5V.  Aidan had previously come up with a solution for interfacing with the 4.99 kΩ pull up resistors in the FSS boxes input logic (see PSL:1573).  This was to add a ~810 Ω resistor in parallel with the acromag's internal 10 kΩ resistor to bring the off state voltage below the 0.8 V threshold of the HCT157D input chip (see option B in attached). This is satisfactory for ensuring that the off-state of the acromag forms a voltage divider back from the +5V rail to ground through the 810 Ω || 10 kΩ of the acromag + added resistor. It gives a value of 0.66 V at the HCT157D's input. However, as it stands, the on state is whatever the minimum of the acromag is (6V). That value exceeds the acceptable limit for the logic.

If we add one more resistor, it is possible to divide down whatever the acromag binary channels put out to 5V while also ensuring the off state is below 0.8 V.  The attached figure shows the four options for configuring a resistor network to adapt the acromags to driving the pulled-up inputs digital logic switches.

  • Option A connects the acromag directly to the digital logic.  In this case the 10 kΩ pull-down inside the acromag along with the 4.99 kΩ pull up to +5V gives an off state voltage of 3.34V (too high for 0.8V switching threshold) and an on-voltage limited to that of the source.
  • Option B  is what we use now for the FSS. This gives a useable off voltage (0.66 V for R1 = 820 Ω and Rpullup = 4.99 kΩ), but is limited to the on voltage of the source.
  • Option C is not great, you would need a really large value for R2 to bring the voltage divided between the input and +5V pullup rail close to 5 V, R1 would then need to be made very small, greatly increasing the current requirment.
  • Option D best.  In the case of the FSS interface, with a pull up voltage of +5V and Rpullup  = 4.99 kΩ, choice of R1 = 820 Ω and R2 = 164 Ω brings the on state to 5.0V (from a 6.0 excitation) and the off state falls to 0.66 V.

The current requirements for option D are also fine.  In the on state the acromag will need to source about 6.7 mA, which is fine. In the off state 0.8 mA will be sourced via the +5V pull up, which is also ok.

Here are the equations for choosing R1 and R2

R_1 = \frac{R_\textrm{pull up}}{(V_\textrm{pullup}/V_\textrm{off}-1)-R_\textrm{pull up}/R_\textrm{pull down}}

where Vpull up is the voltage of the pull up rail, Voff is the desired off state voltage, Rpull up is the pull up resistance of the input logic and Rpull down is the pull down resistance of the driving circuit. Using this value we can also find that the best series resistance is

R_2 = R_1(\frac{V_\textrm{excitation}}{V_\textrm{on}}-1)

where Vexcitation is the  high value of the driving circuit and Von is the desired on (high) voltage at the input logic.

 

Choice of pull down and series R for PMC boards

The pull ups on the PMC board logic are 10 kΩ. So to interface the acromag a good choice would be 1.79 kΩ parallel to acromag 10 kΩ and 359 Ω for the series resistances (in configuration D)

 

Small correction to above equations

Thu Jan 25 22:00:23 2018: I didn't quite include all the stuff in the above equations.  They will give good values for cases where R1<<Rpull up  and R2<<Rpull down. Here is the full equation in the case that the pull up resistor and/or pull up reistor values are comparible to the choices needed for R1 and R2​.

R_1 = \frac{R_\textrm{pull up}}{(V_\textrm{pull up}/V_\textrm{off}-1)-R_\textrm{pull up}/(R_\textrm{pull down}+R_2)}

where Vpull up is the voltage of the pull up rail, Voff is the desired off state voltage, Rpull up is the pull up resistance of the input logic and Rpull down is the pull down resistance of the driving circuit. Likewise

R_2 = \frac{R_1(V_\textrm{excitation}/V_\textrm{on}-1)}{1+(1-V_\textrm{pull up}/V_\textrm{on})R_1/R_\textrm{pull up}}

where Vexcitation is the  high value of the driving circuit and Von is the desired on (high) voltage at the input logic. As is very likely the case the pull up voltage rail is actually the same as the required on voltage and the above equation for R2 reduces to the value in the previous section.

Its a bit cicular here R1<--> R2. Just try some values, if logic pull up and source pull down are ≥5 kΩ then the equations in the second above are fine. 

 

 

Quote:

When Craig restarted the acromag IOC yesterday the North path FSS loop engage binary channel went into a permanent latch off mode.  This is a recurring problem that can be fixed by plugging the 5 V power in line to the acromag binary channels in with the FSS control boxes unplugged. Sometimes you need to plug and unplug a few times.  

It could be an issue with the way we have used 820 Ω resistors to bring the pull up 10 kΩ down to 758 Ω. It probably should be buffered somehow.  For now its good enough to get it working, once it's powered up its fine.

As an intermediate fix I soldered a 1000 µF electrolitic cap in line with the 5V supply to give it juice when first powered up.  This seems to make the latching go away most of the time (90%) when first powering up the units. So... slight improment.

 

Attachment 1: Acromag-to-pullup-logic_options.pdf
Acromag-to-pullup-logic_options.pdf
  2057   Tue Jan 23 23:33:03 2018 awadeDailyProgressTempCtrlMoved table environment temperature monitor

I moved the temperature monitor for the table air temperature away from 20 cm from the vacuum can to closer to the middle of the table around 4 pm today.

Should be more isolated from changes in the vacuum can heater.

  2056   Mon Jan 22 21:43:07 2018 awade, CraigDailyProgressElectronics EquipmentResurrecting mode cleaner electronics

We need to address some of the sources of scatter from the input side of the vaccum can.  Looking into the cavities through the input widow of the vacuum can we can see some ghost beams hitting about 2 cm to the side of the center of the input mirror.  Its a bit hard to take a photo and/or diagnose exactly where this is coming from.  One possibility is that it is coming from the prompt reflection of the cavity coming back from the window. 

If this is the case, the source of light would be junk higher order modes from the poor mode matching. We should expect the cavity to be impedance matched and transmitting most of the light. To test this out we need to resurrect the mode cleaners and their electronics. There is only so much we can do in adjusting positioning of lenses; a good deal of the mode mismatch is the astigmatism of the beams. 

Aidan has made up a 1U box to retrofit the old PMC Servo LIGO VME cards (D980352-A), pictured attachment #1 below.  These include a pair of acromag cards (XT1221 and XT1541) to provide binary channels to engage the loop, readout monitors and provide set points for gain and offset. I've finished wiring these up and checked the connections.  Tomorrow I will update the acromag IOC to add these new cards on IPs 10.0.1.46 and 10.0.1.47 for XT1221 and XT1541 respectvly. Channel allocations will be as follows:

Summary of channel allocations for North PMC box
Channel name Ch number:Card IP Function
C3:PSL-NCAV_PMC_SW1 DIO00:XT1541 10.0.1.47 Mixer engage (1)/Test1 engauge (0)
C3:PSL-NCAV_PMC_SW2 DIO01:XT1541 10.0.1.47 Engage for Test2 input injected at the HV amp stage.
C3:PSL-NCAV_PMC_MGAIN OUT00:XT1541 10.0.1.47 Broadband loop gain setting. 1/10 from ADC to AD602 chip
C3:PSL-NCAV_PMC_OFFSET OUT01:XT1541 10.0.1.47 Offset, applied at U2. 1/10 from ADC to output fo AD797
C3:PSL-NCAV_PMC_RAMP OUT02:XT1221 10.0.1.47 RAMP, injected at HV amp
C3:PSL-NCAV_PMC_LOMON IN1+:XT1221 10.0.1.46 Monitor of LO level
C3:PSL-NCAV_PMC_ERRMON IN2+:XT1221   PMC error monitor
C3:PSL-NCAV_PMC_PMCOUT IN3+:XT1221 10.0.1.46 Fast (actuator) output monitor

 

 

Attachment 1: 2018-01-30_21.29.02.jpg
2018-01-30_21.29.02.jpg
  2055   Mon Jan 22 20:30:11 2018 Craig, awadeDailyProgressTempCtrlDebugging vac can heater circuit issues: oscillations everywhere

awade and I have made some changes to the heater circuit Kevin and Kira made.  The heater circuit is now working as expected, and is currently heating the vaccan again.

1. awade decreased the resistive load of the vaccan heaters from 49.3 Ω to 12.2 Ω.  This caused some expected and unexpected issues.


Expected Issue 1: The lower resistive load gave less heating power per amp of current pulled.  We need to increase the amps our heater circuit could deliver to get. 

Fix: We increased the C3:PSL-HEATERN_OUT limit from 0.5 A to 1.5 A.  We also switched out the 1 amp power switch which shuts itself off when more than one amp flows through it.


Unexpected Issue 1: ~ 1 MHz, 100 mV voltage oscillation ringing in the heater circuit at the drain and source of the MOSFET when 0.2 to 1.5 amps would run through the circuit.  These oscillations would couple to all electronics near the inducting heaters, including the temperature sensors on the can and the table.

Fix: We placed a pair of capacitors, one 10 nF and one 10 pF, over the heater resistive load R_LOAD.  The resistive heaters can act as inductors at high frequency when large currents are run through them.  The capacitors pass the HF through without issue.


Unexpected Issue 2: Changing the power applied to the heater circuit caused our temperature sensors and FSS monitors to flucutate.

Fix: This one is not quite fully fixed.  We did a number of things to isolate the heater circuit from the rest of the electronics rack.  We added two more Sorenson power supplies exclusively for the heater circuit.  We removed the grounding wires from the temperature sensor pins on the acromag ADC.  We stopped the inductive oscillations coupling into the temperature sensors.

Now, the temperature sensors are working and seem to be completely isolated from the heater circuit, likely because there are no more inductive oscillations near them.  However, the FSS Fast out monitors are still coupled to whether the heater circuit is on, despite being on completely different power supplies (if the heater circuit is turned all the way on from 0 to 1.5 amps, the FSS Fast out monitor will move down by about 200 mV). 
We are still investigating this issue, its very important that our electronics are not coupled in a way we don't understand.

EDIT: It seems like our FSS Fast out monitors, and the rest of our electronics, are completely decoupled from the heater circuit's on or off state.

Attachment 1: CTN-Lab-Heater-Circuit.pdf
CTN-Lab-Heater-Circuit.pdf
  2054   Wed Jan 17 23:40:15 2018 awade, CraigDailyProgressTempCtrlDebugging vac can heater circuit issues: oscillations everywhere

More debugging cause of tank heater interfering with temperature sensors

An Isolated Power Supply For The Heater Driver

I extracted a pair of Sorenson rack mounted power supplies from the QIL (ATF) lab CDS rack.  These were then hooked up exclusively to the can heater driver configured for ±24 V.  We powered these up and the heater circuit drove the expected current through the resistive heaters on the tank.  However, the 2.5 MHz oscillations were still present in the circuit. Craig added some resistance between the op amp and the buffering mosfet (see next section), but this didn't quell the ringing. Changing the drive current lowered the frequency a little bit, seems to suggest it might be a saturation thing.

The reported values at the input of the acromag channels for the temperature sensors was low or negative with 1 A applied across the heaters.

In trouble shooting the possible cause we realized that the acromag temperature channels had been configured for floating input; the temperature sense box gives a signal referenced to a common ground.  The acromags have differential input ADC terminals, when operating a floating input it is best to ground the -in terminal of each channel. Grounding at both ends is bad practice so it was removed from the ADC side.  This didn't resolve the problem.

Operating off a separate power supply reproduced the same issues as before.

Adding Some Resistance On The Heater Op Amp Output

As mentioned in the previous post (PSL:2052) the MOSFET introduces a capacitive load to the op amp of order 800 pF.  Although the OPA140 is unity gain stable, this is not necessary the case when a capacitor is added.  This can add phase lag and instability with low closed loop gain. Yesterday Craig added a 1 kΩ resistor between the op amp and the IRF630.  In hindsight this was too much, 50 Ω is probably enough combined with the open loop output impedance of ~16 Ω from the op amp itself.

To address the ringing we should also add feed back capacitance as well to pre compensate the op amp.

Some reading on op amp instablity with capacitive loads: http://www.analog.com/en/analog-dialogue/articles/ask-the-applications-engineer-25.html

Floating supplies of the temperature sensor board

We stuck an oscilloscope probe into the temperature sensor PCB and found that the same ringing of the heater circuit (2.5 MHz) was making into the temperature sensor box. Unplugging all the sensors it went away.  Plugging them each in, one at a time, we found that the oscillations were dominate in sensors attached directly to the can. The circuit in the temperature sensing box is very simple, two power regulators and a LT1125 chip configured as a transimpedance amplifier gain of gain 30 kΩ.  It seems like it is picking up oscilations from the heaters somehow.

We changed power supplies to a ±15 V plug pack and found it was still there. Finally we took a SR560 and powered the temperature sensor board entirely off its batteries. We took care the only connection to the table/outside was a single sensor at a time connected to the tank.  The oscilloscope was hooked up with a high impedance (non-active) probe. Some senors had larger oscillations in the signal than others. Comparing the waveform to that coming out of the heater (on the same trigger + screen) it seems they are correlated: see attached screen shot. 

One theory is the heater circuit is driving some RF into the heaters, the temperature sensors and other electronics have some pick up. When we look at the DC levels of the temperature sensor channels, it is sitting at ~8.9 V on ±12 V supply rails to the LT1125. Excitations maybe driving the op amp beyond its linear range close to the rail giving a lower apparent average voltage.  

We may need to address these heater driver issues to fix what is going on elsewhere in the experiment.

 


Sorry, no proper screenshot of this scope trace, only one scope connects to the network.

Attachment 1: 2018-01-17_21.20.58.jpg
2018-01-17_21.20.58.jpg
  2053   Wed Jan 17 13:55:29 2018 awadeDailyProgressTempCtrlSoldering Kapton heaters

I revisited soldering of polyimide (Kapton heaters) late last week. 

Attempting a different type of solder

I purchased Pb97.5Ag1.5Sn1 solder recommended by Omega.  Here was the method used

  1. Stripped 28 AWG hookup wire
  2. Roughed up pads with sand paper
  3. Flattened and tinned wire ends with Pb97.5Ag1.5Sn1 solder (used iron temperature 630 F, below this it didn't flow well), I used quickchip low residue no-clean flux as the real purchased didn't have rosin core
  4. Added small amount of low residue no-clean flux to pad of Polyimide (Kapton) heater and soldered with a large tipped soldering iron at 630 F.  

This method created a slightly better contact, was strong to sheer applied force. However, with a vertical pull relative to the plane of the contact it snapped off brittlely. After a couple of tries the contacts wouldn't take at all. Put this down as a failure.

Maybe an acid flux or go the other way to Pb free solder. It shouldn't be this hard.

Shock welding the joint

The only strategy I found in the internet that actually seemed to be demonstrated to work was this shock welding technique: https://youtu.be/-XnbtGYRc54

I'm guessing they are just dumping a bunch of current through the joint with a little solder present.  

Commercially available Kapton heaters with leads already attached usually have a layer of glue over the end joints.  These wires aren't vacuum good, so no good for us. They also sell heaters with tinned solder pads.  I think this is some kind of pad welded/glued on the heater traces.

Clipping/crimping

Resistive traces can be peeled back a little from the Polyimide.  Clipping/crimping might be the only reliable way to contact.

 

Quote:

...Looking around online, some other people had the same problem.  I found a data sheet from Omega, Kapton heater KIT, that gives some instructions for soldering.  Its suggested to use a high
temperature solder (Pb97.5Ag1.5Sn1) with about 288°C (550°F) solder tip. We didn't have this in stock, so I've order a non-rosin core spool from McMaster. I'll try this out in the next few days.

  2052   Tue Jan 16 19:10:29 2018 awade, CraigDailyProgressTempCtrlDebugging vac can heater circuit: heat, oscillations, leaking current.

 

Time systematically address this.

How hot? Too hot.

Contrary to previous verbal reports the sense resistor and mosfet are very hot. 60C>= just by touch.  We either need to increase heater resistance or rethink circuit configuration to make the not be so close to the middle of its switching range.

Its also bad to run the sense resistor as such high current.  We are supposed to be reference our circuit voltage-to-current conversion to a reference impedance not frying the reference resistor.

Basic checks: DC supplies check, broken parts or grounding issues through thermal links

  • I stuck a multimeter into various test points in the circuit (reattached below). Regulators LM7815 and LM7915 regulators are receiving ±24 V on their input pins, ground pins are correct, they output expected ±15 V at their respective out pins. Op amp is receiving this dual supply so everything is good on the DC supplies.
  • 1 Ω sensing resistor is reading 1.2 Ω.  However, here the probes I was using measured 0.2 Ω. I suspect the extra 0.2 Ω is oxidation of the multimeter lead ends.
  • The IRF630 MOSFET is mounted to the case with a mica pad and apparently also thermal paste (picture below).  A metal screw was used, but it has an insulating plastic washer. I checked and its an open circuit from the TO-220 casing to the 1U box. So all good here.
  • Gave all wires a good tug.  Negative -24 V input came off.  Suspect.  
  • Negative 15V link from LM7915 to op amp pin 4 looks a little dry jointed to me, same with the link from LP filter into pin 3 (non-inverting op amp). They probably conduct, but need fixing anyway.
  • Also, looking at the underside of the board, the path to ground for the sense resistor is a convoluted set of linked traces
  • There was a 60 mV offset of the 1 Ω sense resistor ground side and the ground and the main link into the board.  This would suggest almost 17 Ω of impedance between these two points. Seems like a lot for copper traces

I resoldered the -24V input wire, touched up the solder on the above noted dry joints and added some extra links to rout grounding along a central axis of the board for a more direct route to the LP filter cap and sense resistor.  

Retesting board and checking voltages again 

Plugged everything back in and put a test voltage of 1.0 V into voltage input. 

  • Supply voltages look fine on the op amp and to the MOSFET
  • Two pole switch on power supply in has 1Ω across both junction, means that +24V supply rail is actually 23V at 1 A current draw.
  • Voltages across the sense resistor, heater, and MOSFET are 0.966 V, 12.40 V, 9.6 V.  This is the full total 23V we expect. 
  • Voltage from sense resistor 'ground' to the ground at the input of the box is now halved to 32 mV.  The above attempt to make a better ground plane only sort of worked.

The MOSFET is super hot because — as pointed out in a previous post (PSL:2048) — there is a large voltage * current drop across it. Its doing most of the heavy lifting in the half switched off state.

 

Looking with an oscilloscope

We set up an oscilloscope with a probe and poked around.  Found that there was a 2.55 MHz oscillation across the sense resistor.  This looks like about 2.5 MHz at the MOSFET gate as well. The Vpp of the oscillation at the sense resistor is 60 mV and 88 mV at the mosfet gate (op amp output pin).  Bad.

Input capacitance on the IRF630 is given to be 800 pF.   Maybe this is introducing some delay or lag in the response of the FET junction that is making the op amp loop unstable? Its likely to be more of a problem with low source impedance configurations. The solution is probably to add a capacitor somewhere. The internet suggests maybe a resistor between the MOSFET and the op amp. Some places on the web suggested a capacitor between gate and source: this seems wrong as it increases total capacitance across the gate-source junction. A better place might be directly before resistor to MOSFET and connecting to the non-inverting pin of the op amp. Combined with a resistor this can add a pole to compensate the feedback.

Current flow: table to rack

I put a multimeter between the table and the ground plane of the heater circuit.  I saw 80 mA current flow back from the table. This went away when I turned the heater current off. I turned the heater current off and it dropped down to 30 mA. Maybe there is a small nick in the heater pads on the tank. 

The residual flow back is suspicious too.  Little effort has been made to ensure we have robust grounding with a star/tree grounding pattern.  Something to put on the distant future to do list.

 

Re last post don't think this is a supply issue:

Quote:

The heated vaccan is no longer oscillating in temperature because of nonlinear spikes in current draw.  The vaccan heated to 30 celsius in about 5 hours.  The heater circuit HEATERN_OUT has settled at drawing 1 amp to heat the vaccan.  The INLOOP and OOL temperature sensors are systematically one degree away from one another.

It is notable that the "ENVMON" temperature sensor on the table, which is placed very close to the vaccan, reports cooling during this time of about half a degree C.  Not sure if this is a real effect, or due to the temperature sensors circuit interfering with one another.  The AC Thermostat sensor is also connected to the same circuit, and that seems normal.

The PLL beatnote is flat because the PLL locker lost lock overnight when the beatnote strength increased to the point of saturating the PLL electronics.  The beatnote strength increased because the RFPD bandwidth is 125 MHz, and heating the can brought the beatnote frequency from 230 MHz down to around 50 MHz.

 

Attachment 1: 20180115_HeaterCircuit.pdf
20180115_HeaterCircuit.pdf
Attachment 2: 2018-01-16_19.08.57.jpg
2018-01-16_19.08.57.jpg
  2051   Tue Jan 16 10:57:50 2018 CraigDailyProgressTempCtrlHigh Current Draw for Vaccan Temp Control Causing Nonlinear Voltage Spikes

The heated vaccan is no longer oscillating in temperature because of nonlinear spikes in current draw.  The vaccan heated to 30 celsius in about 5 hours.  The heater circuit HEATERN_OUT has settled at drawing 1 amp to heat the vaccan.  The INLOOP and OOL temperature sensors are systematically one degree away from one another.

It is notable that the "ENVMON" temperature sensor on the table, which is placed very close to the vaccan, reports cooling during this time of about half a degree C.  Not sure if this is a real effect, or due to the temperature sensors circuit interfering with one another.  The AC Thermostat sensor is also connected to the same circuit, and that seems normal.

The PLL beatnote is flat because the PLL locker lost lock overnight when the beatnote strength increased to the point of saturating the PLL electronics.  The beatnote strength increased because the RFPD bandwidth is 125 MHz, and heating the can brought the beatnote frequency from 230 MHz down to around 50 MHz.

Quote:

I replaced the OP27 inside the vaccan temperature controller circuit with an OPA140.  I then restarted the vaccan temperature control PID, with setpoint 30 celsius, to see if the problem persists.

After some extensive testing, the circuit appears to be working as expected.  The only exception is the affect the temperature control circuit has on all other electronics connected to the +24V kepco power supply.  The model number on our electronics rack is ATE 36-3M, and is a Size "B" Quarter Rack model, rated for "Approx 100 watts" of power, with max DC voltage of 36V and max current of 3 amps, according to Table 1.1 of the manual.  Our current readings on the power supply show between 1 and 2.5 amps at 24 volts, with the current depending on the 0 to 1.5 amps the vaccan heater draws.  So our max power output from the power supply is 60 watts, well within the power limits.

 

Attachment 1: 20180116_OvernightVaccanReheat.png
20180116_OvernightVaccanReheat.png
  2050   Mon Jan 15 21:35:14 2018 CraigDailyProgressTempCtrlHigh Current Draw for Vaccan Temp Control Causing Nonlinear Voltage Spikes

I replaced the OP27 inside the vaccan temperature controller circuit with an OPA140.  I then restarted the vaccan temperature control PID, with setpoint 30 celsius, to see if the problem persists.

After some extensive testing, the circuit appears to be working as expected.  The only exception is the affect the temperature control circuit has on all other electronics connected to the +24V kepco power supply.  The model number on our electronics rack is ATE 36-3M, and is a Size "B" Quarter Rack model, rated for "Approx 100 watts" of power, with max DC voltage of 36V and max current of 3 amps, according to Table 1.1 of the manual.  Our current readings on the power supply show between 1 and 2.5 amps at 24 volts, with the current depending on the 0 to 1.5 amps the vaccan heater draws.  So our max power output from the power supply is 60 watts, well within the power limits.

  2049   Mon Jan 15 14:43:53 2018 awadeDailyProgressFSSAdded cap to FSS acromag binary channel 5V supply

Edit Thu Apr 12 22:09:21 2018 (awade): WRONG, THE ISSSUE IS THAT ACROMAG NEED MIN 6V TO OPERATE BINARY INPUTS

When Craig restarted the acromag IOC yesterday the North path FSS loop engage binary channel went into a permanent latch off mode.  This is a recurring problem that can be fixed by plugging the 5 V power in line to the acromag binary channels in with the FSS control boxes unplugged. Sometimes you need to plug and unplug a few times.  

It could be an issue with the way we have used 820 Ω resistors to bring the pull up 10 kΩ down to 758 Ω. It probably should be buffered somehow.  For now its good enough to get it working, once it's powered up its fine.

As an intermediate fix I soldered a 1000 µF electrolitic cap in line with the 5V supply to give it juice when first powered up.  This seems to make the latching go away most of the time (90%) when first powering up the units. So... slight improment.

Attachment 1: 2018-01-14_20.24.02.jpg
2018-01-14_20.24.02.jpg
  2048   Sun Jan 14 20:40:26 2018 awadeDailyProgressTempCtrlHigh Current Draw for Vaccan Temp Control Causing Nonlinear Voltage Spikes

We checked the ±24 V lines with a multimeter with the heat at max current (1.5 A) and heater off.  There is no observed overdraw on current, the voltages at the end of the LIGO three pin 24 DC strip stays at their 24 V set points about ground.  So unlikely there is an overload on the power supply.

We also checked the signals out of the acromag, these were not being altered by settings of the can heater channel.

With the switch to lower impedance heater configuration the MOSFET is doing a lot more heavy lifting in dissipating voltage-current.   Can heater resistance is now 12.5 Ω.  So the voltage drop across heaters at 1.2 A is 15 V.  This means that from the 24 V rail the drain of the IRF630 MOSFET is sitting at 9 V from ground.  Total drop across the MOSFET = 9V - 1.2A * 1 Ω = 7.8 V. So the MOSFET is dissipating a voltage-current product of P = VI = 7.8 V*1.2 A = 9.36 W.  Maybe you want to check if it is heat sinking properly to the case. Open up the heater box and probe voltages at various points to make sure they make sense.

As for the pulling of values of other circuits: it is not clear what is going on.  Keep in mind that the transimpedance gain of the AD590 temperature sensor box is 30 kΩ. Its geared to convert a 1µA/K change in current into a 30 mV/K output.  If there is some grounding issue, this could easly interfering the convertion to volts. Maybe unplug the heater and see if there is any resistive path to the table, or back to the rack through either of the heater pins.  Have a close look at the heater circuit to make sure it wouldn't do anything like that either. The casing of the IRF630 is grounded to the source pin (I think) which can do funny things if its accidentally passing current through the casing: be sure to check that. Once again, open the box, stick some probes in there to make sure things make sense.

Also check Kira's adaptor box that connects to the Sub-D25.

The temperature sense boxes really should be generating a floating differnetial voltage output before transmitted accross the lab.  This can be done easly with a pair of op amps for each channel, but there is no space on the current board to do this. This would be a good project if we can find a keen undergraduate student: redo the transimpedace with low drift low noise FET input or FET input buffered op amp and add a single-sided to floating diff output then design a Eurocard side PCB with a few channels. Right now temp sense box is using a LT1128 which is low noise but not FET, I imagin its got more LF noise than it needs to. 

FYI: The reason the positive rail draws more current in general is that people habitually make electronics with extra small sub circuits that run on the positive rail. There is just more stuff hooked up to plus. 

I moved the last post to Category = temperature control

Quote:

awade has recently increased the power delivered to our vaccan to enable heating up to 40 celsius.  However, the high current draw from the +24 kepko power supply is causing nonlinear behavior in our control loop sensors, and affects the output voltage of everything powered by the +24V supply.  Most notably, the fast out signal on our FSS interfaces goes up and down along with the control signal on our vaccan temp loop, C3:PSL-HEATERN_OUT. 

HEATERN_OUT is how many volts is supplied to the MOSFET gate, which controls how many volts are over our 1Ω control resistor, which controls how much current runs through our load.  Kevin's elog provides the schematic

Plotted below is our beatnote frequency, out-of-loop temperature sensor, HEATERN_OUT, and our inloop temperature sensor.  Our in-loop and out-of-loop sensors are also powered by the +24V power supply, and when the power draw for controlling the vaccan temperature hit some critical level between 1.0 and 1.2 volts, our rtd sensors would spike in their reported temperature, causing the vaccan temp loop, and our cavity beatnote, to oscillate.

I turned off the temperature control loop, and turned off the voltage supplied to the vaccan so it will return to room temperature. 

The +24V power supply is still reporting ~ 1 amp of current draw, even with the vaccan heater off.  The -24V power supply is reporting 400 milliamps.  The maximum current on the front panels is 3 amps.  Perhaps we were pushing the +24V power supply too close to its current limit.


To do:
Open up the temperature control circuit and make sure everything is wired correctly.
Figure out why the +/-24V power supplies' current draw is so different, even with the vaccan temp loop off.

 

  2047   Sun Jan 14 16:52:53 2018 CraigDailyProgressTempCtrlHigh Current Draw for Vaccan Temp Control Causing Nonlinear Voltage Spikes

awade has recently increased the power delivered to our vaccan to enable heating up to 40 celsius.  However, the high current draw from the +24 kepko power supply is causing nonlinear behavior in our control loop sensors, and affects the output voltage of everything powered by the +24V supply.  Most notably, the fast out signal on our FSS interfaces goes up and down along with the control signal on our vaccan temp loop, C3:PSL-HEATERN_OUT. 

HEATERN_OUT is how many volts is supplied to the MOSFET gate, which controls how many volts are over our 1Ω control resistor, which controls how much current runs through our load.  Kevin's elog provides the schematic

Plotted below is our beatnote frequency, out-of-loop temperature sensor, HEATERN_OUT, and our inloop temperature sensor.  Our in-loop and out-of-loop sensors are also powered by the +24V power supply, and when the power draw for controlling the vaccan temperature hit some critical level between 1.0 and 1.2 volts, our rtd sensors would spike in their reported temperature, causing the vaccan temp loop, and our cavity beatnote, to oscillate.

I turned off the temperature control loop, and turned off the voltage supplied to the vaccan so it will return to room temperature. 

The +24V power supply is still reporting ~ 1 amp of current draw, even with the vaccan heater off.  The -24V power supply is reporting 400 milliamps.  The maximum current on the front panels is 3 amps.  Perhaps we were pushing the +24V power supply too close to its current limit.


To do:
Open up the temperature control circuit and make sure everything is wired correctly.
Figure out why the +/-24V power supplies' current draw is so different, even with the vaccan temp loop off.

Attachment 1: 20180114_CTNLab_TempControlSpikes_6hours.png
20180114_CTNLab_TempControlSpikes_6hours.png
  2046   Thu Jan 11 19:50:58 2018 CraigDailyProgressTempCtrlLab Temperature Control

I plotted all the available CTN lab temperature info in Celcius from January 3 to January 10, 2018. 

We have a vacuum can heater, in-loop, and out-of-loop sensor.  We have some temperature sensors on the table and near the AC units.  We have one working cavity shield heater on the north path, with no working sensors. 

We have some NPRO laser slow voltages for low-bandwidth high-range locking of the laser frequency to our Fabry-Perot cavities.  As of this writing we do not have a good direct estimate of the cavity temperatures from the slow voltages, but do have a good ∆T/∆V estimate.  This number is used to calibrate the slow volts into temperature, with some offset to put the temperatures in a reasonable range (~30 C for both cavities).

Some notable temperature events:

On January 7, I removed the insulating foam from the transmission window in order to mitigate scattering.  On January 9, I removed the insulating foam from the reflection window for the same reason.  Currently the vacuum can is not insulated.  According to the second plot, the vacuum can temperature controller was unable to apply enough heat to keep the can at 30 C.  awade has very recently increased the power capabilities of the vaccan heating circuit, we shall see if the vacuum can temperature is more stable.

 

Attachment 1: 20180111_200150_CTNLab_AllTemperatures.pdf
20180111_200150_CTNLab_AllTemperatures.pdf
Attachment 2: 20180111_200150_CTNLabTemperature_VaccanControl.pdf
20180111_200150_CTNLabTemperature_VaccanControl.pdf
Attachment 3: 20180111_200150_CTNLabTemperature_CavitySlowVolts.pdf
20180111_200150_CTNLabTemperature_CavitySlowVolts.pdf
Attachment 4: 20180111_200150_CTNLabTemperature_TableTemp.pdf
20180111_200150_CTNLabTemperature_TableTemp.pdf
Attachment 5: 20180111_200150_CTNLabTemperature_ACThermostatTemp.pdf
20180111_200150_CTNLabTemperature_ACThermostatTemp.pdf
  2045   Thu Jan 11 19:15:43 2018 awadeDailyProgressTempCtrlFollow up: Lowing can heater effective resistance

Rewiring and updating channels

I just re configured the heater driver to the configuration C illustrated in the last post. 

When I actually opened up the sub-D15 connector I found the previous configuration was not what I thought it was. The actual wiring was optimally bad, see attached schematic.  The resister network paralleled 38 Ω and 70 Ω heater blocks and then puts those pairs in series.  This meant uneven heating and higher resistance.  

I've rewired to put all heaters in parallel and updated the IOC db file entry for the heater voltage channel to max out at 1.5 V (1.5A at output). We should make a point of putting fuses in the heater driver circuits soon, to prevent a meltdown from mistakes in input voltage: this is now a going concern with the low heater impedance.  And/or implement a zenor diode crimp at the input.

Issue with high current affecting temperature readout

I ramped the drive current all the way up to 1.5 A, this is 27.7 W. This was max 9.7 W max before.  The out-of-loop sensor showed an immediate rise in temperature climbing at 0.16 K/min; the can can get from room temperature to 30 C in 1h20min which is much better than the 4-6 hours before.

However, I noticed when I cycled the current up and down this changed the reported temperature (in the out of loop sensor!). Which is very bad.  The transimpedance temperature sensor box that converts AD590 1µV/K sensor current to voltage runs off the same 24V lines as the heater. However, this voltage is regulated down to 12 V (pretty sure) and should be well bypassed on the board. So I'm not sure what the interaction is here. There should be a wide margin for the sensor board voltage regulators and the AD590 should also be pretty immune to any voltage dip (if there is one).

The temperature sensor board is reporting a full 0.6 degrees below when the heater driver is turned all the way up.  This needs to be debugged pronto in case all our temperature readouts are bogus.

Issue with the PID

Another issue I found was with the PID.  I don't think this is related to the above temperature dip with heater drive current changes.  The PID (settings P = -0.50, I  = 0.00020, D = 0) was actually reducing the current drive to 0.700 A when started from 1.5 A, even 5 K below the set point.  It should slowly wind up to the actuator limit and stay there until the set point is exceeded.  I have two theories:

  1. that the high value of P and the upper current limit have a ratcheting effect: upward movements in current hit an upper bound and the finite difference approximation used by the PID python locker subtracts the previous iteration's value.  This onesidedness effectively creates a negative integration term (in the wrong direction) pulling the process value down; and 
  2. there is an issue with the finite difference approximation of the PID loop itself, due to rounding errors.  We only really need a small integration term to push the temperature to set point.  The chosen value mainly determines how fast the can comes to temperature*. Setting it too high ends up with integrator wind up and oscillating overshoot.  I wonder if implementing a finite difference version of the PID introduces accumulated rounding errors when the P and I terms are very different magnitude.  A different implementation of the loop would keep computation of P, I and D separate until the last step of summing them all together. In that scheme one would only accumulate value on the integration term, avoiding many add subtract operations that may round the last digit off a much smaller integrator compared to other terms.

I turned down the P term and this seemed to solve the problem, but it shouldn't do this in the first place.  I've had a suspicion for a while that adjusting the P value was actually introducing an integration affect (usually in the opposite sign to the integrator term). However, it wasn't this extreme in previous cases.

This python PID script is a direct port of an earlier Perl script used at the 40m.  Its a very Perly way of solving the problem: using a single vector each for the process and error and performing shifting and pop on that vector. A python noob might write a script more like that given on Wikipedia

Changing the script isn't a lot of work, but working out what is wrong and why might be harder.

*it will also effect the bandwidth of the loop I guess.

 

Quote:

For the vacuum can heater, we are limited in the heater driver max power by the positive supply rail voltage and the maximum current permissible through the sense resistor. The 54 Ω of the can heater means that for 0.44 A of drive current, the drop across the heater is 24 V, the maximum voltage available to the circuit.  Thus there is a limit to total heating of 9.77 W, accounting for sense resistor and MOSFET voltage drop.

...For now I will just configure the heaters to all be in parallel (configuration C)...

Attachment 1: HeaterConfigs_badconfig20180111.pdf
HeaterConfigs_badconfig20180111.pdf
  2043   Thu Jan 11 14:24:11 2018 awadeDailyProgressTempCtrlLowing can heater effective resistance

For the vacuum can heater, we are limited in the heater driver max power by the positive supply rail voltage and the maximum current permissible through the sense resistor. The 54 Ω of the can heater means that for 0.44 A of drive current, the drop across the heater is 24 V, the maximum voltage available to the circuit.  Thus there is a limit to total heating of 9.77 W, accounting for sense resistor and MOSFET voltage drop.

The present resistor configuration is illustrated in the attachment (Configuration A).  Heating is proportional to area. With 38 Ω + 70 Ω in two parallel resistive circuits the current is the same through all resistive elements: this deliverers the most even heating per unit are across the can. This is at the penalty of lower heating.

Max power is given by 

P_\textrm{max} = \left(\frac{V_\textrm{cc}-V_\textrm{FET,GS}}{R_\textrm{load}+R_s}\right)^2 R_\textrm{load}

Where Vcc is max supply voltage, VFET,GS is the voltage drop from gate to source of the FET, Rs is the sensor resistor (1 Ω, 3W, 50ppm/K in this case) and Rload is the total resistance of the heaters on the can. I've plotted the maximum power verses load taking the larger of heater load voltage drop or max current as the limiting factor for a few choices of max current. This is attached below.

Of the three configurations of resistive heater hookup the option C provides the most heating within the voltage limits of the circuit but with 3.8 times more heating at the ends compared to the middle.  The time constant on heating is probably long enough for the steal to conduct without too much temperature gradient.  Configuration B is a compromise with relatively even heat with about 21 W heating.

Previous measurements of the heating requirements showed that 24.5 W was enough to hold the vacuum can at stead state 45 C.  Heating of 10 W is sufficient to reach 30 C easly. We would like the option to heat up to 40 C, so configuration B in the attached schematic is enough but doesn't leave much overhead.

For now I will just configure the heaters to all be in parallel (configuration C).

Self heating and drift

We want to avoid driving the sense resistor towards it maximum current, it will self heat and drift by

\Delta P_\textrm{Drift} = -2 \left(\delta R_s/R_s \right ) \times P_\textrm{heater}

to first order.  Where the fraction is the ppm/K expected change and P_\textrm{heater} is the average set point heating power. For the resistor used (1Ω) this temperature coefficient is 50ppm/K with a maximum power dissipation of 3 W.  From some previous tests running the the heater driver at 2 A I found the sense resistor seemed to heat up by about 3-5K (just by touch feel).  This could mean that a 20 W heater setting could self heat the circuit to a offset of 10 mW. We want to limit current to some reasonable value, choose 1.5 A for now.

Attachment 1: HeaterConfigs.pdf
HeaterConfigs.pdf
Attachment 2: MaxHeatingPower_CurrentAndVoltageLimited.pdf
MaxHeatingPower_CurrentAndVoltageLimited.pdf
  2042   Thu Jan 11 13:31:36 2018 awadeMiscLab InfrastructureAnts nesting in PSL lab

There are a large number of ants making a trail from the ATF lab to the PSL lab.  They seem to be heading into a hole next to the lab door.  I just saw a queen ant poke its head out of the hole.  

Ants in the ATF lab are taking their ussual route along the AC conduit.  There are Terro baits laid ever 3-4 meters and they have almost emptied every one. The trails continue to the mechanical plant room accross the hall (room 259ME)

Laying down Terro ant poison now. Will buy more.

Thu Jan 11 13:33:04 2018

Attachment 1: 20180111_antnestforming.jpg
20180111_antnestforming.jpg
  2041   Thu Jan 11 12:33:06 2018 awadeSummaryScheduleWeekly todo list W2:2018

Updated todo list is up on the labdata git ctn_labdata/issue/10.

Priority over the next week is mittigating ghost beams within the vacuum can. Its liklely we will need to vent to reangle cavities. 

  2040   Thu Jan 11 11:35:36 2018 CraigHowToComputersGetting data from our frame builder from anywhere

Steps to getting data from our framebuilder, since many of the steps are pretty hard to remember.  These steps were performed on a Macbook running OSX Sierra 10.12.5.  Dataviewer was opened locally using XQuartz.


1) SSH into PSL Lab computer ws3 through the public port 2022:

$ ssh -Y controls@131.215.115.216 -p 2022   (I often alias this on my PC .bashrc as some command: $ alias ws3="ssh -Y controls@131.215.115.216 -p 2022")

2) SSH into framebuilder4 fb4:

$ fb4 ("fb4" is already aliased on ws3 to be the command $ ssh -Y controls@10.0.1.156)

3) Launch dataviewer:

$ dvlaunch ("dvlaunch" is an alias to $ LIGONDSIP=localhost dataviewer.  This tells dataviewer to look locally for frames.)

4) Dataviewer will launch.  Click the "Signal" tab.  Click the "Slow" button.  Channel options "C3" and "C4" should appear.  The PSL Lab is "C3".  Choose what channels you want to plot.

5) Click the "Playback" tab.  Click "Second Trend" because other modes don't work.  Unclick "Min" and "Max".  Select X Axis Time as "GPS".  Choose your start and stop times you want plotted. Finally select your signals.  Click "Start".

Wait a while.  The terminal you ran dvlaunch from should give a progress report.  After all data is retreived a plots page should automatically appear with the channels and plot start/stop times you requested.


If you want to save the data from your plot:

6) Click on the plot you want to save data from.  It will have little black boxes in the corners of the plot when selected.

7) Click the "Data -> Export -> ASCII" tab.  A window called "Grace: Write sets" should open.

8) Click on the option under "Write set(s)", and change the Format from "%.8g" to "%.10g" to get all the digits of the GPS time.  Change Selection to whatever you want to name your datafile.  Click "OK". 

Your data should be saved.  Make sure it is formatted well.

  2039   Thu Jan 11 00:28:07 2018 awadeNotesTempCtrlSoldering Kapton heaters

This afternoon I revisited the soldering of Polyimide (Kapton) heaters.  All the tests to date have failed to form a nice joint between the exposed pads and hookup wire.  The standard Kester Pb63Sn37 solder usually balls up and refuses to take, any joint it does make is brittle and too unreliable for vacuum use.

I tested two theories:

  1. that the pads weren't taking because they were not up to temperature; and
  2. it is just a surface tension/chemistry problem and a different alloy with lower contact angle will take better.

Testing with more and uniform heating

I set up a metal plate with the portable vice grips, attached a heater sticker and heated it with the soldering heat gun to just under 200C (max limit of the Kapton). I then tried soldering with 287 C soldering tip taking care to really heat up the joint.  Standard Pb63Sn37 solder still didn't take, even with heat high enough to damag the Polyimide. No amount of heat is going to improve this.  I repeated again after rouging up the surface with an abrasive. This helped a little but the wire easily detached with a little tug. Too brittle.

Trying different solders

I found a few different types of solder in the PSL lab.  I also looked at the 40m, EE workshop and ATF lab draws, but they only had the standard Pb/Sn combinations.  I found that Sn60Cu? actually worked a bit better.  The flow was about the same, but the solder wetted much closer to the pad.  The joint was better than before but with a little more force (0.1-0.2 N) it also snapped off brittlely.  There was no solder residue left on the pad, it all stuck to the wire. No good.

Looking around online, some other people had the same problem.  I found a data sheet from Omega, Kapton heater KIT, that gives some instructions for soldering.  Its suggested to use a high
temperature solder (Pb97.5Ag1.5Sn1) with about 288°C (550°F) solder tip. We didn't have this in stock, so I've order a non-rosin core spool from McMaster. I'll try this out in the next few days.

 

Quote:

I looks like the only feasible way to attach wires to our Kapton heaters in vacuum is to solder them on. I can't find any suitable clips/crimps.  There are two issues. First, I and Yinzi both found it hard to form a reliable solder joint to the Kapton pads in our initial tests: the surface type and the fact it is thin and holds no heat means that the solder beads onto the wire and doesn't wet to the flat surface of the heater electrical contact.  Secondly, solder has rosin and other contaminants that are potentially bad for outgassing and redepositing on optics in vacuum. Furthermore, it is difficult to bake because it has such a low melting point.

  2038   Wed Jan 10 16:04:03 2018 awadeDailyProgressTempCtrlNorth Cavity Temperature Fluctuations from Slow Voltage Laser Control

This estimate seems much larger than the actual beat note drift would suggest.

The beat note frequency is drifting around by order 10 kHz at the moment.  That would suggest a relative temperature shift on order of 60 µK. I  more inclined to believe the BN inferred temperature shift than deriving the value from laser slow voltage temperature control.  With the laser slow voltage, this can be traded off for fast PZT offset, the PID is oscillating around a little,  you need to look at the long term average rather than the twiddle of the lowest digit on the medm screen.

Quote:

We have a calibration of the north slow voltage control into laser frequency of \dfrac{df}{dV_{\text{slow}}} = 3.481 \,\dfrac{\text{GHz}}{\text{V}_\text{slow}}.

From this number we can estimate how much the cavity temperature is fluctuating based on what laser frequencies lock our cavities:

\dfrac{\Delta L}{L} = -\dfrac{\Delta f}{f}, where L is the cavity length of 3.68 \,\text{cm} and f is the frequency of the resonating laser in the cavity.  

When the temperature of the cavity changes, we change the length of the cavity as well according to

\Delta L = \alpha \,L \,\Delta T, where \alpha is the coefficient of linear expansion of 5.5 \times 10^{-7} \,\text{K}^{-1}, and T is temperature.

Combining the above gives

\dfrac{df}{dT} = -\alpha \,f = -155 \,\dfrac{\text{MHz}}{\text{K}}

Combining with the calibration of the laser slow voltage:

\dfrac{df}{dV_{\text{slow}}} \bigg/ \dfrac{df}{dT} = -22.5 \dfrac{\text{K}}{\text{V}_\text{slow}}

We usually get fluctuations in \text{V}_\text{slow} on the order of 10^{-4} \,\text{V}, which would mean our cavity is fluctuating in temperature by ~2\, \text{mK}.

 

  2037   Wed Jan 10 11:31:03 2018 awadeSummaryLab InfrastructureLasers cycled on and off

I had a guy come and fix the lighting in the lab.  I tried to replace the tubes yesterday and found that the ballast had died for one row of lights.

A facilities guy came this morning and replaced the ballast on the outer row on the north side of the lab.  I turned the lasers off from 10 am till 11:30 am Wed Jan 10 to make  it easier for him.  The lasers are now back on and cavities are locked. I didn't relock the PLL. 

  2036   Mon Jan 8 14:46:09 2018 CraigDailyProgressscatterNorth Cavity Transmission Backreflection

Yesterday I optimized the North cavity visibility for a long time, and achieved around 60% visibility, which is pretty good for us.  This was all to realign the North cavity to eliminate ghost beams which are visibly interfering with our main beam.  After all of the optimization, however, the ghost beam is still visible when the cavity is unlocked, and fringing with the main beam when the cavity is locked.

I figured I would take out the post-cavity lens (PLCX-25.4-154.5-UV) to see if we could eliminate the ghost beam that way (Photo 1).  However, something very unusual happens when I take out the post-cavity lens.  North REFL starts oscillating in a square wave fashion (yellow oscilloscope trace in Photo 2).  These REFL oscillations also occurs if I put the lens back in, but adjust the alignment to be normal with the vaccan window.

This leads me to believe that there must be some direct backreflection from the transmission table into the north cavity, which affects the levels of reflected light on the North FSS RFPD.  This could explain why our scatterer is so robust, but it does not explain why we see the scatterer on transmission even when the cavity is unlocked.

I am now going to try to track down the source of the north cavity backscatterer.  We have looked for this type of scattering before, and no obvious candidates stood out.


Edit: The backscattering is coming from the NCAV Transmission DCPD.  Work is underway to eliminate this backscatterer.


Edit 2: The North cavity REFL oscillations were caused by the ISS still being on, not backscatter.  The post-cavity lens position only mattered because some positions allowed light on the North Trans DCPD and some did not, and if any light was on the DCPD, the ISS tried to stabilize it's intensity.  The ISS is now off.  The true North path scatterer is still present.  Work still focused on eliminating this ghost beam.

Attachment 1: CTNTransTable.jpg
CTNTransTable.jpg
Attachment 2: CTN_NCAV_REFL_Oscillations_without_postcavity_lens.jpg
CTN_NCAV_REFL_Oscillations_without_postcavity_lens.jpg
Attachment 3: CTNTransTablePart2.jpg
CTNTransTablePart2.jpg
  2035   Mon Jan 8 14:07:06 2018 awadeDailyProgressTempCtrlCavity shield heater circuit

Here are notes on the heater driver circuit for the cavity shield heaters and its schematic.  Its built in a standard busboard black box on one of their generic SMT protoboards.

The build is robust within the box, but heat sinking of MOSFETs and voltage regulators onto the casing and the various solder reworkings make the casing a little difficult to dissemble and reassemble.  

Circuitry consists of a Sallen-Key 2nd order LP filter set at 159 mHz corner, followed by an op amp buffer wrapped around a IRF630 MOSFET. I used OPA827 JFET input op amps (recommended in 40m:8125), these were the ones we actually had in stock at the time. They also have very low typical offset drift, quoted as 0.5 µV/K in the spec sheet. The schematic is attached below with an estimate of the voltage noise up to the FET buffer stage. I don't have a model of the IRF630's noise, but the largest issue is actually the very low frequency drift of the op amps and input voltage from the DAC.

So heater driver box gives two channels of 0.1 A/V conversion with some low pass filtering. Input is through two BNCs.  Output is into a Sub-D 9 connector with the following pinouts:

1-6: heater on cavity (south)

4-8: heater on cavity (north)

I checked the resistance of the south path heater.  It looks like its still hooked up but has a resistance of 85.6 Ω compared to the 156.8 Ω of the north cavity heater.

Power and current limits

The voltage-to-current conversion is made using a 10 Ω ± 1 %, 1 W, wire wound 20 ppm/K resistor (Vishay RS01A10R00FE70). The maximum current, within the 1 W power spec of the resistor is 320 mA: maximum voltage is therefore also 3.2 V. It is advisable to operate well below this to ensure the resistor is not self heating too much. I considered adding zenors to crimp the input voltage, but wasn't sure if they cause unexpected noise/behavior at the time. It would be advisable to put some kind of voltage crimp on future reworks to prevent overdriving the circuit with user error.

The logic of the FET buffer is that the op amp will adjust the voltage of the gate until the source pin matches the non-inverting input of the second op amp.  Current is sourced through the heater load (~170 Ω loading) sufficient to generate voltage across the 10 Ω sensing resistor.  The exact resistance of the heater load is unimportant to the voltage to current conversion (0.1 A/V), except where resistance is so high that the source voltage overhead is not large enough to drive the set point current into the drain.

The FET is essentially adjusting the voltage divider at the output formed between the heater load and 10 Ω sense resistor.  The set point voltage at which the heater load is saturated, i.e. where driving voltage isn't high enough to provide enough current through a resistive heater, is given by

V_\textrm{set} = \frac{V_{cc}-V_{FET,GS}}{R_\textrm{load}/R_s+1}

where V_set is the max set point voltage at the non-inverting input of the op amp. Vcc is the supply voltage, V_FET,GS is the gate to source potential drop, Rload is the heater load attached to the drain pin, and, Rs is the sensing resistor.

The north cavity shield heater impedance is 156 Ω. Thus maximum drive voltage is 0.867V, this means a drive current of 87 mA.  The maximum heating is therefore P = I^2R = 1.17 W. Initially we expected cavity heating requirements to be 0.8 W, so it should be sufficient, although it might have been a good idea to have a little more overhead. Presently the heater driver is st to 0.7692 W for a 66 MHz offset of the north from the south cavity.  

Drift 

The Sallen-Key low pass is only to cut off any higher frequency noise from the DAC.   Very low frequency drift in the driver circuit is another issue. From Craig's previous estimates from CTE (PSL:2027) the temperatures to cavity frequency conversion is 155 MHz/K.  The OPA827 has a typical offset voltage 75 µV with a drift of 0.5 µV/K (max 2.0 µV/K).  To first order set point voltage drift to heating power fluctuation ( due to op amps) is

\Delta P_\textrm{drift} = 2 \delta V_s V_s R_\textrm{load}/R_s^2 \approx 2.2 \delta V_s\Delta P_\textrm{drift} = 2 \delta V_s V_s R_\textrm{load}/R_s^2 \approx 2.2 \delta V_s [W/V]

Where delta V_s is the drift of the set point voltage and R_load is the heater impedance (156 Ω).  For worst case 4 µV/K drift (in op amps) we should see 8.7 µW/K change over a heating power of about 700 mW, or 12 ppm/K. 

The sensing resistor may also have small temperature drift.  Its rated to 10 Ω ± 1% with a 20 ppm/K drift. Drift in power output to first order is

\Delta P_\textrm{drift} = -2(\delta R_s/R_s)\left(Vs/R_s\right)^2V_\textrm{load} = -2(\delta R_s/R_s) \times P_\textrm{heater}

So a drift of 20 ppm/K means a heater power output change of -40 ppm/K in power or 28 µW/K over heating power of about 700 mW.

It is likely that the dominant source of very-LF drift in the heater will probably come from the input voltage from the DAC.  The Acromag XT1541 8 Ch output Ethernet module has a spec of ± 50 ppm/K output ambient temperature drift. This would make its output heating drift of order ± 110 µW/K out of 700 mW or 157 ppm/K. 

Estimating north shield temperature

At this stage we are not sure what the true temperature of the north cavity is.  The plan is to step the north heater power down by 0.2 W with the FSS locked and track the shift in the slow laser voltage overnight.  From there we can estimate the thermal settling time and temperature. The facts that we are missing is the true effective emissivity. See known heater shield properties in PSL:1737). First order heat flux as a function of temperature above vacuum can is* 

\Delta Q = \varepsilon_\texrm{Cu} \sigma A T_\textrm{can}^3\delta T

Where \varepsilon_\textrm{Cu} is emissivity of copper 0.03 to 0.1 depending on degree of polish and oxidation. \sigma is Stefan–Boltzmann constant 5.67e-8 [W/m^2/K^4] and A is the area exposed to radiative transfer. Estimating emissivity of 0.05 and vacuum can temperature of 303 K this would give 6.5e-3 W/K.  Doesn't really make sense. We need a measurement, or a sensor.

* This is ignoring cylindrical geometry.

Implementing a PID

Nope. I didn't do this.  I tried for a while to tune this over the Christmas break, but the time constants are very long (30 mins to 60 min depending on the heater step). After turning it off the PID I found that the once the heater settled, an acromag set point voltage of 0.69774 V (0.7692 W) gave a beat note of ~66 MHz drifting around in a range of 5-10 kHz over the course of 30 mins.  This is a task for the todo list: look around for an auto tuner or make a model of the plant to speed up the tuning.

Attachment 1: Basic-two-channel-heater-driver.pdf
Basic-two-channel-heater-driver.pdf
Attachment 2: LISO_model_PSLlabLNHeaterUptoIRF630Mosfet.png
LISO_model_PSLlabLNHeaterUptoIRF630Mosfet.png
Attachment 3: DSCF3508.JPG
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Attachment 4: DSCF3509.JPG
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Attachment 5: DSCF3510.JPG
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Attachment 6: DSCF3506.JPG
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Attachment 7: DSCF3507.JPG
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  2033   Sun Jan 7 17:53:26 2018 Craig, awadeDailyProgressscatterScattering Sources Probably From Inside Vaccan

TL;DR We have (1) poor alignment of light into the cavities and (2) ghost beams interfering with our resonant light from inside the vaccanThis could be the cause of our scattering shelf.


We have been messing around on the transmission table, trying to determine where are 50 Hz scattering shelf is originating from. We have a pretty good candidate for the scatter source in the north path fringer, with what appears to be a ghost beam coming out of the north cavity window.  The photo below was taken when the cavities were unlocked and with half wave plates making the back table light p-polarized, generating these beams ghost beams which are always there and interfering with the main light.

We took off the vaccan's foam on the back, and looked in with the IR viewer.  I can count at least five spots of IR light, some bright and focused, some diffuse and clipped. 

I'm going to play around with cavity alignment and try to eliminate the ghost beams.

Attachment 1: TransmissionMonitors_UnlockedCavities_PPolarizedLight.jpg
TransmissionMonitors_UnlockedCavities_PPolarizedLight.jpg
  2032   Fri Jan 5 20:49:20 2018 awadeMiscOtherPower pins broken off on spare (original PSL lab) NF1811 detector

I borrowed the spare (original) New Focus 1811 (DC-125 MHz) photo detector from the PSL lab to use in the WOPO experiment.  It is broken: spoiler alert the pins on the power in have been bent and some look snapped off, see pictures.

On plugging it into a power supply and looking at the DC coupled output, I found that it was on the negative rail with a high impedance load and when 50 Ω loaded had a 25 Hz oscillation.  This confirms a previous vague suspicion that something was up. We had previously switched this particular unit out for one borrowed from the 40m.  It looks like it may have not been measuring a beat note after all, although the issue preventing us from first seeing a BN was also a bad set of temperature tunings for the lasers at the time.

I thought maybe some internal circuitry was busted, but when looking closer at the pins in the power-in plug it looks like they have been damaged. If its missing ground, most LM79/79 regulators will pass full voltage, otherwise only a single supply will also prevent proper operation.

At some stage I'll open it up, work out the feed through dimensions and replace it with a more robust connector. Or an identical one if Digikey has it.  It might also be a good time to inspect the RF transimpedance stage and see what our options are for lowering the 7kΩ first stage amplification.  It looks like this will need to be a chip replacement, I haven't yet found a 14 pin SOIC replacement for the NE5210D

Attachment 1: IMG_1810.JPG
IMG_1810.JPG
Attachment 2: IMG_1818.JPG
IMG_1818.JPG
Attachment 3: IMG_1811.JPG
IMG_1811.JPG
Attachment 4: IMG_1812.JPG
IMG_1812.JPG
  2031   Fri Jan 5 12:29:55 2018 CraigDailyProgressPLLPLL Mixer Circuit Board Made and Attached to Rack #EOM
Attachment 1: PLLMixerCircuitboard.jpg
PLLMixerCircuitboard.jpg
  2030   Tue Jan 2 15:39:01 2018 awadeDailyProgressOtherN2 bottle is turned off

Table refloated at Tue Jan 2 15:38:59 2018.  PLL locker script was still running but had dropped lock; PLL now been relocked but seems to have excess noise. FSS North and South were both still locked.

 

Quote:

Sun Dec 24 03:23:25 2017 Turned N2 bottle off for christmas break. Refloat and realign after christmas.  Laser have been left and cavity + PLL is still locked if we want to test some PID locking on the beat note and heater.

 

  2029   Sun Dec 24 03:23:10 2017 awadeDailyProgressOtherN2 bottle is turned off

Sun Dec 24 03:23:25 2017 Turned N2 bottle off for christmas break. Refloat and realign after christmas.  Laser have been left and cavity + PLL is still locked if we want to test some PID locking on the beat note and heater.

  2028   Fri Dec 22 14:38:24 2017 awadeDailyProgressscatterFixed bad PDH error signal

It turns out that the bad PDH signal caused by monitor daughter board (D040424) note being properly seated in its plug.  It wasn't screwed in and I must have just bumped it out when putting in some test leads.

I have screwed it in properly now.

Quote:

As pictured in PSL:1994, there was a burn mark in the center of the final lens before the north reference cavity.  I've switch out this PLCX-25.4-77.3-UV-1064 for a new lens. Cavity is realigned with a Vmin,Vmax = 0.492V,1.56V, giving a visibility of 52% (about as good as it will get with this MM solution). Polarization is well aligned with the cavity basis.

Something is up with the PDH error signal (see attachment below), last time we checked this it was the textbook shape. Nothing has changed electronically, I need to double check if things are ok further up the path going through EOMs.  North has been locked nicely for about 3 weeks now with pretty much 100 % uptime.  Power into the cavity is 1.08 mW, so not a saturating thing. It could be AM modulation. I'll check systematically the EOM polarization alignment and that of the AEOM tomorrow. If that fails I'll look at the electronics.

I also blew some dust of optics around the periscope with the ion gun, this may need to be repeated a few times as there is a lot of dust on the table and will take a while to purge out of the tent after each blast.

 

  2027   Fri Dec 22 10:54:24 2017 CraigDailyProgressTempCtrlNorth Cavity Temperature Fluctuations from Slow Voltage Laser Control

We have a calibration of the north slow voltage control into laser frequency of \dfrac{df}{dV_{\text{slow}}} = 3.481 \,\dfrac{\text{GHz}}{\text{V}_\text{slow}}.

From this number we can estimate how much the cavity temperature is fluctuating based on what laser frequencies lock our cavities:

\dfrac{\Delta L}{L} = -\dfrac{\Delta f}{f}, where L is the cavity length of 3.68 \,\text{cm} and f is the frequency of the resonating laser in the cavity.  

When the temperature of the cavity changes, we change the length of the cavity as well according to

\Delta L = \alpha \,L \,\Delta T, where \alpha is the coefficient of linear expansion of 5.5 \times 10^{-7} \,\text{K}^{-1}, and T is temperature.

Combining the above gives

\dfrac{df}{dT} = -\alpha \,f = -155 \,\dfrac{\text{MHz}}{\text{K}}

Combining with the calibration of the laser slow voltage:

\dfrac{df}{dV_{\text{slow}}} \bigg/ \dfrac{df}{dT} = -22.5 \dfrac{\text{K}}{\text{V}_\text{slow}}

We usually get fluctuations in \text{V}_\text{slow} on the order of 10^{-4} \,\text{V}, which would mean our cavity is fluctuating in temperature by ~2\, \text{mK}.

  2026   Thu Dec 21 18:57:12 2017 awadeDailyProgressscatterSwitch out final MM lens in North path

As pictured in PSL:1994, there was a burn mark in the center of the final lens before the north reference cavity.  I've switch out this PLCX-25.4-77.3-UV-1064 for a new lens. Cavity is realigned with a Vmin,Vmax = 0.492V,1.56V, giving a visibility of 52% (about as good as it will get with this MM solution). Polarization is well aligned with the cavity basis.

Something is up with the PDH error signal (see attachment below), last time we checked this it was the textbook shape. Nothing has changed electronically, I need to double check if things are ok further up the path going through EOMs.  North has been locked nicely for about 3 weeks now with pretty much 100 % uptime.  Power into the cavity is 1.08 mW, so not a saturating thing. It could be AM modulation. I'll check systematically the EOM polarization alignment and that of the AEOM tomorrow. If that fails I'll look at the electronics.

I also blew some dust of optics around the periscope with the ion gun, this may need to be repeated a few times as there is a lot of dust on the table and will take a while to purge out of the tent after each blast.

Attachment 1: IMG_1464.JPG
IMG_1464.JPG
Attachment 2: 20171121_ReplacedFinalNorthMMLens.jpg
20171121_ReplacedFinalNorthMMLens.jpg
  2025   Thu Dec 21 16:59:21 2017 awadeDailyProgressBEATTune up: FSS settings checkup

The values on the FSS sliders have been tweaked and changed over the last few weeks.  This wasn't done systematically so I've done a tune up so we have a checkpoint of some good parameters with known loop properties.

North

North FSS + PID settings
Setting Value
Common gain 2.29899 V
Fast gain 1.71654 V
Offset -0.33850 V
Slow Freq offset 3.4597 V
Slow P 0.00217
Slow I 0.00026
Slow D 0.00158

This gave a common loop UGF of 177 kHz (-145 degree phase) and a cross over frequency of about .  Hear I've used the direct Out1/Out2 method for speed and convenience. 

South

South was at Common UGF of 80 kHz and a crossover of 6.2 kHz. New values are below.

South FSS + PID settings
Setting Value
Common gain 2.94295 V
Fast gain 2.26449 V
Offset -0.2095 V
Slow Freq offset -6.1263 V
Slow P 0.00217
Slow I 0.00026
Slow D 0.00158

This gave a common loop UGF of 128 kHz (-146 degrees) and a crossover frequency of 10.5 kHz (-104 degrees). Again, I've used the direct out1/out2 method.

 

Edit awade Fri Dec 22 16:11:49 2017: I've revised North common gain down to 1.93899 V as there appears to be some gain peaking around the higher PZT resonances (~200 kHz and 400 kHz), this gives a UGF of 162 kHz (phase -145 degrees). Offset adjusted to -0.31630 V.

Edit awade Fri Dec 22 16:20:14 2017: Had to back off the south common gain as well for the same reason.  Now set to 2.38295V, UGF of 115 kHz (-145 degrees).  Offset adjusted to -0.2250 V

  2024   Thu Dec 21 14:38:17 2017 awadeDailyProgressOtherFloating the table on N2

We recieved a cylinder of gas. The compressor was at the lower end of presure so I decided switch in the new N2 bottle. The used bottle has been moved to the bottle cage in the ATF (QUIL) lab until facilities comes to pick it up.

Initial bottle preasure is 2300 psi.

I've applied 35 psi to the legs.  A presure of 25 psi is about enought to just float the legs, so 35 psi is ample.

Table is back and floating again.

  2023   Thu Dec 21 08:58:39 2017 CraigDailyProgressscatterScatter Shoulder Fit

I've been learning about scattering from here and here.  It seems most scattering equations are left in an arbitrary form of x_{\text{scatter}}(t), and simply measure seismic noise then use it as x_{\text{scatter}}(t) in the following FFT:

S_f(\omega) = \int_{-\infty}^{\infty} \sin\left(\dfrac{4 \pi}{\lambda} x_{\text{scatter}}(t)\right) e^{i \, \omega \, t} dt

If the sine wave were perfectly sinusoidal, say x_{\text{scatter}}(t) = x_0\,\Omega \, t, our FFT would yield delta functions at \omega = \pm \Omega.  However, scattering is rarely so clean.

If x_{\text{scatter}}(t) \ll \lambda where lambda is the laser wavelength, then our sine wave is approximately linear, and we get a clean spectrum at frequencies above the seismic noise.

When x_{\text{scatter}}(t) \approx \lambda , we get "upconversion" of scatter noise, i.e. the higher order modes of the sine wave start to matter, and this extends the scatter shelf into higher frequencies.


I fit a scattering shelf of the functional form S_{\text{Hz}}(f) = A \, e^{-\pi \,\Gamma \,f}, where A is a scatter coupling coefficient in units of hertz, and \Gamma is a half width half maximum (HWHM) of an underlying Lorentzian L(t) = \dfrac{1}{\pi} \, \dfrac{\Gamma}{\left(t - t_0\right)^2+ \Gamma^2}.

I found A = 34.0 \, \text{Hz} and \Gamma = 0.04 \, \text{s}.


We can think of the HWHM as a function of overall scattering displacement and velocity: \Gamma \approx \dfrac{x_{\text{scatter}}}{v_{\text{scatter}}}

If x_{\text{scatter}}(t) \approx \lambda, this gives v_{\text{scatter}} \approx 27 \, \dfrac{\mu \text{m}}{\text{s}}

Attachment 1: PLLNoisebudget_20171221_111006.pdf
PLLNoisebudget_20171221_111006.pdf
  2022   Tue Dec 19 22:44:48 2017 awadeDailyProgressscatterBuzz test of beat board

I used the buzzer to help locate sources of scatter in the transmission beat board. 

Here I used the SR785 in swept sine mode: 10kHz to 20 Hz, 10 mV source, a 35 mV offset, amplified by a thorlabs HV amplifier (x15). I took the mechanical transfer function from the buzzer to the PLL actuation signal (the same used to measure the displacement spectrum).

Below I've noted the frequencies and the amplitude in units of dB (ratio of PLL signal to excitation 10 mV). The frequencies listed in the upper half of the labels are buzz applied horizontally on the mount, those on the lower half are resonances excited with vibrations coupled in the vertical direction.  I would have made this labeling clearer, but I saved accidentally closed as jpeg so its been flattened.

Vertically applied excitations seem to couple down into the board more easily and common frequencies may well be same the same scatter source being excite from different points in the board. There are some standard resonances from vertical excitations at 1.03 kHz, 1.13-1.15 kHz, 3.11-3.2 kHz. These can also be seen by exciting the elevated breadboard directly. Maybe we can apply some sorbothane dampeners to the underside of the board to kill these resonances. 

Horizontal resonances were less likely to be common between mounts.  

Note that since I took this picture the ND filters were removed from directly before the PD.

 

Attachment 1: 20171119_BeatBoard_BuzzMap.JPG
20171119_BeatBoard_BuzzMap.JPG
  2021   Tue Dec 19 14:30:05 2017 awadeDailyProgressBEATPID control of north cavity heater, up for 24 hours

I've implemented the PID control of the north cavity heater through the heater driver box.  Its been running for almost 24 hours continuously now. It takes the computed beat note frequency from the PLL carrier and control frequency to the Marconi and feeds back to the heater power IOC channel.

Physical parameters are:

North cavity heater impedance = 158 Ω

Gain heater box = 0.1 A/V with 159 mHz LPF

Mean heater level  = 0.75 W

Thermal response = TBC with a model and a step function response.

---

I've gently tuning the PID parameters to improve settling time and reduce oscillations in the beat note frequency. It seems to have a response on the order of 4-8 minutes.  Current settings are too slow in reaching a set point from, say, a 10 MHz change; but at least it doesn't push the PLL so hard it drops. Settings are:

P = -1.0000; I  = -0.00006; D = -0.00050

Setpoint = 50 MHz; Average heating power = 0.75 W

Wander about set point is less than 10.0 kHz at the moment -- equivalent to 10kHz/(155MHz/K) = 64 µK -- but we would like much less for the final measurement of Brownian noise with small VCO slope.

There is nothing particularly systematic about the way I came to these numbers. I started with a little bit of I and then added some P once it got close to the set point.  I added some D to help damp the overshoot, but its not clear if this objectively improved things much. Everything moves on such a slow time scale that its hard to see the whole picture of the loop dynamics without a systematic approach. We want something that reaches the set point much faster and with more dampening but it will do for now.  I'm just noting the settings here so there is something to go back to. 

Quote:

I've changed the NCav heater slider to units of Watts. Voltage applied to the heater driver is computed using a soft IOC channel. This is so we have an actuator slider that is linear injected heat power rather than quadratic. 

However, when I rebooted the acromag1 IOC the channel didn't come up.  Craig had a look at the db file and found that I was using a calcout record for the intermediate power to volts conversion. To the error was that the power and voltage were both being written to the acromag which and output of zero volts regardless of the slider value.  Heater was down for about 60 minutes and has taken a long time get it back to <100MHz range of the south cavity. 

 

  2020   Tue Dec 19 11:28:47 2017 awadeDailyProgressBEATTune up: Minor PLL and FSS tuning, lowering beat note amplitude

I have Brittany's notes. It looks like their main modifications were to the biasing of the PD (I think), and adjusting the AC coupling network -- including the inductor -- before the NE5210 7kΩ all-in-one transimpedance chip.  I'll put these notes into the PSL electronics section of the ATF wiki.

It doesn't seem like we have have access to transimpedance gain as a knob to turn without compleatly removing the first stage of the AC path and making some PCB 'art'. Maybe there is an equivalent chip with some gain selector pins out there...

Quote:

Brittany & Jon have some holometer 1811 notes that would be helpful. I think they lowered the transimpedance there for this reason.

 

  2019   Tue Dec 19 10:26:47 2017 ranaDailyProgressBEATTune up: Minor PLL and FSS tuning, lowering beat note amplitude

Brittany & Jon have some holometer 1811 notes that would be helpful. I think they lowered the transimpedance there for this reason.

  2018   Mon Dec 18 19:00:47 2017 awadeDailyProgressBEATTune up: Minor PLL and FSS tuning, lowering beat note amplitude

Yeah, it would be much better to lower the detector gain. As things are I think the dark noise we measured today on the NF1811 was on the order of 200-300 nV/sqrtHz. This was measured from low passed output of the level 13 mixer used in our PLL.  We might have been a factor of 2 off there from 50 Ω into SR785.  With the factor of two this would put us at 20-30 mHz/sqrtHz in our noise budget for the current preamp-Gain = 20, Marconi=10kHz/V configuration. That will not let us clear Brownian noise of 10 mHz/sqrtHz @ 100 Hz.

It looks like these 1811 have a fixed 7kΩ all-in-one transimpedance amplifier chip strait after the ac coupling capacitor.  So no option to modify that, unless there is a pin-for-pin replacement on the market.  Do we have any non-resonant ~100 MHz photo-detectors or designs that are good? We don't need it right, scatter hunting is the task at hand, but would be good to have something lined up to switch in, if we have it.

Quote:

best to detect all the optical power and just have a lower transimpedance gain somehow

using any ND filters or power attenuators is against the rules of low noise measurement

 

  2017   Mon Dec 18 15:45:26 2017 ranaDailyProgressBEATTune up: Minor PLL and FSS tuning, lowering beat note amplitude

best to detect all the optical power and just have a lower transimpedance gain somehow

using any ND filters or power attenuators is against the rules of low noise measurement

  2015   Mon Dec 18 13:49:14 2017 awadeDailyProgressBEATTune up: Minor PLL and FSS tuning, lowering beat note amplitude

I've changed the NCav heater slider to units of Watts. Voltage applied to the heater driver is computed using a soft IOC channel. This is so we have an actuator slider that is linear injected heat power rather than quadratic. 

However, when I rebooted the acromag1 IOC the channel didn't come up.  Craig had a look at the db file and found that I was using a calcout record for the intermediate power to volts conversion. To the error was that the power and voltage were both being written to the acromag which and output of zero volts regardless of the slider value.  Heater was down for about 60 minutes and has taken a long time get it back to <100MHz range of the south cavity. 

  2014   Mon Dec 18 13:42:21 2017 awadeHowToComputersHow to: update and restart the fb4 framebuilder

fb4 is now logging channels in the c3 (PSL) lab.  To update channels you need to:

  1. ssh into 10.0.1.156 with username controls (ssh -Y controls@10.0.1.156);
  2. cd to dir /opt/rtcds/caltech/c4/chans/daq/ and edit entries in C3CTN.ini, the setout and units are obvious from the file;
  3. find the daqd process with "$ps -e | grep daqd" and then kill it with "$kill -9 <PID from previous command>";
  4. Check with dataviewer, this can be launched by setting environment variable with "$LIGONDSIP=localhost" and then running "$dataviewer" on the fb4 machine.  You will need to have window forwarding on.

I have updated the environment temperature sensors and they should be logging properly now.  Unfortunately some channel name changes meant they were not logged over the weekend.

  2013   Sat Dec 16 22:01:25 2017 awadeDailyProgressTempCtrlTemperature sensors, vacuum can and surrounding environment

Now moved to the table.

Quote:

might be best to plant the room temperature sensor on the optical table rather than hang it in the air (less noisy)

 

  2012   Sat Dec 16 16:39:31 2017 awadeDailyProgressBEATTune up: Minor PLL and FSS tuning, lowering beat note amplitude

Adjustments to the BN amplitude: fixing saturation

I just checked the BN amplitude.  Its around 10 dBm (2.06Vpp) @ 170 Mhz which is way too high. This reduced down to about 6 dBm at about 59 MHz, not sure what changed.  Could have been the ISS or me changing the alignment somewhere in the beat board. I can see harmonics at x2 main peak, we are saturating.  

Optical power measured just after the combining beam splitter on the beat board was 86 uW North and 95 uW South.  After the beam splitter and a OD1.3 ND filter the total combined power was reduced to 19 µW total on the BN detector. For beat note signal of

 \small BN = 20\log_{10}[2\eta RG\sqrt{P_NP_S}] - 10log_{10}[50 \Omega]+30 dBm

where R = 0.8 A/W, G = 40e3, PN = PS = 10e-6 W, this would suggest an overlap effiecency of eta = 0.7. Ok I guess. However, RF power is too high for NF1811 detector, 

I have changed ND filter to 2.0OD before the beat detector to bring the RF beat note power down to -3 dBm.  Its a little bit on the low side but plenty to lock the PLL. I didn't have a single ND filter handy between these values that was clean enough to use.  Stacking multipe ND filters is a bad idea, given we are concerned with scatter etc.

I also checked power incident of the cavities: south = 1.39 mW and north  = 1.17 mW.


CCD cameras on the beat board

I inserted right angle connectors on the two CCD cameras to get the BNC cable out of the way of the NF1811 area. We need to angle the detector into the board so a black glass capture beam dump can be installed.  Right now the prompt beam from the NF1811 is reflected off the beat board and is not dumped.


Relocking PLL with -3 dBm BN, adding PID to heater shield

For now we want to optimize scatter, we can address issues of PLL noise floor later.  With the lower beat note the SR560 preamp has been turned up to gain = 100 (low noise mode) with no filters engaged.  Marconi slope was turned back up to 10kHz/V, we are nowhere near the noise floor for this actuation range in the <3 kHz band so there is no point in lowering this slope yet.  The loop is the standard 1/f OLG shape with a UGF of 30.5 kHz and phase 60 degrees.  Phase drops off pretty quickly after this point with 90 degrees at 49.8 kHz and 180 just above 100 kHz.   This will  limit turning it up further I guess, but is not a bit issue for the main band of interest.

I added some soft channels to implement a PID loop on the North cavity heater.  I have switched out the bench top power supply for a variable heater driver I built last week.  The bench top power supply was driving the beat note all over the place.  When I pointed a heat gun at the bench supply for 30 seconds, the beat note frequency responded strongly, moving by 10s of MHz. Although this is less scientific than comparing sensors from the vacuum can and room temp (we are measuring this now), it is good enough evidence for immediately ditching what should have always have been a temporary solution.

I haven't documented this new heater driver on the elog yet. I will try to get to this in the next few days.  It is essentially a Sallen-Key 2nd order LP filter set at 159 mHz corner followed by an op amp buffer wrapped around a IRF630 MOSFET.  So similar to Kevin and Kira's heater driver (PSL:1903) but with an active LP filter.  I used FET input op amps and used a 10 Ω sensing resistor (20 ppm/K), so the conversion is 0.1 A/V.  We need to heat the north cavity with about 0.8 W to get a beat note <150 MHz.  North heater resistance is 158 Ω.

Craig's Marconi netgpib re ranging script keeps the PLL locked, but periodically adjusts the carrier frequency of the Marconi. The software carrier frequency change kicks the PLL every time it reranges, so its not ideal. Knowing the VCO slope setting and and actuation voltage applied to the Marconi, provides enough information to estimate the current BN frequency.  There is already a soft channel dedicated to this (being continuously logged). The plan is to implement a PID to pull the BN to a set point.  The PLL can then be locked at all times with large deviations handled by the Marconi locker script and a PID pulling the BN to a desired value. Playing around by hand, it is possible to hold the BN to a value around 70 MHz with just the north cavity heater driver.

I've updated the PLL medm screen and added a heater driver PID as a service to acromag1.  The only issue is that the Marconi software locker is run as a bash line script, rather than a service with epics channels to configure it. The PID is blind to whether the PLL is actually locked and will rail if the Marconi locker dies.  Its something that will need babysitting in case PLL, FSS or other loops die. Also there is a sqrt(2) error in the kHz/V conversion that needs fixing.

New medm PLL screen, fast BN slew is me changing heater a lot.
  2011   Sat Dec 16 13:11:11 2017 ranaDailyProgressTempCtrlTemperature sensors, vacuum can and surrounding environment

might be best to plant the room temperature sensor on the optical table rather than hang it in the air (less noisy)

  2010   Fri Dec 15 14:19:39 2017 awade, CraigDailyProgressTempCtrlTemperature sensors, vacuum can and surrounding environment

We need to see how well the vac can PID temperature control is actually performing at suppressing room temperature fluctuations. We have four AD590 sensors on the vacuum can that have been logging for about a month now into the frame builder.  One sensor is used to control the temperature of the can and the other three are out of loop.  There is a slight reported gradient of about 1.05 K across the vacuum can from one side to the other.

However, we need environmental monitoring to see the degree to which the can temperature is perturbed by room temperature fluctuations.  I've placed one AD590 hanging just above the vacuum can, just below the hepafilter output. This will give the local air temperature about the experiment: currently reporting 15.48 C, which is pretty low. Another is placed next to the AC thermostat. 

We only have four transimpedance channels to work with, maybe in the future we would like to add a sensor on the table itself and right at the AC outlet so we can directly track the actuation into the room. I am updating the the frame builder to log these new channels so we have some data collected over the weekend.

  2009   Tue Dec 12 13:53:53 2017 awadeDailyProgressTempCtrlHeat circuits for cavities

We're finding that the beat note is slewing through the range of the PLL too quickly.  This is limiting the minimum practical actuation slope on the Marconi as the frequency of the center frequency reranging is often (preventing averaging of spectrum) and it limits the loop gain as we are constantly railing on the higher gain settings as the BN pushes towards the edge of the actuator limits.

I've made a cavity shield heater unit.  It consists of a Sallen-Key LP filter stage (1/f^2 starting at 159 mHz) along with a OP27 buffer rapped around IRF610 MOSFET sourcing ground through a 10 Ω (20 pp/C) wire wound resistor. Schematic is attached below. I looked at the noise introduced by the Sallen-Key, this is plotted in the second attachement.  

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