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  15505   Wed Jul 29 11:57:59 2020 ranaUpdateBHDIn-air BHD - CDS and wiring summary

3. I agree - this whitening will be handy to have for diagnostics.

4. I think in principle, we can ask a company to make the custom cables for us to save us some hand labor. Rich/Chub probably know the right companies to do small numbers of dirty cables.

5. Can't we just a single Noliac PZT in the same way that the OMC does? Or is the lead time too long?

6. Do we need active steering for this in-air test? I'm not even sure how we would get the alignment signal, so maybe that's a good reason to figure this out.

  15507   Thu Aug 6 00:34:38 2020 YehonathanUpdateBHDMonte Carlo Simulations

I've pushed an MCMC simulation to the A+ BHD repo (filename MCMC_TFs.ipynb). The idea is to show how random offsets around ideal IFO change the noise couplings of different DOFs to readout.

At each step of the simulation:

1. Random offsets for the different DOFs are generated from a normal distribution. The RMSs are taken from experimental data and some guesses and can be changed later. The laser frequency is tuned to match the CARM offset.

These are the current RMS detunings I use:

DOF RMS Taken from
DARM 10fm PHYSICAL REVIEW D 93, 112004 (2016), Table 2
CARM 1fm PHYSICAL REVIEW D 93, 112004 (2016), Table 2
MICH 3pm PHYSICAL REVIEW D 93, 112004 (2016), Table 2
PRCL 1pm PHYSICAL REVIEW D 93, 112004 (2016), Table 2
SRCL 10pm PHYSICAL REVIEW D 93, 112004 (2016), Table 2
OMCL 0.1pm Guess
OMC Breadboard angle 1\mu rad Guess
Differential arm loss 15ppm Guess
BHD BS imbalance 10% Guess
OMC finesse imbalance 5ppm Guess

2. A transfer function is computed for the noisy DOFs.

3. Projected noise is calculated.

These are the noise level for the DOFs:

DOF Noise Taken from
MICH 2e-16 m PHYSICAL REVIEW D 93, 112004 (2016), Fig 9
PRCL 0.5e-17 m PHYSICAL REVIEW D 93, 112004 (2016), Fig 9
SRCL 5e-16 PHYSICAL REVIEW D 93, 112004 (2016), Fig 9
OMCL 2.5e-17*(100/f)^(1/2) LIGO-G1800149
OMC Breadboard angle 1nrad Guess
RIN 2e-9 Optics Letters Vol. 34, Issue 19, pp. 2912-2914 (2009)

 

The attachments show the projected noise levels for the noisy DOFs. Each curve is a different instance of random offsets. The ideal case - "zero offsets" is also shown.

OMC Comm and OMC diff refer to the common and differential length change of the OMCs.

Attachment 1: MICH_Aplus_MCMC.pdf
MICH_Aplus_MCMC.pdf
Attachment 2: PRCL_Aplus_MCMC.pdf
PRCL_Aplus_MCMC.pdf
Attachment 3: SRCL_Aplus_MCMC.pdf
SRCL_Aplus_MCMC.pdf
Attachment 4: OMC_Comm_Aplus_MCMC.pdf
OMC_Comm_Aplus_MCMC.pdf
Attachment 5: OMC_Diff_Aplus_MCMC.pdf
OMC_Diff_Aplus_MCMC.pdf
Attachment 6: OMC_Angle_Yaw_Aplus_MCMC.pdf
OMC_Angle_Yaw_Aplus_MCMC.pdf
Attachment 7: OMC_Angle_Pitch_Aplus_MCMC.pdf
OMC_Angle_Pitch_Aplus_MCMC.pdf
Attachment 8: L0_RIN_Aplus_MCMC.pdf
L0_RIN_Aplus_MCMC.pdf
  15509   Fri Aug 7 11:23:47 2020 ranaUpdateBHDMonte Carlo Simulations

that's great. I think we would like to figure out how to present this so that its clear what the distribution of TFs is. Maybe we can plot the most likely curve as well as a shaded region indicating the 5% and 95% values?

Quote:

I've pushed an MCMC simulation to the A+ BHD repo (filename MCMC_TFs.ipynb). The idea is to show how random offsets around ideal IFO change the noise couplings of different DOFs to readout.

and then we add the loops

  15512   Mon Aug 10 07:13:00 2020 YehonathanUpdateBHDMonte Carlo Simulations

I fixed some stuff in the MCMC simulation:

1. Results are now plotted as shades from minimum to maximum. I tried making the shade the STD around a mean but it doesn't look good on a log scale when the STD is bigger than the mean.

2. Added comparison with aLigo. The OMCL diff and comm motions in A+ are both compared to the single OMCL DOF of aLigo.

3. I fixed a serious error in the code that produced incorrect results.

4. Imbalances in the IFO such as differential arm loss are generated randomly at the beginning and stay fixed for the rest of the simulation instead of being treated as an offset.

5. The simulation now runs with maxtem=2. That is, TEM modes up to 2nd order are considered.

The results are attached.

 

Attachment 1: MICH_AplusMCMC.pdf
MICH_AplusMCMC.pdf
Attachment 2: PRCL_AplusMCMC.pdf
PRCL_AplusMCMC.pdf
Attachment 3: SRCL_AplusMCMC.pdf
SRCL_AplusMCMC.pdf
Attachment 4: OMC_Comm_AplusMCMC.pdf
OMC_Comm_AplusMCMC.pdf
Attachment 5: OMC_Diff_AplusMCMC.pdf
OMC_Diff_AplusMCMC.pdf
Attachment 6: OMC_Angle_Yaw_AplusMCMC.pdf
OMC_Angle_Yaw_AplusMCMC.pdf
Attachment 7: OMC_Angle_Pitch_AplusMCMC.pdf
OMC_Angle_Pitch_AplusMCMC.pdf
Attachment 8: L0_RIN_AplusMCMC.pdf
L0_RIN_AplusMCMC.pdf
  15513   Mon Aug 10 16:52:04 2020 gautamUpdateBHDWorkable setup prepared

All the details are in E2000436, and documents linked from there, I think an elog would be much too verbose. In summary, a workable setup consisting of

  • 2 DCPDs interfaced with the realtime CDS system. Note that because this circuit is single-ended, while the AA and ADC are differential receiving, there is an overall gain of 0.5. Explicitly, for the 300 ohm DC transimpedance, the conversion is ~350 cts/mW.
  • A local oscillator beam delivered via fiber that is mode-matched (roughly) with the IFO AS beam.
  • A PZT mounted mirror to control the homodyne phase. The PZT (S320) is an obsolete part and it's hard to find a datasheet for it, but if its specs are comparable to the more modern S330, the full stroke is 10 um, for a max applied voltage of 100 V DC, so 100nm/V. c.f. 200V for 3um full stroke of the Noliac.

was prepared.

Last night, I locked the PRMI with the carrier resonant, and convinced myself that the DCPD null stream was sensing the MICH degree of freedom (while it was locked on AS55_Q) with good SNR below ~60 Hz. Above ~60 Hz, in this configuration, the ADC noise was dominating, but by next week, I'll have a whitening board installed that will solve this particular issue. With the optical gain of MICH in this configuration, the ADC noise level was equivalent to ~500 nrad/rtHz of phase noise above ~60 Hz (plots later).

Now, I can think about how to commission this setup interferometrically.

  15514   Tue Aug 11 23:20:29 2020 gautamUpdateBHDSome first tests with air BHD setup

Some tests done today:

All of these tests were done with the PRMI locked with carrier resonant in the recycling cavity (i.e. sidebands rejected to REFL port). I then actuated the BS length DOF with a sine wave at 311.1 Hz, 40 cts amplitude (corresponding to ~8 pm of peak-to-peak displacement).

  1. Attempt to balance the DCPDs
    • I tried to tune the digital gains of the two DCPDs so as to minimize the appearance of this line in the SUM channel
    • but no matter how I tuned the gains, I couldn't make the line in the SUM channel disappear entirely - in fact, the best I could do was to make the line height in SUM and NULL channels (yes I recognize the poor channel name choice, I'll change "NULL" to "DIFF" at the next model recompile) the same. See Attachment #1.
    • The lobes around the main peak are indicative of some scattering?
    • Attachment #2 shows a wider frequency range. The homodyne phase isn't controlled, so the "NULL" channel is not necessarily measuring the correct quadrature to be sensing MICH motion.
    • I think I can back out something about the contrast defect from this fact, but I need to go back to some modeling.
  2. A simple test of the homodyne phase actuator
    • I wanted to check that this PI S320 piezo actually allows me to actuate the optical path length of the local oscillator.
    • I'm using the OMC HV driver to drive said PZT - so there are two DAC channels available, one to dither the optic and one to apply a control signal. I think mainly this is to avoid using up DAC range for the dither signal, the overall dynamic range is still limited by the HV supply.
    • I can't find the maximum voltage that can be applied on the datasheet - so conservatively, I limited the HV output to saturate at 100 V DC, as this is the maximum for the S330 piezos used for green steering, for which there is a manual.
    • The S320 manual does say the full stroke of each PZT element is 10 um - so the actuation coefficient is ~100 nm/V. I then drove this actuator with a sine wave of 500 cts amplitude, at 314.1 Hz (corresponding to 15 nm of motion). With only the LO beam incident on the PDs, I saw no signal in either DCPD - as expected, so this was good.
    • Then, with the PRMI locked, I repeated the test. If there is no DC light field (as expected for the PRMI in this configuration), I wouldn't expect this drive signal to show up in the DCPDs. But in fact, I do. Again, this supports the presence of some (for now unquantified) contrast defect.

While it would seem from these graphs that the RIN of the LO beam at these frequencies is rather high, it is because of the ADC noise. More whitening (to be installed in the coming days) will allow us to get a better estimate, should be ~1e-6 I think.

I was just playing today, still need to setup some more screens, DTT templates etc to do more tests in a convenient way.

Now, I can think about how to commission this setup interferometrically.

Attachment 1: PRMI_RFlock.pdf
PRMI_RFlock.pdf
Attachment 2: PRMI_RFlock_fullscale.pdf
PRMI_RFlock_fullscale.pdf
  15532   Mon Aug 17 23:41:50 2020 gautamUpdateBHDWhitening and air BHD dark noise

Summary:

With the chosen transimpedance of 300 ohms, in order to be able to see the shot noise of 10 mW of light in the digitized data streams, we'd need all 3 stages of whitening. If we want to be shot noise limited with 1 mW of LO light, we'd need to increase said transimpedance I think.

Details:

The measurements were taken with

  1. No light incident on the DCPDs.
  2. The flat whitening gain was set to 0 dB.
  3. Whitening engaged sequentially, stage by stage, shown as (Blue, Red, Orange and Green) curves corresponding to (0, 1, 2, 3) stages of whitening.

Of course, it's unlikely we're going to be shot noise limited for any configuration in the short run. But this was also a test of 

  1. My soldering.
  2. Change of whitening corner frequencies.
  3. Test of the overall whitening board assembly.

All 3 tests passed.

Attachment 1: BHD_whitening.pdf
BHD_whitening.pdf
  15535   Fri Aug 21 15:27:00 2020 gautamUpdateBHDBetter BHD mode-matching

Summary:

The mode-matching between the LO and AS beams is now ~50%. This isn't probably my most average mode-matching in the lab, but I think it's sufficient to start doing some other characterization and we can try squeezing out hopefully another 20-30% by putting the lenses on translation stages, tweaking alignment etc.

Details:

The main change was to increase the optical path length of the IFO AS path, see Attachment #1. This gave me some more room to put a lens and translate it.

  • The LO path uses two lenses, f=200mm and f=100mm to focus the collimator output beam, which is supposedly ~1200um diameter, to something like 400um diameter (measured using beam profiler but not very precisely).
  • This beam is  fairly well collimated, and the beam size is close to what the PMC cavity will want, I opted not to tweak this too much more.
  • For the AS beam, the single bounce reflection from ITMY was used for alignment work.
  • There is a 2" f=600mm lens upstream (not seen in Attachment #1). This supposedly makes a beam with waist ~80um, but I couldn't numerically find a good solution numerically if this assumption is true, so I decided to do the mode-matching empirically.
  • A single f=150mm lens got me a beam that seemed pretty well collimated, and roughly the same size as the LO beam, so I opted to push ahead with that. Later, I measured with the beam profiler that the beam is ~600um in diameter, so the beam isn't very well matched to the LO spot size, but I decided to push ahead nevertheless.
  • Patient alignment work enabled me to see interference fringes.
    • Note that the ITM reflection registers 30 cts (~80 uW). Assuming 800mW transmission through the IMC, I would have expected more like 800mW * 5.637% * 50% * 98.6% * 50% * 10% * 30% * 50% * 50% = 80uW, so this is reasonable I guess. The chain of numbers corresponds to T_PRM * T_BS * R_ITM * R_BS * T_SRM * T_vac_OMC_pickoff * R_in_air_BS * R_homodyneBS.
    • The IFO AS beam appears rather elliptical to the eye (and also on the beam profiler). It already looks like this coming out of the vacuum so not much we can do about it right now I guess. By slightly rotating the f=150mm focusing lens so that the beam going through it at ~10 degrees instead of normal incidence, I was able to get a more circular beam as measured using the beam profiler.
    • With the AS beam blocked, the LO beam registers 240 cts on each DCPD (~0.7 mW). 
    • The expected fringe should then be (sqrt(240) + sqrt(30))^2 - (sqrt(240) - sqrt(30))^2 ~ 440 cts pp.
    • The best alignment I could get is ~200 cts pp, see Attachment #2.

Next steps:

Try the PRMI experiments again, now that I have some confidence that the beams are actually interfering.

See Attachment #3 for the updated spectra - the configuration is PRMI locked with carrier resonant and the homodyne phase is uncontrolled. There is now much better clearance between the electronics noise and the MICH signal as measured in the DCPDs. The "LO only" trace is measured with the PSL shutter closed, so the laser frequency isn't slaved to the IMC length. I wonder why the RIN (seen in the SUM channel) is different whether the laser is locked to the IMC or not? The LO pickoff is before the IMC.

Attachment 1: IMG_7548.JPG
IMG_7548.JPG
Attachment 2: BHD_MM.png
BHD_MM.png
Attachment 3: PRMI_DCPDs.pdf
PRMI_DCPDs.pdf
  15539   Tue Aug 25 05:51:29 2020 YehonathanUpdateBHDMonte Carlo Simulations

I re-plotted the MCMC results as semi-transparent lines so that probable lines stick out.

This also reveals what is behind the extreme sparsity in the aLIGO simulation results (In the previous post).

There seem to be some bi-stability/phase transition/whatever in the aLIGO simulation. The aLIGO transfer functions are very sensitive to one or more of the DOFs. Not sure which yet.

Attachment 1: MICH_AplusMCMC.pdf
MICH_AplusMCMC.pdf
Attachment 2: PRCL_AplusMCMC.pdf
PRCL_AplusMCMC.pdf
Attachment 3: SRCL_AplusMCMC(1).pdf
SRCL_AplusMCMC(1).pdf
Attachment 4: OMC_Diff_AplusMCMC.pdf
OMC_Diff_AplusMCMC.pdf
Attachment 5: OMC_Comm_AplusMCMC.pdf
OMC_Comm_AplusMCMC.pdf
Attachment 6: OMC_Angle_Yaw_AplusMCMC.pdf
OMC_Angle_Yaw_AplusMCMC.pdf
Attachment 7: OMC_Angle_Pitch_AplusMCMC.pdf
OMC_Angle_Pitch_AplusMCMC.pdf
Attachment 8: Main_Laser_RIN_AplusMCMC.pdf
Main_Laser_RIN_AplusMCMC.pdf
  15540   Wed Aug 26 00:52:55 2020 gautamUpdateBHDBHD activities

Listing some talking points from the last week of activity here.

  1. LO delivery fiber cable may be damaged.
    • The throughput itself doesn't suggest any problems, I get almost all the light I put in out the other end.
    • However, even when I slightly move the fiber, I see huge amplitude fluctuations in the DCPD readouts. This shouldn't be the case, particularly if the light is well matched to one of the special axes of the PM fiber. I checked with a PBS at the output that this is indeed the case, so something else must be funky?
    • In any case, I don't think it's a great idea to use this 70m long fiber for bringing the light from the PSL table to the adjacent AP table. Chub has ordered a 10m patch cable.
    • I was a bit too hasty this morning, thinking we had a patch cable in hand, and so I removed the fiber from the AP table. So right now, the LO beam doesn't make it to the BHD setup. Depending on the lead time for the new patch cable, I may or may not resurrect this old setup.
    • I have also located some foam and rigid plastic tubing which I think will help in isolating the fiber from environmental length(phase) modulation due to acoustic pickup.
  2. BHD commissioning activities
    • Basically, I've been trying to use the Single Bounce ITM reflection/ Michelson / PRMI with carrier locked to get some intuition about the BHD setup. These states are easily prepared, and much easier to understand than the full IFO for these first attempts.
    • One concern I have is the angular stability (or lack thereof). When the PRMI is locked, the DC light level on each DCPD fluctuates between ~0 (which is what it should be), up to ~30 cts (~85uW).
    • Using the empirically determined attenuation factor between the DCPDs and the dark port of the beamsplitter, I estimate the power can be as high as 20mW. This is a huge number, considering the input to the interferometer is ~800mW. I assume that all the light is at the carrier frequency, since the PRC should reject all the sideband light in this configuration. In any case, the total amount of sideband light is ~20mW, and the carrier stays resonant in the PRC even when there are these large ASDC excursions, so I think it's a reasonable assumption that the light is at the carrier frequency. Moreover, looking at the camera, one can see a clear TEM10/01 profile, indicative of imperfect destructive interference at the beamsplitter due to beam axis misalignment.
    • The effect of such excursions on the BHD readout hasn't yet been quantified (by me at least), but I think it may be hampering my attempts to dither the homodyne phase to estimate the LO phase noise.
  3. High voltage coil driver project - see thread for updates.
  4. Trek HV driver has arrived.
    • I haven't opened the box yet, but basically, what this means is that I can dither the mirror intended for homodyne phase control in a reasonable way.
    • Previously, I was using the OMC HV driver to drive the PZTs - but this dither signal path has a 2kHz high pass filter (since the OMC length dither is a kHz dither). I didn't want to futz around with the electronics, particularly since the unit was verified to be working.
    • So the plan now would be to drive the input of the Trek with a DAC output (an appropriate AI chassis has been prepared to interface with the CDS system).
    • Hopefully, there's enough DAC dynamic range to dither the PZT and also do the homodyne phase locking using a single channel. Else, we'd need to use two channels and install a summing amplifier.
    • We definitely need more high-voltage amplifiers/supplies in the lab:
      • Any Thorlabs HV drivers we can recover? 
      • Eventually, we will need HV for coil drivers, OMC PZTs, steering PZTs, homodyne phase control PZT. 
  5. PMC bases have arrived.
    • Joe Benson from the machine shop informed me today afternoon that the bases were ready for pickup.
    • We have 3 bases in hand now. The finish isn't the greatest in the world, but I think it'll work. You can see some photos here.
    • I will hold off on putting this together while I work on the basic airBHD commissioning tests. We can install the PMCs later.
  6. AS port WFS project
    • We now have in hand almost all the components for stuffing the ISC whitening and LSC demod boards.
    • Rich, Chub, Luis and I had a call on Monday. The advise from Rich/Luis was:
      • Choose an inductance that has Z~100 ohms at the frequency of interest, for the resonant transimpedance part.
      • Choose a capacitance that gives the appropriate resonant frequency.
      • Don't stuff more notches than you need - start with just a 2f notch (so 110 MHz for us), and make sure to place the highest frequency notch closest to the photodiode.
      • Rich also suggested looking at the optical signal with a non-optimized head, get an idea of what the field content is, and then tune the circuit as necessary. There are obviously going to be many issues that only become apparent once we do such a test.
    • The aLIGO modulation frequencies are only 20% different from the 40m modulation frequencies. So I thought it is best if for our first pass, we stick to the inductance values used in the aLIGO circuits (same footprint, known part etc etc). Then, we will change the capacitance so that we have a tuning range that is centered our modulation frequencies.
    • The parts have been ordered.
  7. ISS project
    • Half of the LO light on the BHD breadboard is diverted for the purpose of sensing the LO intensity noise, for eventual stabilization. Right now, it is just getting dumped.
    • A PD head has been located. It has a minimalist 1kohm transimpedance amplifier circuit integrated into the head.
    • Our AOM driver has an input range of 0-1V DC. We want to map the servo output of +/-10V DC (or +/-4V DC if we use an SR560 based servo for a first pass) to this range.
    • I wanted to do this for once in a non-hacky way so I drew up a circuit that I think will serve the purpose. It has been fabricated and will be tested on the bench in a couple of days.
    • Once I get a feel for what the signal content is, I will also draw up a interface board to the PD head that (i) supplies the reverse bias voltage and +/-15 V DC to the PD head and (ii) applies some appropriate HPF action and provides a DC monitor as well.
  8. Summary pages are dead.
  9. General lab cleanup
    • I moved all the PPE from the foyer area into the designated cabinets along the east arm.
    • Did some basic cleanup of the lab in preparation for crane inspection. Walkways are clear.
    • I de-cluttered the office area a bit, but today I received ~10 packages from Digikey/FrontPanelExpress etc. So, in fact, it got even more cluttered. Entropy will go down once we ship these off to screaming circuits for stuffing the PCBs.
  15545   Fri Aug 28 23:33:38 2020 gautamUpdateBHDSome more hardware changes

Just a quick set of notes detailing changes so that there are no surprises, more details to follow.

  1. Trek driver has been temporarily placed on top of the KEPCO supply east of the OMC electronics rack. Cabling to it has been laid out as well. I turned both off so neither should be energized now.
  2. A new AI chassis (and associated cabling including the DAC SCSI cable and +/-24 V DC cable) has been installed in 1X2.
  3. To map the DAC range to what the Trek driver wants, I've configured the inverting summing amplifier with gain of 1/8. The offset voltage is set to 5V DC instead of 10V as intended, because the DAC can only drive +/-5 V when connected to a single ended receiving/sending unit.
  4. The LO delivery fiber was re-laid, and the interference between the IFO AS beam and LO beams were restored.

I briefly tried some LO PZT mirror dithering tonight, but didn't see the signal. Needs more troubleshooting.

  15549   Sat Aug 29 22:46:29 2020 gautamUpdateBHDNew homodyne-phase control electronics

Summary:

The electronics chain used to drive the three elements of the PI PZT on which a mirror is mounted with the intention of controlling the LO phase has been changed, to now use the Trek Mode603 power amplifier instead of the OMC high voltage driver. Attachment #1 shows the new configuration.

Details:

The text of Attachment #1 contains most of the details. The main requirement was to map the DAC output voltage range, to something appropriate for the Trek amplifier. The latter applies a 50V/V gain to the signal received on its input pin, and also provides a voltage monitor output which I hooked up to an ADC channel in c1ioo. The gain of the interfacing electronics was chosen to map the full output range of the DAC (-5 to +5 V for a single-ended receiving config in which one pin is always grounded) to 0-2.5 V at the input of the Trek amplifier, so that the effective high voltage drive range is 0-125 V. I don't know what the damage threshold is for the PI PZT, maybe we can go higher. The only recommendation given in the Trek manual is to not exceed +/-12 V on the input jack, so I have configured D2000396 to have a supply voltage of 11.5 V, so that in the event of electronics failure, we still don't exceed this number.

On the electronics bench, I tested the drive chain, and also measured the transfer function, see Attachment #2. Seems reasonable (the Trek amplifier was driving a 3uF capacitive load used to protect the SR785 measurement device from any high voltage, hence the roll-off). The gain of D2000396 was changed from 1/8 to 1/4 after I realized that the DAC full range is only +/- 5 V when the receiving device is single-ended at both input and output. Maybe the next iteration of this curcuit should have differential sending, to preserve the range.

Testing:

To test the chain, I used the single bounce beam from the ITM, and interfered it with the LO. Clear fringing due to the seismic motion of the ITM (and also LO phase noise) is visible. In this configuration, I drove the PZT mirror in the LO path at a higher frequency, hoping to see the phase modulation in the DCPD output. However, I saw no signal, even when driving the PZT with 50% of the full DAC range. The voltage monitor ADC channel is reporting that the voltage is faithfully being sent to the PZTs, and I measured the capacitance of the PZTs (looked okay), so not sure what is going on here. Needs more investigation.

Update Aug 30 5pm: Turns out the problem here was a flaky elbow connector I used to pipe the high voltage to the PI PZT, it had some kind of flaky contact in it which meant the HV wasn't actually making it to the PZT. I rectified this and was immediately able to see the signal. Played around with the dark fringe Michelson for a while, trying to lock the homodyne phase by generating a dither line, but had no success with a simple loop shape. Probably needs more tuning of the servo shape (some boosts, notches etc) and also the dither/demod settings themselves (frequency, amplitude, post mixer LPF etc). At least the setup can now be worked on interferometrically.

Attachment 1: zetaDrive.pdf
zetaDrive.pdf
Attachment 2: trekTFs.pdf
trekTFs.pdf
  15553   Wed Sep 2 00:49:47 2020 gautamUpdateBHDSome notes about homodyne phase

Summary:

Using a heterodyne measurement setup to track both quadratures, I estimated the relative phase fluctuation between the LO field and the interferometer output field. It may be that a single PZT to control the homodyne phase provides insufficient actuation range. I'll also need to think about a good sensing scheme for controlling the homodyne phase, given that it goes through ~3 fringes/sec - I didn't have any success with the double demodulation scheme in my (admittedly limited) trials.

For everything in this elog, the system under study was a simple Michelson (PRM, SRM and ETMs misaligned) locked on the dark fringe.

Details:

​This work was mainly motivated by my observation of rapid fringing on the BHD photodiodes with MICH locked on the dark fringe. The seismic-y appearance of these fringes reminded me that there are two tip-tilt suspensions (SR2, SR3), one SOS (SRM) + various steering optics on seismic stacks (6+ steering mirrors) between the dark port of the beamsplitter and the AS table, where the BHD readout resides. These suspensions modulate the phase of the output field of course. So even though the Michelson phase is tightly controlled by our LSC feedback loop, the field seen by the homodyne readout has additional phase noise due to these optics (this will be a problem for in vacuum BHD too, the question is whether we have sufficient actuator range to compensate).

To get a feel for how much relative phase noise there is between the LO field and the interferometer output field (this is the metric of interest), I decided to set up a heterodyne readout so that I can simultaneously monitor two orthogonal quadratures.

  • The idea is that with the Michelson locked, there is no DC carrier field from the interferometer.
  • The field incident on the DCPD from the interferometer should be dominated by the 55 MHz sideband transmitted to the dark port given the Schnupp asymmetry. 
  • The LO field is picked off before any RF sidebands are added to it (the PMC modulation sideband should be suppressed by the cavity transmission).
  • Therefore, the LO field should be dominantly at the carrier frequency.
  • By placing a broadband RFPD (PDA10CF) in place of one of the DCPDs, I can demodulate the optical beat between this 55 MHz sideband, which shares the same output path to the location of the DCPD as the audio-frequency sidebands on the carrier from the dark Michelson, to estimate the relative phase noise between the LO and IFO output fields.
  • The point is that with the heterodyne readout, I can track the fringe wrapping, which is not an option for the BHD readout with two DCPDs, and uncontrolled LO phase.

Attachment #1 shows the detailed measurement setup. I hijacked the ADC channels normally used by the DCPDs (along with the front-end whitening) to record these time-series.

Attachments #2, #3 shows the results in the time domain. The demodulated signal isn't very strong despite my pre-amplification of the PDA10CF output by a ZFL-500-HLN, but I think for the purposes of this measurement, there is sufficient SNR.

This would suggest that there are pretty huge (~200um) relative phase excursions between the LO and IFO fields. I suppose, over minutes, it is reasonable that the fiber length changes by 100um or so? If true, we'd need some actuator that has much more range to control the homodyne phasethan the single PZT we have available right now. Maybe some kind of thermal actuator on the fiber length? If there is some pre-packaged product available, that'd be best, making one from scratch may be a whole project in itself. Attachment #3 is just a zoomed-in version of the time series, showing the fringing more clearly.

Attachment #4 has the same information as Attachment #2, except it is in the frequency domain. The FFT length was 30 seconds. The features between ~1-3 Hz support my hypothesis that the SR2/SR3 suspensions are a dominant source of relative phase noise between LO and IFO fields at those frequencies. I guess we could infer something about the acoustic pickup in the fibers from the other peaks.

Attachment 1: heterodyneMICH.pdf
heterodyneMICH.pdf
Attachment 2: unwrappedPhase.pdf
unwrappedPhase.pdf
Attachment 3: unwrappedPhase_zoom.pdf
unwrappedPhase_zoom.pdf
Attachment 4: phaseNoisePSD.pdf
phaseNoisePSD.pdf
  15554   Thu Sep 3 00:00:57 2020 gautamUpdateBHDNew patch cable installed
  • 10m PM1064 cable was installed. I tried a double shielding approach (photos tmrw here), but I suspect the real weak point is where the fiber is plugged into the collimator - it's hard to imagine we can stabilize this sort of arrangement to better than 100um absolute length over long periods of time, I'd think thermal/mechanical strains in the fiber will modulate the length by ~mm (?). Anyways, let's see what the heterodyne measurement tells us.
  • This work required (i) realignment at the input coupler and (ii) change of position of mode matching lenses in the LO path on the AS table to see any interference with the IFO beam. This indicates something was seriously wrong with the old patch cable, as the collimator should set the mode. The MFD of the two fibers may have been different, but I don't know if that alone can account for it.
  • As of now, I have fringes between the ITM single bounce and the LO, but the fringe pk-pk is only 10% of the theoretical pk-pk based on DC values of the LO and AS beams. So the mode matching can be improved significantly (I preivously had ~60%).

To be continued tomorrow. I think it's a good idea to let the newly installed fiber relax into some sort of stable configuration overnight.

  15555   Thu Sep 3 15:55:04 2020 gautamUpdateBHDPhase drift between LO and IFO after fiber replacement

Summary:

After replacement of the fiber delivering the LO beam to the airBHD setup (some photos here), I repeated the measurement outlined here. There may be some improvement, but overall, conclusions don't change much.

Details:

The main addition I made was to implement a digital phase tracker servo (a la ALS), to make sure my arctan2 usage wasn't completely bonkers (the CDS block can be deleted later, or maybe it's useful to keep it, we will see). I didn't measure it today, but the UGF of said servo should be >100 Hz so the attached spectrum should be valid below that (loop has not been done, so above the UGF, the control signal isn't a valid representative of the free running noise). Attachment #1 shows the result. The 1 Hz and 3 Hz suspension resonances are well resolved. Anyways, what this means is that the earlier result was not crazy. I don't know what to make of the high frequency lines, but my guess is that they are electronic pickup from the Sorensens - I'm using clip-mini-grabbers to digitize these signals, and other electronics in that rack (e.g. ALS signals) also show these lines.

It is pretty easy to keep the simple Michelson locked for several minutes. Attachment #2 shows the phase-tracker servo output over several minutes. The y-axis units are degrees. If this is to be believed, the relative phase between the two fields is drifting by 12um ove an hour. This is significantly lower than my previous measurement, while the noise in the ~0.5-10 Hz band is similar, so maybe the shorter fiber patch cable did some good?

I think there is also correlation between the PSL table temperature, but of course, the evidence is weak, and there are certainly other effects at play. At first, I thought the abrupt jumps are artefacts, but they don't actually represent jumps >360 degrees over successive samples, so maybe they are indicative of some real jump in the relative phase? Either fiber slippage or TT suspension jumps? I'll double check with the offline data to make sure it's not some artefact of the phase tracker servo. If you disagree with these conclusions and think there is some meaurement/analysis/interpretation error, I'd love to hear about it.

Next steps:

  1. Budget the offline inferred phase noise spectrum, overlay a seismic noise model, to see if we can disentangle the contributions from the suspensions and that from the LO fiber.
  2. I'll see if I can setup an LO pickup with some RF sidebands on it in parallel to this setup so we can try some of the ideas discussed on the call this week. There are several beams available, but the question is whether I can get this into a fiber without 1 week of optical layout work.

I have left the heterodyne electronics setup at the LSC rack, but it is not powered (because there are some exposed wires). Please leave it as is.

Attachment 1: LOphaseDrift.pdf
LOphaseDrift.pdf
Attachment 2: phaseDrift_tempCorr.png
phaseDrift_tempCorr.png
  15562   Mon Sep 7 23:49:14 2020 gautamUpdateBHDA first look at RF44 scheme

Summary:

Over the last couple of days, I've been working towards getting the infrastructure ready to test out the scheme of sensing (and eventually, controlling) the homodyne phase using the so-called RF44 scheme. More details will be populated, just quick notes for now before I forget.

  • LO beam with RF sidebands needed to be re-coupled into collimator, it wasn't seated tightly and just touching the fiber completely destroyed the alignment.
  • HWP installed before said collimator - IMC wants s-polarized light whereas the IFO field is p-polarized.
  • After my work, the numbers were: ~1.47mW input to collimator, ~1.07mW out of collimator on AS table, ~1mW making it to the BHD board. All seem like reasonable numbers to me.
  • 44 MHz signal synthesis - for now, I use a Marconi (10 MHz synced to Rb clock), I think we could also use a mixer+SLP50 to mix 11 and 55 MHz signals (which are easily available at the LSC rack) to generate this. I looked at Wenzel quadruplers, the specs don't suggest a quadrupler will do much better.
  • CDS model was modified to accept the phase-tracker output as an error signal for the homodyne phase control servo. Compile and install went smooth but I opted against a model restart tonight, I'll do it tmrw.
  • Some trials were done with the Michelson locked to a dark fringe (as was done for the case of the DC LO field beating with the 55 MHz sideband). While the overall spectrum lines up fairly well with earlier trials, the signal looks somewhat more "discontinuous" in its traversal of I/Q space, and it never quite goes to 0. Some offset? What does this mean for locking? More investigations needed....
  15563   Tue Sep 8 01:31:43 2020 KojiUpdateBHDA first look at RF44 scheme

- Loose fiber coupler: Sorry about that. I could not detect something was loose there, although some of the locks were not tightened.

- S incident instead of P: Sorry about that too. I completely missed that the IMC takes S-pol.

  15565   Wed Sep 9 00:05:18 2020 gautamUpdateBHDMore notes on the RF44 scheme

Summary:

  1. With the Michelson locked on a dark fringe, the f2-f1 signal at ~44 MHz does not seem to ever vanish, it seems to bottom out at ~2mV DC. Is this just an electronics offset? Not sure of the implications on using this as a locking signal for the homodyne phase yet.
  2. The inferred relative phase fluctuations between the LO and RF fields using this 44 MHz signal is consistent with that from previous tests.
  3. The laying out of the new, shorter, fiber patch cable seems to have helped to reduce the phase drift over minute time scales.
  4. So far, I have not had any success in using the 44 MHz signal to close a servo loop and keep the homodyne phase locked for more than a few seconds at a time, and even then, the loop shape is sub-optimal as the in-loop error signal is not clean. Maybe some systematic loop shaping will help, but I think the dynamic range requirement on the actuator is too high, and I'm not sure what to make of the fact that the error signal does not vanish.

Details:

Attachment #1 shows the optical setup currently being used to send the LO field with RF sidebands on it to the air BHD setup.

  • You can find a video of the large power fluctuations mentioned in my previous elog here. After tightening the collimator in the mount, the arrangement is still rather sensitive, but at least I was able to see some light on the DCPD on the AS table, at which point I could use this signal and tweak the alignment to maximize it.
  • It is well known that the input beam to the IMC drifts during the day, either due to temperature fluctuations / PMC PZT stroke L2A / some other reason (see Attachment #4 for the power drift over ~12 hours, it is not monotonic with temperature). The fact that our collimating setup is so sensitive to the input pointing isn't ideal, but I noticed the power had only degraded by ~5% today compare to yesterday, so maybe the occassional touch up is all that is required.

Attachment #2 shows spectra of the relative phase drift between LO and IFO output field (from the Dark Michelson). 

  • I still haven't overlaid a seismic model. There was some discussion about the TTs having a 1/f roll-off as opposed to 1/f^2, I don't know if there was any characterization at the time of installation, but this SURF report seems to suggest that it should in fact be 1/f^2 because the passive eddy current dampers are mounted to the main suspension cage on springs rather than being rigidly attached. 
  • The noise at ~100 Hz is ~x2 higher if the spectra is collected during the daytime, when the seismic activity is high. Although this shouldn't really matter at 100 Hz? 
  • There are also huge power-line harmonics - I suspect these are making it difficult to close a feedback loop, as I couldn't add a 60 Hz comb which doesn't affect the loop stability for a UGF of ~30-50 Hz. But if they aren't notched out, the control signal RMS is dominated by these frequencies.

Attachment #3 shows the signal magntiude of the signals used to make the spectra in Attachment #2, during the observation time (10 minutes) with which the spectra were computed. The dashed vertical lines denote the 1%, 50% and 99% quantiles.

  • Koji asked me about the 55 MHz signal and why it doesn't vanish - for the dark Michelson, where the ITMs don't apply any relative phase on reflection to the carrier and RF sideband fields, we expect that the upper and lower sidebands cancel, and so there should be no intensity modulation at 55 MHz (just like we don't expect any for a pure phase modulated light field incident on a photodiode).
  • However, from the I/Q demodulated data that is collected, it would appear that while the size of the signal does vary, it doesn't ever completely vanish. This implies some asymmetry in the sidebands (or at least, the transmission of the sidebands by the Michelson). I didn't estimate the effect of the Schnupp asymmetry, or if this asymmetry is coming from elsewhere, but the point is that for the conclusions drawn from Attachment #2 remain valid even though both the amplitude and phase of the 55 MHz signal is changing. 
  • I also plot the corresponding histogram for the 44 MHz signal. You can see that it never goes to 0 (once I fix the x-label ticks). I don't know if this is consistent with some electronics offset.

Attempts to close a feeddback loop to control the homodyne phase:

  • A digital PLL (a.k.a. Phase Tracker) servo was used to keep the demodulated 44 MHz signal in one (demodulated) quadrature, which can then be used as an error signal.
  • Unlike the ALS case, the quantity to be servoed to 0 is the magnitude of the 44 MHz signal, and not its phase, so that's how I've set up the RTCDS model.
  • I played around with the loop shape to try and achieve a stable lock by actuating on the PZT mounted mirror in the LO path - however, I've not yet had any success so far.
Attachment 1: IMG_3397.JPG
IMG_3397.JPG
Attachment 2: phaseNoisePSD.pdf
phaseNoisePSD.pdf
Attachment 3: magnitudeHist.pdf
magnitudeHist.pdf
Attachment 4: LOpowerDrift.png
LOpowerDrift.png
  15567   Thu Sep 10 15:43:22 2020 JonUpdateBHDInput noise spectra for A+ BHD modeling

As promised some time ago, I've obtained input noise spectra from the sites calibrated to physical units. They are located in a new subdirectory of the BHD repo: A+/input_noises. I've heavily annotated the notebook that generates them (input_noises.ipynb) with aLOG references, to make it transparent what filters, calibrations, etc. were applied and when the data were taken. Each noise term is stored as a separate HDF5 file, which are all tracked via git LFS.

So far there are measurements of the following sources:

  • L1 SRCL
  • H1 SRCL
  • L1 DHARD PIT
  • L1 DSOFT PIT
  • L1 CSOFT PIT
  • L1 CHARD PIT

These can be used, for example, to make more realistic Hang's bilinear noise modeling [ELOG 15503] and Yehonathan's Monte Carlo simulations [ELOG 15539]. Let me know if there are other specific noises of interest and I will try to acquire them. It's a bit time-consuming to search out individual channel calibrations, so I will have to add them on a case-by-case basis.

  15569   Mon Sep 14 07:50:01 2020 YehonathanUpdateBHDMonte Carlo Simulations

Turns out what was causing the instability in the aLIGO plots were the lock commands which I forgot to remove before running the simulation. Removing these also made the simulation much faster.

Other than that I improved other stuff in the simulations:

  • The LO phase in the aPlus simulation is now optimized for the lowest noise at 100Hz.
  • Added RF PDs diagnostics (see attachments 8 for aPlus and 9 for aLIGO). The thresholds (red dashed lines in attachments 8,9) for cutting marginal simulations are set such that roughly 30% of the simulations are discarded.
  • Removed DHARD because it jacks up the RF PD readings in aPlus for some reason.
  • Fixed the sign of laser frequency shift in response to CARM offset.

Still need to do:

  • Incorporate Jon’s noise curves.
  • Add phase noise for LO beam.
  • Add arm reflectivity imbalance.
  • Add mirror curvature imbalance.
  • Include feedback loops using Pytickle.

Feel free to add to the todo list.

Attachment 1: MICH_AplusMCMC.pdf
MICH_AplusMCMC.pdf
Attachment 2: PRCL_AplusMCMC.pdf
PRCL_AplusMCMC.pdf
Attachment 3: SRCL_AplusMCMC(1).pdf
SRCL_AplusMCMC(1).pdf
Attachment 4: OMC_Comm_AplusMCMC.pdf
OMC_Comm_AplusMCMC.pdf
Attachment 5: OMC_Diff_AplusMCMC.pdf
OMC_Diff_AplusMCMC.pdf
Attachment 6: OMC_Angle_Yaw_AplusMCMC.pdf
OMC_Angle_Yaw_AplusMCMC.pdf
Attachment 7: OMC_Angle_Pitch_AplusMCMC.pdf
OMC_Angle_Pitch_AplusMCMC.pdf
Attachment 8: Main_Laser_RIN_AplusMCMC.pdf
Main_Laser_RIN_AplusMCMC.pdf
Attachment 9: aPlus_RF_Diagnostics.pdf
aPlus_RF_Diagnostics.pdf
Attachment 10: aLIGO_RF_Diagnostics.pdf
aLIGO_RF_Diagnostics.pdf
  15575   Tue Sep 15 22:11:52 2020 gautamUpdateBHDMore notes on the RF44 scheme

Summary:

After more trials, I think the phase tracker part used to provide the error signal for this scheme needs some modification for this servo to work.

Details:

Attachment #1 shows a block diagram of the control scheme.

I was using the "standard" phase tracker part used in our ALS model - but unlike the ALS case, the magnitude of the RF signal is squished to (nearly) zero by the servo. But the phase tracker, which is responsible for keeping the error signal in one (demodulated) quadrature (since our servo is a SISO system) has a UGF that is dependent on the magnitude of the RF signal. So, I think what is happening here is that the "plant" we are trying to control is substantially different in the acquisition phase (where the RF signal magnitude is large) and once the lock is enabled (where the RF signal magnitude becomes comparitively tiny).

I believe this can be fixed by dynamically normalizing the gain of the digital phase tracking loop by the magnitude of the signal = sqrt(I^2 + Q^2). I have made a modified CDS block that I think will do the job but am opting against a model reboot tonight - I will try this in the daytime tomorrow. 

I'm also wondering how to confirm that the loop is doing something good - any ideas for an out-of-loop monitor? I suppose I could use the DCPD - once the homodyne phase loop is successfully engaged, I should be able to drive a line in MICH and check for drift by comparing line heights in the DCPD signal and RF signal. This will requrie some modification of the wiring arrangement at 1Y2 but shouldn't be too difficult...


The HEPAs, on the PSL table and near ITMY, were dialled down  / turned off respectively, at ~8pm at the start of this work. They will be returned to their previous states before I leave the lab tonight.

Attachment 1: RF44.pdf
RF44.pdf
  15579   Fri Sep 18 10:47:48 2020 gautamUpdateBHDSensing scheme for homodyne phase

eSummary:

I don't think the proposed scheme for sensing and controlling the homodyne phase will work without some re-thinking of the scheme. I'll try and explain my thinking here and someone can correct me if I've made a fatal flaw in the reasoning somewhere.

Field spectrum cartoon:

Attachment #1 shows a cartoon of the various field components.

  • The input field is assumed to be purely phase modulated (at 11 MHz and 55 MHz) creating pairs of sidebands that are in quadrature to the main carrier field.
  • The sideband fields are drawn with positive and negative imaginary parts to indicate the relative negative sign between these terms in the Jacobi-Anger expansion.
  • For our air BHD setup, the spectrum of the LO beam will also be the same.
  • At the antisymmetric (= dark) port of the beamsplitter, the differential mode signal field will always be in the phase quadrature.
  • I'm using the simple Michelson as the test setup:
    • The ITMs have real and (nearly) identical reflectivities for all frequency components incident on it.
    • The sideband fields are rotated by 90 degrees due to the i in the Michelson transmission equation.
    • The Schnupp asymmetry preferentially transmits the 55 MHz sideband to the AS port compared to the 11 MHz sideband - note that in the simple Michelson config, I calculate T(11 MHz) = 0.02%, T(55 MHz) = 0.6% (both numbers not accounting for the PRM attenuation).
  • I think the cartoon Hang drew up is for the DRFPMI configuration, with the SRC operated in RSE.
    • The main difference relative to the simple Michelson is that the signal field picks up an additional 90 degrees of phase propagating through the SRC.
  • For completeness, I also draw the case of the DRFPMI where the SRC is operated at nearly the orthogonal tuning.
    • I think the situation is similar to the simple Michelson

So is there a 90 degree relative shift between the signal quadrature in the simple Michelson vs the DRFPMI? But wait, there are more problems...

Closing a feedback loop using the 44 MHz signal:

We still need to sense the 44 MHz signal with a photodiode, acquire the signal into our CDS system, and close a feedback loop.

  • The 44 MHz signal is itself supposed to be generated by the interference between the TEM00 55 MHz sideband from the IFO output with the TEM00 11 MHz sideband from the LO field (let's neglect any mode mismatch, HOMs etc for the moment).
  • By splitting this beat signal photocurrent in two, mixing each part with an electrical 44 MHz signal, and digitizing the IF output of said mixers, we should in principle be able to reconstruct the magnitude and phase of the signal.
  • The problem is that we know from other measurements that this signal is going to go through multiple fringes, and hence, we don't have a signal that is linear in the quantity we would like to control, namely the homodyne phase (either quadrature signal can be a candidate linear signal around a zero crossing, but when the signals are going through multiple fringes, neither signal stays linear).
  • One possible way to get around this problem is to use a phase tracker servo - basically, close a purely digital feedback loop, using one of the demodulated quadratures as an error signal, and changing the demodulation phase digitally such that the signal stays entirely in the orthogonal quadrature. However, such a scheme relies on the signal magnitude remaining constant. If the "error signal" goes to zero for multiple reasons (rotation out of the quadrature being considered, or just that the signal itself goes to zero), then this technique won't work. Of course, the phase tracker doesn't know what the "phase" of the signal is, when it's magnitude is (nearly) zero.
  • It is true that we always expect a "background" level of 44 MHz signal, from the 11 MHz and 55 MHz sidebands in the LO beam directly interfering, but this doesn't contain any useful information, and in fact, it'd only contaminate the phase tracker error signal I think.
  • So we can't rely on the error staying in one quadrature (like we do for the regular IFO PDH signals, where there is no relative phase propagation between the LO and RF sideband optical fields and so once we set the demodulation phase, we can assume the signal will always stay in that quadrature, and hence we can close a feedback loop), and we can't track the quadrature. What to do? I tried to dynamically change the phase tracker servo gain based on the signal magnitude (calculated in the RTCDS code using sqrt(I**2 + Q^2), but this did not yield good results...

Next steps:

I don't have any bright ideas at the moment - anyone has any suggestions?🤔

Aside:

I wanted to check what kind of signal the photodiode sees when only the LO field is incident on the photodiode. So with the IFO field blocked, I connected the PDA10CF to the Agilent analyzer in "Spectrum" mode, through a DC block. The result is shown in Attachment #2. To calculate the PM/AM ratio, I assumed a modulation depth of 0.2. The RIN was calculated by dividing the spectrum by the DC value of the PDA10CF output, which was ~1V DC. The frequencies are a little bit off from the true modulation frequencies because (i) I didn't sync the AG4395 to a Rb 10 MHz signal, and (ii) the span/BW ratio was set rather coarsely at 3kHz.

I would expect only 44 MHz and 66 MHz peaks, from the interference between the 11 MHz and 55 MHz sideband fields, all other field products are supposed to cancel out (or are in orthogonal quadratures). This is most definitely not what I see - is this level of RIN normal and consistent with past characterization? I've got no history in this particular measurement. 

Attachment 1: fieldQuads.pdf
fieldQuads.pdf
Attachment 2: PMAMratio.pdf
PMAMratio.pdf
  15596   Tue Sep 22 22:38:11 2020 gautamUpdateBHDSensing scheme for homodyne phase - some analytic calcs

I got some feedback from Koji who pointed out that the phase tracker is not required here. This situation is similar to the phase locking of two lasers together, which we frequently do, except in that case, we usually we offset the absolute frequencies of the two lasers by some RF frequency, and we demodulate the resulting RF beatnote to use as an error signal. We can usually acquire the lock by simply engaging an integrator (ignoring the fact that if we actuate on the laser PZT, which is a frequency actuator, just a proportional feedback will be sufficient because of the phase->frequency conversion), the idea being that the error signal is frequently going through a zero-crossing (around which the sinusoidal error signal is approximately linear) and we can just "catch" one of these zero crossings, provided we don't run of actuation range.

So the question here becomes, is the RF44 signal a suitable error signal such that we can close a feedback loop in a similar way? To try and get more insight, I tried to work out the situation analytically. I've attached my thinking as a PDF note. I get some pretty messy complicated expressions for the RF44 signal contributions, so it's likely I've made a mistake (though Mathematica did most of the heavy lifting), it'll benefit from a second set of eyes. 

Anyways, I definitely think there is some additional complications than my simple field cartoon from the preceeding elog would imply - the relative phases of the sidebands seem to have an effect, and I still think the lack of the PRC/SRC make the situation different from what Hang/Teng et. al. outlined for the A+ homodyne phase control analysis. Before the HEPA failed, I had tried closing the feedback loop using one quadrature of the demodulated RF44 signal, but had no success with even a simple integrator as the loop (which the experience with the PLL locking says should be sufficient, and pretty easily closed once we see a sinusoidally oscillating demodulated error signal). But maybe I'm overlooking something basic conceptually?

Quote:

eSummary:

I don't think the proposed scheme for sensing and controlling the homodyne phase will work without some re-thinking of the scheme. I'll try and explain my thinking here and someone can correct me if I've made a fatal flaw in the reasoning somewhere.

Attachment 1: simpleMich.pdf.zip
  15611   Mon Oct 5 00:37:19 2020 gautamUpdateBHDSingle bounce interferometer locked

Summary:

The simple interferometer, composed of a single bounce reflection from ITMY and the LO beam deilvered via fiber to the AS table, can be locked - i.e. the phase of the LO beam can be controlled such that the DC light level on the DCPDs after the two beams are interfered can be stabilized. This test allows us to confirm that various parts of the sensing and actuation chain (e.g. PI PZT for homodyne phase control, Trek amplifier etc etc) are working.

I will post more quantitative analysis tomorrow.

Optical configuration:

  • LO beam is a pickoff of the main PSL beam from just before it goes into the vacuum. The optical power arriving on each DCPD after the various beamsplitters, coupling loss etc is ~200 uW.
  • IFO beam is the single bounce reflection from ITMY. For this test, ETMY, ITMX and ETMY are misaligned. Optical power arriving on each DCPD is ~80uW.
  • The two beams are interfered on a 50-50 beamsplitter. The mode-matching efficiency was estimated to be ~50% which isn't stellar, but should be fine for this test.
  • So, at half-fringe, we expect the signal on each DCPD to be linearly proportional to the phase difference between the two fields, and so we can use that as an error signal.

Servo topology:

Attachment #2 shows the servo topology.

  • For a first attempt to close the feedback loop, we can consider the two blocks labelled "Sensing Chain" and "Actuation chain" to have a flat frequency response. While this isn't true, for a taget loop with ~100 Hz UGF, I think the approximation is reasonable.
  • From the peak-to-peak value (160 cts) of the DCPD signals when the homodyne phase is uncontrolled, I estimate a sensing response (at half-fringe) of approximately 0.3 ct/nm, since this corresponds to 532nm of relative phase between the two beams. 
  • An inverting summing amplifier is used to map the +/- 2^15 ct DAC range to 0-125V on the PI PZT. Assuming the full stroke of the PZT is 10um per the datasheet, and that this voltage range drives half of the full stroke (this is just a guess since all the old PI PZT circuits were designed to work at 0-250 V), we get an actuation coefficient of 0.075 nm/ct.
  • Using these two numbers, we can then design a digital feedback loop that gives an open loop transfer function with ~100 Hz UGF, and sufficient stability margin.
  • From the earlier measurements, we have an estimate for the amount of phase fluctuations caused by (i) seismic disturbances and (ii) fiber phase noise. This is the quantity we wish to suppress, and the suppression factor will be 1/(1+L), where L is the open loop gain.
  • I didn't do this in any systematic way, but the loop in Attachment #3 seemed like a reasonable shape that would suppress the error signal RMS by ~10x, as shown in Attachment #4. So I decided to try this out.

Other notes:

  1. The idea of offloading the DC control voltage to the ITMY suspension seemed to work fine.
  2. It also seems like the relative phase between the two beams doesn't drift by so large an amount in short time scales, at least at night/quiet seismic conditions. So it is possible to maintain the lock for several seconds without having to offload the DC signal to the suspensions.
  3. I didn't bother adapting the FSS Slow PID script to do this offloading in an automated way, seemed like more trouble than was just doing it by hand. But we may want to automate this in the future.
  4. I couldn't make a clean measurement of the loop transfer function using the usual IN1/IN2 method. Introducing a step offset at the error point, the servo is able to track it (I didn't fit the step response time, but it's not as if the loop bandwidth is <1 Hz or something). I have to compare the measured in-loop error signal ASD to the free-running one to get a feel for what the UGF is, I guess, to rule out a weird loop.
  5. Update 1100 Oct 6 2020: I have now added measured, in-loop, error point spectra to Attachment #4. Looks like there might be significant sensing noise re-injection.
    • Initially, I forgot to turn the HEPA on the PSL down for the measurement. So I have the two traces to compare. Looks like with the HEPA turned up to full, there is more noise in the 50-200 Hz range.
    • The trace marked "highGain" was taken with an overall loop gain that was 3dB higher than the nominal value - I could see some oscillations start to appear, and in the spectrum, maybe the feature at ~150 Hz is evidence of some gain peaking?

Conclusions:

  1. The PI PZT seems to work just fine.
  2. Need to look into the loop shape. I guess it's not reasonable to expect a UGF much higher than 100-200 Hz, because of the various delays in the system, but maybe the low frequency suppression can be made better.
  3. What are the next steps?? What does this mean for the RF44 sensing scheme?
Attachment 1: simpleHomodyne.png
simpleHomodyne.png
Attachment 2: singleBounceIFO.pdf
singleBounceIFO.pdf
Attachment 3: proposedController.pdf
proposedController.pdf
Attachment 4: freeRunningSuppressed.pdf
freeRunningSuppressed.pdf
  15612   Mon Oct 5 00:53:16 2020 KojiUpdateBHDSingle bounce interferometer locked

🤘🤘🤘

 

  15623   Tue Oct 13 11:13:54 2020 gautamUpdateBHDInvestigation into RF44 sensing

Attachment #1: spectra of the phase noise between LO and IFO output fields sensed using the RF44 signal.

  • Measurement setup:
    • LO an IFO fields are combined on a beamsplitter, with ~60% mode-matching efficiency.
    • One port of the BS goes to a DCPD.
    • The other port goes to an RF sensing photodiode, PDA10CF. The spec-ed dark noise NEP is ~12 pW/rtHz at 1.6 um, (so let's say 25 pW/rtHz) and transimpedance is 5kohms into a 50 ohm load. We can convert this to an equivalent sensing noise at the error point of this loop, though it's more likely that the electronics (demod, ADC etc) noise downstream dictate the sensing limit, which I measure by blocking light on the photodiode.
  • The demodulation is done on one of the newly received D0902745 boards - this was just a more compact setup than many cascaded minicircuit components. We don't have the hardware to package this into a chassis to shield against electronics noise pickup yet, so I'm using a bench supply to power this for now (via a voltage regulation board, D1000217.
  • "Dark Noise" = ASD with no light incident on the photodiode. "LO field only" = ASD with only the LO field incident on the photodiode.
  • The "Dark noise" trace and "LO field only" traces are converted from cts/rtHz to rad/rtHz by noting that when the Michelson is locked on a dark fringe, the demodulated RF44 quadratures have a pk-pk amplitude of ~160 cts (corresponding to pi radians of phase shift). Since in these conditions the demodulated quadratures do not undergo any fringe wrapping, I converted the spectra by simple multiplication.
  • For the "RF44 open loop" trace:
    • The DC offset in the demodulated signal (due to the RF44 signal from the LO field only) is digitally compensated, so that the fringing has (roughly) zero offset.
    • The Michelson was locked on a dark fringe, and the demodulated RF44 quadratures were monitored for ~5 mins. Then arctangent (specifically, arctan2 to get the correct quadrant in the IQ plane) of the two signals was taken to convert the fringing signals to phase noise.

Closing a feedback loop:

  • Since it seems like we are sensing a signal (below ~1kHz at least), I tried to close a feedback loop (modelled loop shape shown in Attachment #2, it's just a model because I have to guess what the sensing and actuation gains are, and they're both assumed to be flat, digital delays etc aren't accounted for). I've also added the inferred loop gain by taking the ratio of the in loop and unsuppressed ASDs (though of course I don't account for the flat sensing noise at higher frequencies). At least qualitatively, things line up...
  • While I can get the light level on the DCPD to stabilitze somewhat, the loop is not at all stable, and the suppression isn't very good at all.
  • Not sure how meaningful any of the spectra with the loop closed are, but FWIW, I've put in the spectra of the demodulated RF44 signals with the loop engaged (RF44 Q is used as the error signal). A clear problem is evident at ~120 Hz, and the forest of lines isn't helping for sure. Also unclear to me why the I and Q signals don't have the same profile at low frequencies.

Conclusions/Questions:

  1. What is the reason for the huge forests of lines in the "RF44 open loop" ASD, that are absent in the other two traces? If this were electrical pickup, it should be there in all three traces?
  2. Is the shape of the spectrum reasonable? The roll-off above ~5 Hz doesn't seem quite steep enough to be seismic noise from the suspensions. Can it really be that the Michelson dark field has such high phase noise?
  3. How can we get this scheme to give us cleaner sensing?
  4. The actuation chain was verified to work fine with the single bounce beam from an ITM interfered with the LO field, and using the DC light level as an error signal and locking to the half-fringe point. So the problem is not due to insufficient actuation range. Seems like the error signal is so polluted with these forests of lines that even though there is some suppression of the error signal at low frequencies, the unsuppressed noise is still significant. I can't solve the problem by simply increasing the loop gain...
  5. It is not shown here, but with only the LO field incident on the RFPD, I see a drift of the demodulated signals on the ~5 minute timescale - is this just due to fiber length change? If so, this is potentially problematic, as on long time scales, the true zero of the error point of the servo would be changing on the ~5 minute timescale. This would be true even for the final suspended scheme - if the path length between PR2 and the homodyne BS changes by some microns, we would have to correct this at DC?
Attachment 1: phaseNoisePSD.pdf
phaseNoisePSD.pdf
Attachment 2: loopTF.pdf
loopTF.pdf
  15628   Thu Oct 15 10:42:39 2020 gautamUpdateBHDMore investigation into RF44 sensing

Summary of discussion between Koji and gautam on 14 Oct:

  1. Koji questioned the accuracy of the "open loop" ASD shown here. While it may not be entirely accurate to compute the free-running (homodyne) phase noise simply by taking the arctangent of the I and Q signals (because the magnitude of the signal is also changing), gautam claims the estimate is probably still close to the true homodyne phase, especially since the ratio of the "in-loop" and free-running ASDs gives something that closely approximates the magnitude of the supposed OLG of the system.
  2. Koji suggested the following tests:
    • Investigate the relative stability of the two RF signal generators involved in this system. Since the 44 MHz electrical LO signal (for demodulation) is generated by a separate IFR from the one used to imprint 11 MHz and 55 MHz phase modulation sidebands on the main PSL beam, we want to investigate what the drift is.
    • Try implementing an analog feedback loop using LB1005 - the idea being we should be able to implement higher bandwidth control, for better suppression of the high frequency noise (which looking at the ASD is not only due to seismic phase modulation of the IFO output field). Maybe some combination of this and the Marconi investigation would suggest why we have these forests of lines in the ASDs of the error signal?
    • Turn off the HEPAs on the PSL enclosure completely as a test, to see if that improves (i) phase noise due to air currents and (ii) mechanical pickup on the fiber producing  phase noise.

I tried all of these last night / overnight, here are my findings.

Analog locking of the homodyne phase:

See Attachment #1

  • RF44_I was used as the error signal.
  • The "C1:OMC-ZETA_IMON_OUT" channel is actually looking at the error signal monitor from the LB1005, and is uncalibrated in this plot.
  • The "monitor" port on the demodulator board provides a convenient location for us to route the demodulated signal to an LB1005 box, while simultaneously digitizing both demodulated quadratures.
  • Empirically, I found settings that could engage the lock. I also found that I couldn't increase the gain much more without destroying the lock. 
  • The time domain signals look much "cleaner" in this analog feedback loop than when I achieved similar stabilization using the digital system. But I will quantify this more when I post some spectra of the in loop error signals.
  • I will do some more characterization (loop TF measurement, error point spectrum in lock etc), but in summary, it looks like we still only have ~100 Hz UGF. So something in the loop is limiting the bandwidth. What could it be?
  • The main problem is that the LB1005 isn't well suited to remote enabling/disabling of the lock, so this isn't such a great system.

Relative stability of two IFR2023s synchronized to the same FS725 Rb standard:

The electrical LO signal for demodulation of the 44 MHz photocurrent is provided by an IFR2023 signal generator. To maintain a fixed phase relation between this signal, and the phase modulation sidebands imprinted on the interferometer light via a separate IFO2023 signal generator, I synchronize both to the same Rb timing standard (a 10 MHz signal from the FS725 is sent to the rear panel frequency standard input on the IFR). We don't have a direct 44 MHz electrical signal available from the main IFO Marconi at the LSC rack (or anywhere else for that matter). So I decided to do this test at 55 MHz. 

  • RF input of the demodulator was driven by 5*11.066209 MHz pickoff from the LSC rack.
  • LO input of the demodulator was driven by 5*11.066209 MHz signal from the IFR2023 used for the RF44 demodulation setup.
  • The outputs were monitored overnight. The RF44_Q channel had a DC level of nearly 0. So this channel is nearly a linear sensor of the phase noise between LO and RF signals.
  • To convert ADC counts to radians, I offset the LO Marconi frequency by 100 Hz, and saw that the two quadratures showed pk-pk variation of ~24000cts. So, at the zero crossing, the conversion is 1/(24000/2) rad/ct ~83urad/ct.
  • The result is shown in Attachment #2. The "measurement noise" trace corresponds to the RF. input of the demodulator being terminated to ground with a 50 ohm terminator.
  • For comparison, I also overlay the phase noise estimate of an individual IFR from Rana. In his investigation, the claim is that the PLL that locks the IFR to the Rb timing standard has ~1kHz UGF, but if my measurement is correct, the relative stability between the two signal generators synchronized to the same timing standard already. degrades at ~1 Hz. Could be just a cts/rad calibration error I guess.
  • In any case, we are far from saturating this limit in the homodyne phase lock.
  • There are several sharp lines in this measurement too - but I don't know what exactly the source is. Of course the two marconis are plugged into separate power strips, so that may explain the 60 Hz lines and harmonics, but what about those that aren't a multiple of 60 Hz?

A look at the time domain signal:

With the Michelson locked on the dark fringe, the RF44 I and Q signals in the time domain are shown in Attachment #3 for a 1 minute stretch.

  • The RF44 signal level bottoms out at ~40 cts. Okay, so this is the offset.
  • However, the maximum value of the RF44 signal amplitude seems to be modulated in time. How can we explain this?
Attachment 1: analogZetaLock.png
analogZetaLock.png
Attachment 2: relPhaseNoise.pdf
relPhaseNoise.pdf
Attachment 3: sigMagPhase.pdf
sigMagPhase.pdf
  15631   Fri Oct 16 09:16:37 2020 YehonathanUpdateBHDMonte Carlo Simulations

Pushed another update to MCMC simulation. This includes:

  • Added new imbalances: ITM transmission, ITM & ETM RoCs.
  • Added new static offsets: DHARD, DSOFT, CHARD, CSOFT. All pitch. The RMS is calculated from the data Jon fetched with /input_noises/input_noises.ipynb.
  • SRCL noise ASD and RMS are now taken from data in /input_noises.
  • RF PD diagnostics were redone: Instead of post-discarding marginal simulations, simulations are now discarded when one or more of the RF PDs demodulated signal does not cross zero when the associated DOFs are scanned by 1um in the offset state.

The DOFs<->RFPD associations I use are:

DARM AS_f2_I
CARM REFL_f1_I
MICH POP_f2_Q
PRCL POP_f1_I
SRCL REFL_f2_I

However, one thing that bothers me is that for some reason ~ 15 out of 160 aLigo simulations are discarded while none for A+. It can also be seen that the A+ simulations are more spread-out which might be related.

The new simulation results are attached.

Attachment 1: MICH_AplusMCMC.pdf
MICH_AplusMCMC.pdf
Attachment 2: PRCL_AplusMCMC.pdf
PRCL_AplusMCMC.pdf
Attachment 3: SRCL_AplusMCMC.pdf
SRCL_AplusMCMC.pdf
Attachment 4: OMC_Comm_AplusMCMC.pdf
OMC_Comm_AplusMCMC.pdf
Attachment 5: OMC_Diff_AplusMCMC.pdf
OMC_Diff_AplusMCMC.pdf
Attachment 6: OMC_Angle_Yaw_AplusMCMC.pdf
OMC_Angle_Yaw_AplusMCMC.pdf
Attachment 7: OMC_Angle_Pitch_AplusMCMC.pdf
OMC_Angle_Pitch_AplusMCMC.pdf
Attachment 8: Main_Laser_RIN_AplusMCMC.pdf
Main_Laser_RIN_AplusMCMC.pdf
  15637   Thu Oct 22 11:48:08 2020 YehonathanUpdateBHDMonte Carlo Simulations

I found this H1 alog  entry by Izumi confirming that the calibrated channels CAL-CS_* need the same dewhitening filter.

This encouraged me to download the PRCL and MICH data and using Jon's example notebook. I incorporated these noise spectra into the MCMC simulation. The most recent results are attached.

I am still missing:

  • Laser frequency noise
  • Laser RIN
  • Estimation of the LO phase noise
  • Estimation of the BHD breadboard angular noise

Also, now the MCMC repeats a simulation if it doesn't pass the RF PDs test so the number of valid simulations stays the same. I'm still not sure about why the A+ simulations are much more robust to these tests than aLigo simulations.

Attachment 1: MICH_AplusMCMC.pdf
MICH_AplusMCMC.pdf
Attachment 2: PRCL_AplusMCMC.pdf
PRCL_AplusMCMC.pdf
Attachment 3: SRCL_AplusMCMC.pdf
SRCL_AplusMCMC.pdf
Attachment 4: OMC_Comm_AplusMCMC.pdf
OMC_Comm_AplusMCMC.pdf
Attachment 5: OMC_Diff_AplusMCMC.pdf
OMC_Diff_AplusMCMC.pdf
Attachment 6: OMC_Angle_Yaw_AplusMCMC.pdf
OMC_Angle_Yaw_AplusMCMC.pdf
Attachment 7: OMC_Angle_Pitch_AplusMCMC.pdf
OMC_Angle_Pitch_AplusMCMC.pdf
Attachment 8: Main_Laser_RIN_AplusMCMC.pdf
Main_Laser_RIN_AplusMCMC.pdf
  15684   Mon Nov 23 12:25:14 2020 gautamUpdateBHDBHD MMT Optics delivered

Optics --> Cabinet at south end (Attachment #1)

Scanned datasheets--> wiki. It would be good if someone can check the specs against what was ordered.

Attachment 1: IMG_8965.HEIC
  15685   Mon Nov 23 14:52:10 2020 KojiUpdateBHDBHD MMT Optics delivered

Basically, they repeated our specs and showed the coating performances for HR/AR for 10deg P and PR/AR for 45deg P. There is no RoC measurement by the vendor.
Nevertheless, their RoC (paper) specs should be compared with our request.

  15727   Thu Dec 10 14:48:00 2020 YehonathanUpdateBHDMonte Carlo Simulations

I have rebuilt the MCMC simulation in an OOP fashion and incorporated Lance/Pytickle functionality into it. The usage of the MCMC now is much less messy, hopefully.

I made an example that calculates the closed-loop noise-coupling from SRCL sensing and displacement to DARM in A+. I used the control filters that Kevin defined in his controls example.

The resulting noise budget is in attachment 1. The code is in the 40m/bhd git.

 

I also investigated why aLIGO simulations behave so different than the A+ simulation (See few previous elogs in this thread). That is why aLIGO results are much less variable, and the simulations in aLIGO barely pass the validity checks, while A+ simulations almost always pass.

The way I check for the validity of a kat model is by scanning all the DOFs and checking that the corresponding sensing RFPDs demodulated signals cross zero. Attachment 2 shows these scanning for 3 such RFPDS for 3 cases:

A+ model with maxtem = 2

ALigo model with maxtem = 2

ALigo model with maxtem = 'off'

It seems like the scanning curves for A+ and ALigo with no HOMs are well behaved and look like normal PDH signals, while the ALigo with maxtem = 2 curves look funky. I believe that the aLIGO+HOMS curves indicate that the IFO is not really in a good locking point. All the IFO lockings were done by using the locking methods straight out of the PyKat package. 

Attachment 1: MCMCLance_NoiseBudget_Example.pdf
MCMCLance_NoiseBudget_Example.pdf
Attachment 2: IFO_Check.pdf
IFO_Check.pdf
  15732   Fri Dec 11 09:28:52 2020 ranaUpdateBHDMonte Carlo loop coupling Simulations

Cool. Can you give us Bode plots of the open loop gain for each of the 5 length control loops?

Quote:

I have rebuilt the MCMC simulation in an OOP fashion and incorporated Lance/Pytickle functionality into it. The usage of the MCMC now is much less messy, hopefully.

 

  15734   Mon Dec 14 11:09:28 2020 YehonathanUpdateBHDMonte Carlo loop coupling Simulations

I spent a few hours monkeying around with the control filters. They are totally made up and also it's my first time trying to design control filters.

The OLTFs of the different length controls are shown in attachment 1.

The open-loop couplings of the DOFS to DARM are shown in attachment 2.

Note that in Lance/Pytickle the convention is that CLTFs are 1/(1 - G). Where G is the OLTF.

Quote:

Cool. Can you give us Bode plots of the open loop gain for each of the 5 length control loops?

 

 

Attachment 1: MCMC_LANCE_OLTFs.pdf
MCMC_LANCE_OLTFs.pdf
Attachment 2: MCMC_LANCE_OLCoupling2DARM.pdf
MCMC_LANCE_OLCoupling2DARM.pdf
  15758   Mon Jan 11 16:11:51 2021 YehonathanUpdateBHDMonte Carlo loop coupling Simulations

I dived into the alog to make the OLTFs in the MC_controls example more realistic. I was mainly inspired by these entries:

https://alog.ligo-la.caltech.edu/aLOG/uploads/47116_20190708131007_carmolg_20190702.png

https://alog.ligo-wa.caltech.edu/aLOG/index.php?callRep=18742

https://alog.ligo-wa.caltech.edu/aLOG/index.php?callRep=20466

and Evan's and Dennis's Theses.

Attachment 1 shows the new OLTFs. I tried to make them go like 1/f around the UGF and fall as quickly as possible at higher frequencies. I didn't do more advanced stability checks.

I also noticed that imbalances and detunings in the MC simulation can change the plants significantly. Especially DARM, CARM, and sometimes PRCL. I added the option to fix some OLTFs throughout the simulation. At every iteration, the simulation computes the required control filter to fix the selected OLTFs such that it will match the OLTFs in the undetuned and balanced IFO.

 

 

 

Attachment 1: MC_LANCE_OLTFs.pdf
MC_LANCE_OLTFs.pdf
  15759   Mon Jan 11 19:10:10 2021 ranaUpdateBHDMonte Carlo loop coupling Simulations
  • looking better, but the CARM plot still looks weird.
  • you should plot from 0.01 - 10,000 Hz
  • All of the loops should have true integrators below 1 Hz.
  • I don't think these loops are stable; the Bode plot is not a good way to check stability for LTI systems since you can be fooled by phase wrapping.
  15774   Wed Jan 20 18:07:09 2021 AnchalSummaryBHDHAM-A Coil Driver measurements before modifications

I have taken transfer functions and noise measurements of the two HAM-A coil driver boxes D1100687 #S2100027 and #S2100028. All transfer functions look as expected. I'm not sure about the noise measurements. If anyone sees flaw in my measurement method, please let me know. I'm not sure why in some channels I got 10Hz harmoni peaks in the noise. That was very strange. Also let me know if my current noise estimate is wrong.

Transfer Function Measurement details

  • SR785 source out was connected to the differential amplifier input of D1900068.
  • The one pair of two BNC outputs of this differential amplifier goes directly to the SR785 Input 1 A and B.
  • The DB9 output of the differential amplifier goes to the Coil Input DB9 connector J3.
  • Header W2 was shorted to provide ground to the incoming signal.
  • Header P4 was shorted to enable all the channels manually.
  • Normal operation is the Acquisition mode (Acq) while when pins of header P3 are shorted, we go into the Run mode for respective channel.
  • The “To Satellite Box” DB25 port at the read side was conencted to a DB25 breakout circuit and pins 1-9, 3-11, 5-13 and 7-15 were connected to 36 Ohm resistor to simulate Coil load.
  • The “Output Monitor” on the rear side is then connected to the test switch DB9 port on D1900068.
  • The the pair of BNCs from the test switch is connected to SR785 Input 2 A and B.
  • Measurements are taken with file D1100687_TF.yml and D1100687_TF_LF.yml.
  • A measurement of just cables without the DUT is taken as well.
  • Commands.txt list all the commands used.
  • All data is compiled and plotted in Plotting.ipynb
  • D1100117_S2100027_TF.pdf and D1100117_S2100028_TF.pdf shows all the transfer functions measured.

Spectrum Measurements

  • All channels were kept in disabled mode (Not shorting P4) to ensure their inputs are grounded on the board.
  • I ran two BNC cables with their centers connected to output monitors V2+ and V2- and one of their shields connected to board GND.
  • in SR785, A-B differential mode always runs with grounded shields mode, so effectively the board GND got grounded to SR785 GND through internal 50 Ohm resistor. But all ground loops have been evaded.
  • The two BNC cables were twisted together to minimize the area between the two center cores of the cables as that is the remaining pickoff possible in this measurement.
  • Instrument noise with cables was measured first but shorting the clips of the center cores and one of the shields of the two BNC cables together.
  • Measurements were taken with file D1100687_SP.yml and D1100687_SP_LF.yml.
  • D1100117_S2100027_Voltage_Noise_Spectrum.pdf and D1100117_S2100028_Voltage_Noise_Spectrum.pdf shows the measured voltage noise spectrum at the output monitors when loaded with 36 Ohms.
  • D1100117_S2100027_Current_Noise_Spectrum.pdf and D1100117_S2100028_Current_Noise_Spectrum.pdf shows the esitmate current noise through the coil calculated by dividing the measured voltage noise by 2436 Ohms.
Attachment 1: MeasurementData.zip
Attachment 2: D1100117_S2100027_TF.pdf
D1100117_S2100027_TF.pdf D1100117_S2100027_TF.pdf D1100117_S2100027_TF.pdf D1100117_S2100027_TF.pdf D1100117_S2100027_TF.pdf
Attachment 3: D1100117_S2100028_TF.pdf
D1100117_S2100028_TF.pdf D1100117_S2100028_TF.pdf D1100117_S2100028_TF.pdf D1100117_S2100028_TF.pdf D1100117_S2100028_TF.pdf
Attachment 4: D1100117_S2100027_Voltage_Noise_Spectrum.pdf
D1100117_S2100027_Voltage_Noise_Spectrum.pdf D1100117_S2100027_Voltage_Noise_Spectrum.pdf D1100117_S2100027_Voltage_Noise_Spectrum.pdf D1100117_S2100027_Voltage_Noise_Spectrum.pdf D1100117_S2100027_Voltage_Noise_Spectrum.pdf
Attachment 5: D1100117_S2100028_Voltage_Noise_Spectrum.pdf
D1100117_S2100028_Voltage_Noise_Spectrum.pdf D1100117_S2100028_Voltage_Noise_Spectrum.pdf D1100117_S2100028_Voltage_Noise_Spectrum.pdf D1100117_S2100028_Voltage_Noise_Spectrum.pdf D1100117_S2100028_Voltage_Noise_Spectrum.pdf
Attachment 6: D1100117_S2100027_Current_Noise_Spectrum.pdf
D1100117_S2100027_Current_Noise_Spectrum.pdf D1100117_S2100027_Current_Noise_Spectrum.pdf D1100117_S2100027_Current_Noise_Spectrum.pdf D1100117_S2100027_Current_Noise_Spectrum.pdf D1100117_S2100027_Current_Noise_Spectrum.pdf
Attachment 7: D1100117_S2100028_Current_Noise_Spectrum.pdf
D1100117_S2100028_Current_Noise_Spectrum.pdf D1100117_S2100028_Current_Noise_Spectrum.pdf D1100117_S2100028_Current_Noise_Spectrum.pdf D1100117_S2100028_Current_Noise_Spectrum.pdf D1100117_S2100028_Current_Noise_Spectrum.pdf
  15776   Mon Jan 25 18:18:04 2021 AnchalSummaryBHDSatellite Amplifier Transfer Functions and noise

 

I took transfer function and noise measurement of satellite amplifier box's photodiode transimpedance circuit. For the measurement, I created a makeshift connector to convert backside DB25 into DB9 with the 4 channels for PDA input. The output was taken in differential form at the front PD Output port. To feed current to the circuit, I put in 12 kOhm resistors in series at the inputs, so the V/V transfer function measured was multiplied by 12 kOhm to get the transimpedance of the circuit.


Transfer Function Measurement details

  • SR785 source out was fed into PDA input pins using a makeshift BNC-DB9-DB25 converter.
  • The output from PDOut DB9 port was fed to test switch in D1900068 to separate differential signal.
  • This differential signal was fed back to SR785 at input 2 in A-B configuration.
  • Measurements are taken with file D1002818_TF.yml and D1002818_TF_LF.yml.
  • A measurement of just cables without the DUT is taken as well.
  • Commands.txt list all the commands used.
  • All data is compiled and plotted in Plotting.ipynb
  • D1100117_S2100029_TFandNoiseSpectrum.pdf shows all the transfer functions measured.

Spectrum Measurements

  • Two pair of BNC cables were twisted together and clips were added at ends.
  • One of the GND was connected to board GND. Rest were left unconnected to avoid ground loops.
  • Each pair of signal was connected to PDOutP/N.
  • The PDA inputs were shorted together to make zero input current to the board.
  • Instrument noise with cables was measured by shorting the clips of the center cores and one of the shields of the two BNC cables together.
  • Measurements were taken with file D1002818_SP.yml and D1002818_SP_LF.yml.
  • Input referred current noise spectrum was calculated by dividing the output voltage noise spectrum by the measured transfer function.
  • D1100117_S2100029_TFandNoiseSpectrum.pdf shows all the output votlage noise spectrum and input referred current noise spectrum measured.

Edit Wed Feb 10 15:14:13 2021 :

THE NOISE MEASUREMENT WAS WRONG HERE. SEE 40m/15799.

Attachment 1: D1002818_S2100029_TFandNoiseSpectrum.pdf
D1002818_S2100029_TFandNoiseSpectrum.pdf D1002818_S2100029_TFandNoiseSpectrum.pdf D1002818_S2100029_TFandNoiseSpectrum.pdf
Attachment 2: D1002818_Testing.zip
  15780   Thu Jan 28 12:53:14 2021 AnchalSummaryBHDHAM-A Coil Driver measurements before modifications

I took some steps to reduce the coupling of 60 Hz harmonics in noise measurement. The box was transferred to the floor instead of on top of another instrument. Measurement was immediately converted into single-ended using SR560 in battery mode with a gain of 10. All of the setups was covered in aluminum foil to increase isolation.

Spectrum measurement details

 

Attachment 1: D1100117_S2100027_Current_Noise_Spectrum.pdf
D1100117_S2100027_Current_Noise_Spectrum.pdf D1100117_S2100027_Current_Noise_Spectrum.pdf D1100117_S2100027_Current_Noise_Spectrum.pdf D1100117_S2100027_Current_Noise_Spectrum.pdf D1100117_S2100027_Current_Noise_Spectrum.pdf
Attachment 2: D1100117_S2100027_Voltage_Noise_Spectrum.pdf
D1100117_S2100027_Voltage_Noise_Spectrum.pdf D1100117_S2100027_Voltage_Noise_Spectrum.pdf D1100117_S2100027_Voltage_Noise_Spectrum.pdf D1100117_S2100027_Voltage_Noise_Spectrum.pdf D1100117_S2100027_Voltage_Noise_Spectrum.pdf
Attachment 3: D1100117_S2100028_Current_Noise_Spectrum.pdf
D1100117_S2100028_Current_Noise_Spectrum.pdf D1100117_S2100028_Current_Noise_Spectrum.pdf D1100117_S2100028_Current_Noise_Spectrum.pdf D1100117_S2100028_Current_Noise_Spectrum.pdf D1100117_S2100028_Current_Noise_Spectrum.pdf
Attachment 4: D1100117_S2100028_Voltage_Noise_Spectrum.pdf
D1100117_S2100028_Voltage_Noise_Spectrum.pdf D1100117_S2100028_Voltage_Noise_Spectrum.pdf D1100117_S2100028_Voltage_Noise_Spectrum.pdf D1100117_S2100028_Voltage_Noise_Spectrum.pdf D1100117_S2100028_Voltage_Noise_Spectrum.pdf
Attachment 5: SpectrumMeasurement.zip
  15781   Thu Jan 28 18:04:55 2021 AnchalSummaryBHDHAM-A Coil Driver measurements After modifications

I did the recommended modifications on of the boards with serial number S2100028. These included:

  • R13, R27: 160 -> 75
  • C11, C21: 470 nF -> 68nF
  • C19: 4.7 uF -> 470 nF
  • R15: 3.23 kOhm -> 1.82 kOhm

I took transfer function measurements with same method as in 40m/15774 and I'm presenting it here to ensure the modifications are correct and if I should proceed to the next board as well. I didn't have the data used to make plots in here but I think the poles and zeros have landed in the right spot. I'll wait for comments until tomorrow to proceed with changes in the other board as well. I'll do noise measurements tomorrow.

Attachment 1: D1100117_S2100027_TF.pdf
D1100117_S2100027_TF.pdf D1100117_S2100027_TF.pdf D1100117_S2100027_TF.pdf D1100117_S2100027_TF.pdf D1100117_S2100027_TF.pdf
Attachment 2: AfterChanges.zip
  15782   Thu Jan 28 21:44:45 2021 gautamSummaryBHDHAM-A Coil Driver measurements After modifications

Looks fine to me visually but the verdict can only be made once the z:p locations are quantitatively confirmed, and the noise tests pass. It would be interesting to see what kind of time-domain transient (in N of force) switching on the de-whitening introduces, i guess best done interferometrically.

Quote:

I'll wait for comments until tomorrow to proceed with changes in the other board as well. I'll do noise measurements tomorrow.

  15784   Fri Jan 29 15:39:30 2021 AnchalSummaryBHDHAM-A Coil Driver measurements After modifications TF and Noise S2100027

I fitted zeros and poles in the measured transfer function of D1100687 S2100027 and got zeros at 130 Hz and 234 Hz and poles at 10Hz and 2845 Hz. These values are different from the aimed values in this doc, particularly the 234Hz zero which was aimed at 530 Hz in the doc.

I also took the noise measurement using the same method as described in 40m/15780. The noise in Acquisition mode seems to have gone up in 10 Hz - 500 Hz region compared to the measurement in 40m/15780 before the modifications.

All channels are consistent with each other.


Edit Mon Feb 1 12:24:14 2021:
Added zero model prediction after the changes. The measurements match with the predictions.


Edit Wed Feb 3 16:46:59 2021:

Added zero modeled noise in the noise spectrum curves. The acquisition mode curves are in agreement with the model. The noise in Run mode is weirdly lower than predicted by zero.

Attachment 1: D1100687_S2100027_After_Modifications_Jan28.jpg
D1100687_S2100027_After_Modifications_Jan28.jpg
Attachment 2: D1100117_S2100027_TF.pdf
D1100117_S2100027_TF.pdf D1100117_S2100027_TF.pdf D1100117_S2100027_TF.pdf D1100117_S2100027_TF.pdf D1100117_S2100027_TF.pdf
Attachment 3: D1100117_S2100027_Voltage_Noise_Spectrum.pdf
D1100117_S2100027_Voltage_Noise_Spectrum.pdf D1100117_S2100027_Voltage_Noise_Spectrum.pdf D1100117_S2100027_Voltage_Noise_Spectrum.pdf D1100117_S2100027_Voltage_Noise_Spectrum.pdf D1100117_S2100027_Voltage_Noise_Spectrum.pdf
Attachment 4: D1100117_S2100027_Current_Noise_Spectrum.pdf
D1100117_S2100027_Current_Noise_Spectrum.pdf D1100117_S2100027_Current_Noise_Spectrum.pdf D1100117_S2100027_Current_Noise_Spectrum.pdf D1100117_S2100027_Current_Noise_Spectrum.pdf D1100117_S2100027_Current_Noise_Spectrum.pdf
Attachment 5: AfterChanges.zip
  15787   Tue Feb 2 11:57:46 2021 AnchalSummaryBHDHAM-A Coil Driver measurements After modifications TF and Noise S2100028

I have made the modifications on the other board D1100687 S2100028 as well. The measurements were taken as mentioned in 40m/15784. All conclusions remain the same as 40m/15784. The attached zip file contains all measurement data, before and after the modifications.


Edit Wed Feb 3 16:44:51 2021 :

Added zero modeled noise in the noise spectrum curves. The acquisition mode curves are in agreement with the model. The noise in Run mode is weirdly lower than predicted by zero.

Attachment 1: D1100687_S2100028_After_Modifications_Feb01_2021.jpg
D1100687_S2100028_After_Modifications_Feb01_2021.jpg
Attachment 2: D1100117_S2100028_TF.pdf
D1100117_S2100028_TF.pdf D1100117_S2100028_TF.pdf D1100117_S2100028_TF.pdf D1100117_S2100028_TF.pdf D1100117_S2100028_TF.pdf
Attachment 3: D1100117_S2100028_Voltage_Noise_Spectrum.pdf
D1100117_S2100028_Voltage_Noise_Spectrum.pdf D1100117_S2100028_Voltage_Noise_Spectrum.pdf D1100117_S2100028_Voltage_Noise_Spectrum.pdf D1100117_S2100028_Voltage_Noise_Spectrum.pdf D1100117_S2100028_Voltage_Noise_Spectrum.pdf
Attachment 4: D1100117_S2100028_Current_Noise_Spectrum.pdf
D1100117_S2100028_Current_Noise_Spectrum.pdf D1100117_S2100028_Current_Noise_Spectrum.pdf D1100117_S2100028_Current_Noise_Spectrum.pdf D1100117_S2100028_Current_Noise_Spectrum.pdf D1100117_S2100028_Current_Noise_Spectrum.pdf
Attachment 5: AfterChanges.zip
  15788   Tue Feb 2 17:09:17 2021 yehonathanUpdateBHDSOS assembly

I set up a working area on the table next to the south flow bench (see attachment). I also brought in a rolling table for some extra space.

I covered all the working surfaces with a foil from the big roll between 1x3 and 1x4.

I took the SOSs, SOS parts and the OSEMS from the MC2 table to the working area.

I cleaned some LN Allen keys with isopropanol and put them on the working table, please don't take them.

Attachment 1: 20210202_165501.jpg
20210202_165501.jpg
Attachment 2: 20210202_162452.jpg
20210202_162452.jpg
  15790   Tue Feb 2 18:24:54 2021 KojiUpdateBHDSOS assembly

You can remove the components of the optical table enclosure (black ones) and use the optical table as your working area too.

 

  15793   Wed Feb 3 16:27:19 2021 AnchalSummaryBHDSatellite Amplifier Transfer Functions and noise After modifications

I have made modifications recommended in this doc. The changes made are:

  • R24: 19.6k to 4.99k Ohms
  • R20: 19.6k to 4.99k Ohms
  • R23: 787 to 499 Ohms
  • Removed C16.

I took transfer function measurements, fitted them with zeros and poles and plotted it against the zero model of the circuit. The zeros and poles we intended to shift are matching well with 3Hz zero and 30 Hz pole. The later pole at 1500 Hz is at a higher value from what is predicted by zero.

I also took noise measurements and they are in good agreement with the noise predicted by zero.


Edit Wed Feb 10 15:14:13 2021 :

THE NOISE MEASUREMENT WAS WRONG HERE. SEE 40m/15799.

Attachment 1: D1002818_S2100029_TFAfterChanges.pdf
D1002818_S2100029_TFAfterChanges.pdf D1002818_S2100029_TFAfterChanges.pdf D1002818_S2100029_TFAfterChanges.pdf D1002818_S2100029_TFAfterChanges.pdf D1002818_S2100029_TFAfterChanges.pdf D1002818_S2100029_TFAfterChanges.pdf D1002818_S2100029_TFAfterChanges.pdf D1002818_S2100029_TFAfterChanges.pdf
Attachment 2: D1002818_S2100029_OutputNoiseSpecAfterChanges.pdf
D1002818_S2100029_OutputNoiseSpecAfterChanges.pdf D1002818_S2100029_OutputNoiseSpecAfterChanges.pdf D1002818_S2100029_OutputNoiseSpecAfterChanges.pdf D1002818_S2100029_OutputNoiseSpecAfterChanges.pdf D1002818_S2100029_OutputNoiseSpecAfterChanges.pdf D1002818_S2100029_OutputNoiseSpecAfterChanges.pdf D1002818_S2100029_OutputNoiseSpecAfterChanges.pdf D1002818_S2100029_OutputNoiseSpecAfterChanges.pdf
Attachment 3: D1002818_S2100029_InputRefferedNoiseSpecAfterChanges.pdf
D1002818_S2100029_InputRefferedNoiseSpecAfterChanges.pdf D1002818_S2100029_InputRefferedNoiseSpecAfterChanges.pdf D1002818_S2100029_InputRefferedNoiseSpecAfterChanges.pdf D1002818_S2100029_InputRefferedNoiseSpecAfterChanges.pdf D1002818_S2100029_InputRefferedNoiseSpecAfterChanges.pdf D1002818_S2100029_InputRefferedNoiseSpecAfterChanges.pdf D1002818_S2100029_InputRefferedNoiseSpecAfterChanges.pdf D1002818_S2100029_InputRefferedNoiseSpecAfterChanges.pdf
Attachment 4: D1002812_S2100029_After_Modifications_Feb3.jpg
D1002812_S2100029_After_Modifications_Feb3.jpg
Attachment 5: AfterChanges.zip
  15796   Thu Feb 4 15:14:55 2021 YehonathanUpdateBHDSOS assembly

I gathered all the components I could find from the SOS towers and the cleanroom and put it all on the table next to the flow bench (See attachment).

I combed through the cleanroom cabinet for SOS parts but didn't find all the parts listed in the procurement spreadsheet. I did find some extra items that were not listed.

This table compares the quantities in the spreadsheet to the quantities collected on the table. Green rows are items I found more than in the procurement spreedsheet while red rows are items I found less.

ITEM DCC # Qty required Qty in procurement spreadsheet How much I found
SENSOR/ACTUATOR PLATE D960002 14 21 21
SUSPENSION BLOCK D960003 7 9 9
TOWER BASE D960004 7 10 11
RIGHT SIDE PLATE D960005 7 12 13
LEFT SIDE PLATE D960006 7 12 13
UPPER MIRROR CLAMP D960007 7 8
7+1 teflon
LOWER CLAMP D960008-1 7 8 8
LOWER CLAMP, OPPOSITE D960008-2 7 8 8
WIRE CLAMP 1205308-1 10 17 9
CLAMP, SUSPENSION BLOCK D960134 14 19 21
STIFFENER PLATE D960009 7 9 9
DUMBBELL STANDOFF D970075 50 10 7
SAFETY STOP, LONG D970313 14 2 10
OSEM assy D960011 35 2 13 wire wound osem housings (gold)
WIRE STANDOFF D970187 20 7 0
GUIDE ROD D970188 10 9 0
MAGNET D960501 50 54 51 rusted + 37 slightly rusted. Didn't put on table
SAFETY STOP, SMALL D970312 28 0 4
SAFETY STOP D970311 28 0
16+9 stained w/o spring
SS Spring Plunger 8498A999 35 4 27
Attachment 1: 20210204_144007.jpg
20210204_144007.jpg
  15797   Wed Feb 10 11:45:59 2021 AnchalSummaryBHDSatellite Amplifier Very Low frequency noise After modifications

As suggested, I wrapped the satellite amplifier box D10028128 S2100029 in blanket and foam and took very low frequency spectrum starting from 32 mHz to 3 Hz. The results are attached along with stiched high frequency measurements from 40m/15793.

Very Low Frequency Spectrum Measurement

  • D1002818 S2100029 box was powered and covered in a foam blanket.
  • Additionally, it was covered from all sides with foam to reduce wind and temperature effects on it.
  • The rear panel DB25 connector was connected to a breakout board where pins od PDA input and GND were shorted, shorting the transimpedance circuit input.
  • The output was read from PDMon DB9 output at front panel which was converted to 4 BNC channels using breakout board.
  • Two channel noise was measured at once using D1002818_SP.yml parameter file.
  • Instrument noise at all the used input ranges were measured separately by shorting the input of the BNC cables.

Edit Wed Feb 10 15:14:13 2021 :

THIS MEASUREMENT WAS WRONG. SEE 40m/15799.

Attachment 1: FrontsideLook.jpg
FrontsideLook.jpg
Attachment 2: BacksideLook.jpg
BacksideLook.jpg
Attachment 3: InnerFoamBlanket.jpg
InnerFoamBlanket.jpg
Attachment 4: D1002818_S2100029_OutputNoiseLFSpecAfterChanges.pdf
D1002818_S2100029_OutputNoiseLFSpecAfterChanges.pdf D1002818_S2100029_OutputNoiseLFSpecAfterChanges.pdf D1002818_S2100029_OutputNoiseLFSpecAfterChanges.pdf D1002818_S2100029_OutputNoiseLFSpecAfterChanges.pdf D1002818_S2100029_OutputNoiseLFSpecAfterChanges.pdf D1002818_S2100029_OutputNoiseLFSpecAfterChanges.pdf D1002818_S2100029_OutputNoiseLFSpecAfterChanges.pdf D1002818_S2100029_OutputNoiseLFSpecAfterChanges.pdf
Attachment 5: D1002818_S2100029_InputRefCurrentNoiseLFSpecAfterChanges.pdf
D1002818_S2100029_InputRefCurrentNoiseLFSpecAfterChanges.pdf D1002818_S2100029_InputRefCurrentNoiseLFSpecAfterChanges.pdf D1002818_S2100029_InputRefCurrentNoiseLFSpecAfterChanges.pdf D1002818_S2100029_InputRefCurrentNoiseLFSpecAfterChanges.pdf D1002818_S2100029_InputRefCurrentNoiseLFSpecAfterChanges.pdf D1002818_S2100029_InputRefCurrentNoiseLFSpecAfterChanges.pdf D1002818_S2100029_InputRefCurrentNoiseLFSpecAfterChanges.pdf D1002818_S2100029_InputRefCurrentNoiseLFSpecAfterChanges.pdf
Attachment 6: AfterChangesLFSpectrum.zip
  15799   Wed Feb 10 15:07:50 2021 AnchalSummaryBHDSatellite Amplifier Output Offset measurements

I measured the output DC voltage of the satellite amplifier box at PDMon port when the PDA input was shorted and got following offsets:
 

CH Output Offset (mV) CH Output Offset (mV)
1 6 5 750
2 140 6 120
3 350 7 537
4 40 8 670

However, I think I'm making a mistake while measuring this offset as well as all the noise measurements of this satellite amplifier box so far. Since it is a current input, transimpedance circuit, the noise of the circuit should be measured with open input, not closed. Infact, by shorting the PDA input, I'm giving DC path to input bias current of AD833 transimpedance amplifier to create this huge DC offset. This won't be the case when a photodiode is connected at the input which is a capacitor and hence no DC path is allowed. So my issue of offset was bogus and past two noise measurements in 40m/15797 and 40m/15793 are wrong.

  15800   Wed Feb 10 15:25:45 2021 gautamSummaryBHDSatellite Amplifier Output Offset measurements

Why not just do this test with the dummy suspension box and CDS system? I think Rich's claim was that the intrinsic LED RIN was dominant over any drive current noise but we can at least measure the quadrature sum of the two (which is after all the relevant quantity in terms of what performance we can realize) and compare to a model.

ELOG V3.1.3-