Amplitude of 29.485MHz input sine wave [dBm] | Value of channel C1:IOO-MC_DEMOD_LO
-------------------------------------------- | -----------------------------------
-10 | -0.000449867
-8 | -0.000449867
-6 | -0.000449867
-4 | 0.000384331
-2 | 0.00526733
0 | 0.0199163
2 | 0.0492143
4 | 0.0931613
6 | 0.161523
8 | 0.229885
10 | 0.293364
- Use LEDs of a wavelength that the OSEMs don't see. LEDs are also cool so that the
Suspension won't drift in alignment.
- Use 2 power supplies so that the power is balanced.
- Make is +/-12 V twisted AWG 14 wire so that the EMI is contained. Should also
be shielded cable.
The attached plot shows that someone broke the MC_SUM_MON channel around 10:30 AM this past Wednesday the 11th. This is the EPICS monitor of the MC error point.
Come forward now with your confession and I promise that I won't let Steve hurt you.
I checked a broken QPD, which was placed for PSL angle monitor, and finally I cocluded one segment of the quadrant diode was broken.
The broken segment has a offset voltage of -0.7V after 1st I-V amplifier. It means the diode segment has a current offset without any injection of light.
Tomorrow I will check a new QPD for replacement.
Today I checked out the SR560 around the lab. I confirmed that the one with the label "channel A noisy" is effectively mulfuctioning.
It was coonected to the lock-in amplifier set up for the drum mode peak readout.
I repleaced that with a working one.
I borrowed SR785 to measure AA, AI noise and TF.
Eric Gustafson is handling the old HP4291A rehabilitation. Tarac picked both units up today.
March of 2008 Tucker Electronics failed to fix it's intermittent ~25MHz 0.5V oscillation at the swept sine output
See 40m-elog id:398 on 3-24-2008 by Rob Ward
Pete, Rob, Alberto,
yesterday we thought that some of the problems we were having in locking the IFO might be related to a change of the length of the mode cleaner. So today we decided to measure it again.
We followed the Sigg-Frolov technique (see 40m Wiki, Waldman, Fricke). For the record, the MC_AO input corresponds to IN2 on the MC Servo board.
We obtained: L = 27.092 +/- 0.001 m
From the new measurement we reset the frequencies of the Marconis to the following values:
For the 40m Upgrade, we plan to eliminate the Mach-Zehnder and replace it with a single EOM driven by all three modulation frequencies that we'll need: f1=11MHz, f2=5*f1=55MHz, fmc=29.5MHz.
A frequency generator will produce the three frequencies and with some other electronics we'll properly combine and feed them to the EOM.
The frequency generator will have two crystals to produce the f1 and fmc signals. The f2 modulation will be obtained by a frequency multiplier (5x) from the f1.
The frequency multiplier, for the way it works, will inevitably introduce some unwanted harmonics into the signals. These will show up as extra modulation frequencies in the EOM.
In order to quantify the effects of such unwanted harmonics on the interferometer and thus to let us set some limits on their amplitude, I ran some simulations with Optickle. The way the EOM is represented is by three RF modulators in series. In order to introduce the unwanted harmonics, I just added an RF modulator in series for each of them. I also made sure not to leave any space in between the modulators, so not to introduce phase shifts.
To check the effect at DC I looked at the sensing matrix and at the error signals. I considered the 3f error signals that we plan to use for the short DOFs and looked at how they depend on the CARM offset. I repeated the simulations for several possible amplitude of the unwanted harmonics. Some results are shown in the plots attached to this entry. 'ga' is the amplitude ratio of the unwanted harmonics relative to the amplitude of the 11 & 55 MHz modulations.
Comparing to the case where there are no unwanted harmonics (ga = 0), one can see that not considerable effect on the error signals for amplitudes 40dB smaller than that of the main sidebands. Above that value, the REFL31I signals, that we're going to use to control PRCL, will start to be distorted: gain and linearity range change.
So 40 dB of attenuation in the unwanted harmonics is probably the minimum requirement on the frequency multiplier, although 60dB would provide a safer margin.
I'm still thinking how to evaluate any AC effect on the IFO.
** TODO: Plot DC sweeps with a wider range (+/- 20 pm). Also plot swept sines to look for changes in TFs out to ~10 kHz.
The beam scan (which has been living in the bridge subbasement for a bit now) is in a state of imperfection.
I noticed that:
I took some pictures to try and illuminate the situation - The inverted images are included to make it easier to see the flecks (?) in the slits
I am not sure how to figure out if any bit of the scan is/has been fried.
Pending further investigation, enjoy large error bars in your scan measurements!
PICTURES OF BOTH SLITS ON THE BEAMSCAN HEAD:
Koji and I were looking for an extender card to aid with MZ board testing. Rob went off on a quest to find one. He found 2 (in addition to the one in the drawer near the electronics bench which says "15V shorted"), and put them in some empty slots in 1X1 to test them out. Somehow, this burned a few pins on each board (1 pin on one of them, and 3 pins on the other). We now have 0 functioning extender cards: unfortunately, both extender cards now need fixing. The 2 slots that were used in 1X1 now have yellow electrical tape covering the connectors so that they do not get used, because the ends of the burnt-off pins may still be in there.
In other, not-Rob's-fault news, the Martian network is down...we're going to try to reset it so that we have use of the laptops again.
This happened when I plugged the cards into a crate with computers, which apparently is a no-no. The extender cards only go in VME crates full of in-house, LIGO-designed electronics.
I took the pictures of all racks of electronics yesterday, and then uploaded these pictures on the wiki.
You can see them by clicking "pictures" in the wiki page.
Its back in and re-centered. Our next move on IPPOS should be to replace its steering mirror with something bigger and more rigid.
20K -> 3.65 K (R6, R20, R42, R31) (unused)
20K -> 3.65 K (R7, R21, R32, R43, R11, R24, R35, R46)
If you look in the schematic (D990272), you see that its an AD797 transimpedance stage with a couple of LT1125 stages set to give some switchable gain. It looks like some of these
switches are on and some are not, but I am not sure where it would be controlled from. I've attached a snapshot of one quadrant of the schematic below.
The schematic shows the switches in the so-called 'normally closed' configuration. This is what the switches do with zero volts applied to the control inputs. As the schematic also shows,
just disconnecting the 'switch' inputs cause the switch's control inputs to go high (normally open configuration, i.e. pins 2-3 connected, pin4 open). For the record, the default positions of the IPPOS switches are:
** EDIT (Nov 2, 2009): I forgot to attach the before and after images; here they are:
I tried to compare the IP_POS time series with the IPANG and MC_TRANS but was foiled at first:
1) The IPANG scan rate was set to 0.5 second, so it doesn't resolve the pendulum motions well. Fixed in the .db file.
2) Someone had used a Windows/DOS editor to edit the .db file and it was filled with "^M" characters. I have removed them all using this command: tr -d "\r" <ETMXaux.db > new.db
3) The MC_TRANS P/Y channels were on the MC Lock screen but had never been added to the DAQ. Remember, if there's a useful readback on an EPICS screen. its not necessarily in the frames unless you add it to the C0EDCU file. I have done that now and restarted the fb daqd. Channels now exist.
4) Changed the PREC of the IPPOS channels to 3 from 2.
5) changed the sign for the IBQPD (aka IPANG) so that bigger signal is positive on the EPICS screen.
In ----o-------- | | --------o-------- Out
_ 1uF R 7.5 kOhms
I have been working about multi-resonant EOM since last week.
In order to characterize the behavior of the each components, I have made a simple LC tank circuit.
You can see the picture of the circuit below.
Before constructing the circuit, I made an "ideal" calculation of the transfer function without any assumptions by my hand and pen.
Most difficult part in the calculation is the dealing with a transformer analytically. Eventually I found how to deal with it in the analytical calculation.
The comparison of the calculated and measured transfer function is attached.
It shows the resonant frequency of ~50MHz as I expected. Those are nicely matched below 50MHz !!
For the next step, I will make the model of the circuit with stray capacitors, lead inductors, ... by changing the inductance or something.
I have measured the impedance of the LC tank circuit which I referred on my last entry.
The configuration of the circuit is exactly the same as that time.
In order to observe the impedance, by using Koji's technique I injected a RF signal into the output of the resonant circuit.
In addition I left the input opened, therefore the measured impedance does not include the effect of the transformer.
- - - - - - - - - - - - results
The measured impedance is attached below; "LCtank_impedance.png"
The peak around 50MHz is the main resonance and it has impedance of ~1500 [Ohm], which should go to infinity in the ideal case (no losses).
In fact the impedance looked from the input of the circuit gets reduced by 1/n^2, where "n" is the turn ratio of the transformer.
By putting the n=4, the input impedance of the circuit should be 93 [Ohm]. This is a moderate value we can easily perform impedance-matching by some technique.
I also fitted the data with a standard model of equivalent circuit (see attachment 2).
In the figure.2 red component and red letter represents the design. All the other black stuff are parasites.
But right now I have no idea the fitted value is reasonable or not.
For the next I should check the input impedance again by the direct way; putting the signal into the input.
I measured the input impedance of the LC tank circuit with the transformer. The result is attached.
Note that the input impedance can be derived as follower;
(input impedance) = L1 + Z/n^2.
Where L1 is the inductance of the primary coil, Z is the load in the secondary loop and n is the turn ratio.
I think now I am getting ready to enter the next phase \(^o^)/
It looked like the Busby Low Noise Box had too much low frequency noise and so I upgraded it. Here is a photo of the inside - I have changed out the 0.8 uF AC coupling cap with a big, white, 20 uF one I found on Rob's desk.
The Busby Box is still working well. The 9V batteries have only run down to 7.8V. The original designer also put a spare AD743 (ultra low current FET amp) and a OP27 (best for ~kOhm source impedances) in there.
Here's the noise after the fix. There's no change in the DC noise, but the AC noise is much lower than before:
I think that the AC coupled noise is higher because we are seeing the current noise of the opamp. In the DC coupled case, the impedance to ground from the input pins of the opamp is very low and so the current noise is irrelevant.
The change I implemented, puts in a corner frequency of fc = 1/2/pi/R/C = 1/2/pi/10e3/20e-6 = 0.8 Hz.
Overall, the box is pretty good. Not great in terms of current noise and so it misses getting an A+. But its easily a solid A-.
I've measured the voltage noise of the SR560's lead acid battery outputs; they're not so bad.
Steve ordered us some replacement lead-acid batteries for our battery powered pre-amps (SR560). In the unit he replaced, I measured the noise using the following setup:
SR560 Busby Box
(+12V/GND) -------------AC Input Out ---------------- SR785
The SR785 was DC coupled and auto-ranged. The input noise of the SR785 was measured via 50 Ohm term to be at least 10x less than the SR560's noise at all frequencies.
Its clear that this measurement was spoiled by the low frequency noise of the Busby box below 10 Hz. Needs a better pre-amp.
The circuit design of multi-resonant EOM have proceeded.
In fact there are 6 parameters (Lp, L1, L2, Cp, C1, C2) in the circuit to be determined.
In general the less parameter gives the less calculation time with performing the numerical analysis. Of course it looks 6 parameters are little bit large number.
In order to reduce the arbitrary parameters, I put 4 boundary conditions.
Each boundary conditions fixed resonant peaks and valleys; first peak=11MHz, third peak=55MHz, first valley=19MHz, second valley=44MHz.
So now the remaining arbitrary parameters successfully get reduced to 2. Only we have to do is optimize the second peak as it to be 29.5MHz.
Then I take C1 and C2 as free parameters seeing how the second peak agree with 29.5MHz by changing the value of the C1 and C2.
the red color represents the good agreement with 29.5MHz, in contrast blue contour represents the bad.
You can see some best choice along the yellow belt. Now what we should do is to examine some of that and to select one of those.
In designing the whole circuit it is better to know the characteristic of the EOM.
I made impedance measurement with the EOM (New Focus model 4064) and I found it has capacitance of 10pF.
This is good agreement with the data sheet which says "5-10pF".
The measured plot is attached below. For comparison there also plotted "open" and "10pF mica".
In the interested band( from 1MHz to 100MHz), EOM looks just a capacitor.
But indeed it has lead inductance of 12nH, resistance of 0.74[Ohm], and parasitic capacitance of 5.5pF.
In some case we have to take account of those parasites in designing.
How can I get those values from the figure?
But indeed it has lead inductance of 12nH, resistance of 0.74[Ohm], and parasitic capacitance of 5.5pF.
Now I am studying about the behavior of the Q-factor in the resonant circuit because the Q-factor of the circuit directly determine the performance as the EOM driver.
Here I summarize the fundamental which explains why Q-factor is important.
The EOM driver circuit can be approximately described as shown in figure below
Z represents the impedance of a resonant circuit.
In an ideal case, the transformer just raise the voltage level n-times larger. Rin is the output impedance of the signal source and usually has 50[Ohm].
The transformer also makes the impedance Z 1/n^2 smaller. Therefore this configuration gives a following relation between Vin and Vout.
Where G is the gain for the voltage. And G goes to a maximum value when Rin=Z/n2. This relation is shown clearly in the following plot.
Note that I put Rin=50 [Ohm] for calculating the plot.
Under the condition Rin=Z/n2( generally referred as impedance matching ), the maximum gain can be expressed as;
It means that larger Z makes more efficient gain. In our case, interested Z is considered as the impedance at a resonance.
The key point of the story is:
"The recipe to exploit maximum benefit from a resonant EOM"
- Make a resonant EOM circuit. Measure the impedance Z at the resonance.
- This Z determines the optimum turn ratio n of the step-up transformer.
(n2 = Z/Rin where Rin is 50Ohm in our case.)
- This n gives the maximum gain Gmax (= n/2) that can be obtained with the step up transformer.
And, the impedance matching is also satisfied in this condition.
OK: The larger Z, the better. The higher Q, the Z larger, thus the better.
(Although the relationship between Z and Q were not described in the original post.)
So, how can we make the Q higher? What is the recipe for the resonant circuit?
=> Choose the components with smaller loss (resistance). The details will be provided by Kiwamu soon???
When I was young (3 months ago), I thought...
I was just too thoughtless. In reality, they are closely related each other.
A high Q resonant circuit has a high residual resistance at the resonant frequency. As far as the impedance is higher than the equivalent output impedance of the driving circuit (i.e. Z>Rin n2), we get the benefit of increasing the turn ratio of the transformer. In other words, "the performance of the resonant EOM is limited by the turn ratio of the transformer." (give us more turns!)
OK. So can we increase the turn ratio infinitely? No. Once Rin n2 gets larger than Z, you no longer get the benefit of the impedance transforming. The output impedance of the signal source yields too much voltage drop.
There is an optimum point for n. That is the above recipe.
So, a low Q resonant EOM has a destiny to be useless. But high Q EOM still needs to be optimized. As far as we use a transformer with a low turn ratio, it only shows ordinary performance.
Need to measure the length of the cable, but too lazy to use a measuring tape?
Then you too can become an expert cable length measurer by just using an RF signal generator and a scope:
The T is kind of acting like a beamsplitter in an asymmetric length Michelson in this case. Just as we can use the RF phase shift between the arms to measure the Schnupp asymmetry, we can also use a T to measure the cable length. The speed of light in the cable is documented in the cable catalog, but in most cases its just 66% of the speed of light in the vacuum.
I shorted the input to the box and then put its output into the SR560 (low noise, G = 100, AC). I put the output of the SR560 into the SR785.
*** BTW, the 2nd channel of the SR785 is kind of broken. Its too noisy by a factor of 100. Needs to go back for repair once we get started in the vac.
The attached PNG shows its input-referred noise with the short.
The picture shows the inside of the box before I did anything. The TO-5 package metal can is the meaty super dual-FET that gives this thing all of its low noise power.
In the spectra on the right are two traces. The BLUE one is the noise of the box as I found it. The BLACK one is the noise after I replaced R1, R6, R7, & R10 with metal film resistors.
The offset at the output of the box with either an open or shorted input is +265 mV.
I think we probably should also replace R2, R3, & R1, but we don't have any metal film resistors lower than 100 Ohms in the kit...but hopefully Steve will read this elog and do the right thing.
The validation for high impedance measurement has been well done.
The impedance measurement is one of the keys for designing the EOM circuit.
So far I was very struggling to measure the high impedance ( above several 1000 Ohm) at RF because the EOM circuit has a high impedance at its resonance.
Finally I realized that the measured impedance was suppressed by a parasitic resistance, which especially reduces the impedance at the resonance.
Also I found that we can extract the TRUE impedance data by subtracting the effect of the parasitic resistance from resultant data.
In order to confirm whether this subtraction works correctly or not, the impedance was directly re-measured with another analyzer for crosscheck.
The measurement has been performed with help from Peter and Frank. ( Thank you !)
By using network analyzer AG4395A with the impedance test kit AG43961A (these are at the PSL lab.), the impedance of resonant circuit with EOM was measured.
The picture of setup is attached. This impedance test kit allows to measure typically 0.1 [Ohm]-1M [Ohm] and frequency range of 100kHz-500MHz.
The resultant plot is attached. In the plot the blue curve represents the impedance measured by usual analyzer at 40m.
Note this curve is already subtracted the effect of the parasitic resistance.
( the parasitic resistance is in parallel to the circuit and it has ~7.8k Ohm, which is measured while the probe of the analyzer stays open. )
The red curve is the re-measured data using the impedance test kit.
The important point is that; these two peak values at the resonance around 40MHz show good agreement in 10%.
The resonant frequencies for two data differs a little bit, which might be the effect of a stray capacitance ( ~several [pF] )
The red curve has a structure around 80MHz, I think this comes from the non-coaxial cables, which connect the circuit and analyzing kit.
You can see these cables colored black and red in the picture.
( conclusion )
Our measurement with the subtraction of the parasitic resistance effect is working reliably !
I found that MCT QPD has dependence of the total output on the position of the spot. Since the QPD needs the supply and bias voltages from the sum/diff amp, I could not separate the problems of the QPD iteself and the sum/diff amplifier by the investigation on Tuesday. On Wednesday, I investigated a generic quad photodiode interface module D990692.
This is indeed sad. But, we can perhaps bypass all of this by just using the individual segment outputs. According to the circuit diagram and the c1iool0 .db file, we should be able to just do the math on the segments and ignore the VERT/HOR/SUM signals completely. In that case, we can just use high impedance for the sum/diff buffers as Koji says and not suffer from the calibration errors at all I think.
Unfortunately, the signals for individual segments also suffer from the voltage drop as all of the low impedance amplifiers are hung from the same input.
In order to utilize the individual channels, we anyway have to remove the resistors for the VERT/HOR/SUM amps.
That is possible. But does it disable some fast channels for future ASC purposes?
FYI: I stored the Universal PDH boxes in the RF cabiner in the Y arm.
The first design of the triple resonant EOM circuit has been done.
If only EOM has a loss of 4 Ohm, the gain of the circuit is expected to be 11 at 55MHz
So far I've worked on investigation of the single resonant circuit and accumulated the knowledge about resonant circuits.
Then I started the next step, designing the triple resonant circuit.
Here I report the first design of the circuit and the expected gain.
( What I did )
At first in order to determine the parameters, such as inductors and capacitors, I have solved some equations with numerical ways (see the past entry).
In the calculation I put 6 boundary conditions as followers;
(first peak=11MHz, second peak=29.5MHz, third peak=55MHz, first valley=22MHz, second valley=33MHz, Cp=18pF)
The valley frequencies of 22MHz and 33MHz are chosen in order to eliminate the higher harmonics of 11MHz, and Cp of 18pF represents the capacitance of the EOM.
Basically the number of parameters to be determined is 6 ( L1, L2, ...,), therefore it is completely solved under 6 boundary conditions. And in this case, only one solution exists.
The point is calculation does not include losses because the loss does not change the resonant frequency.
( results )
As a result I obtained the 6 parameters for each components shown in the table below.
Then I inserted the loss into the EOM to see how the impedance looks like. The loss is 4 Ohm and inserted in series to the EOM. This number is based on the past measurement .
Let us recall that the gain of the impedance-matched circuit with a transformer is proportional to square-root of the peak impedance.
It is represented by G = sqrt(Zres/50) where Zres is the impedance at the resonance.
As you can see in the figure, Zres = 6.4 kOhm at 55MHz, therefore the gain will be G=11 at 55MHz.
Essentially this gain is the same as that of the single resonant circuit that I've been worked with, because its performance was also limited mainly by the EOM loss.
An interesting thing is that all three peaks are exactly on the EOM limited line (black dash line), which is represented by Zres = L/CR = 1/ (2pi f Cp)^2 R. Where R = 4 Ohm.
( next plan )
There are other solutions to create the same peaks and valleys because of the similar solution.
It is easy to understand if you put Cp' = Cp x N, the solutions must be scaled like L1'=L1/N, C1'=C1 x N, ..., Finally such scaling gives the same resonant frequencies.
So the next plan is to study the effect of losses in each components while changing the similar solution.
After the study of the loss I will select an optimum similar solution.