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ID Date Author Type Categoryup Subject
  605   Mon Jun 30 15:56:22 2008 JenneUpdateElectronicsFixing the LO demod signal
To make the alarm handler happy, at Rana and John's suggestion I replaced R14 of the MC's Demod board, changing it from 4.99 Ohms to 4.99 kOhms. This increased the gain of the LO portion of the demod board by a factor of 10. Sharon and I have remeasured the table of LO input to the demod board, and the output on the C1:IOO-MC_DEMOD_LO channel:

Input Amplitude to LO input on demod board [dBm]: | Value of channel C1:IOO-MC_DEMOD_LO
------------------------------------------------- | -----------------------------------
-10 | -0.00449867
-8 | 0.000384331
-6 | 0.0101503
-4 | 0.0296823
-2 | 0.0882783
0 | 0.2543
2 | 0.542397
4 | 0.962335
6 | 1.65572
8 | 2.34911
10 | 2.96925
  888   Tue Aug 26 18:19:16 2008 ranaOmnistructureElectronicsResistor Noise at the 40m
As Stefan points out in his recent ISS ilog entries at LLO, Daniel Sigg recently wrote a
recommendation memo on resistor and capacitor
choices: T070016.

While working on the PMC I have had to use leaded resistors and wondered about the noise. As it turns
out we have the RN series of 1/4 W resistors from Stackpole Electronics. The RN series are
metal film resistors (datasheet attached); metal film is what Sigg recommends for lowest flicker
noise.

So we are OK for using the Stackpole 1/4 W leaded resistors in low noise circuits.
Attachment 1: SEI-RN_RNM.pdf
SEI-RN_RNM.pdf SEI-RN_RNM.pdf
  915   Wed Sep 3 18:43:19 2008 YoichiConfigurationElectronicsTwo more active probes
I found two active probes, an HP41800A and a Tektronix P6201.
Thanks John for telling me you saw them before.
Now we have three active probes, wow !
We have to find or make a power supply for P6201.
The manual of the probe is attached.
Attachment 1: Tektronix-P6201-active-probe.pdf
Tektronix-P6201-active-probe.pdf Tektronix-P6201-active-probe.pdf Tektronix-P6201-active-probe.pdf Tektronix-P6201-active-probe.pdf Tektronix-P6201-active-probe.pdf Tektronix-P6201-active-probe.pdf Tektronix-P6201-active-probe.pdf Tektronix-P6201-active-probe.pdf
  939   Wed Sep 10 13:28:25 2008 YoichiSummaryElectronicsIOO rack lost -24V (recovered)
Alberto, Yoichi

This morning, the MC suddenly started to be unwilling to lock.
I found a large offset in the MC servo board.
It turned out that -24V was not supplied to the Eurocard crate of the IOO rack.
We found two loose cables (violet, that means -24V) around the cross connects with fuses.
We connected them back, and the -24V was back.
The MC locks fine now, and Alberto can continue his arm length experiment.
  1036   Wed Oct 8 22:23:43 2008 YoichiConfigurationElectronicsElectronics work bench cleanup
Yesterday, I cleaned up the electronics work bench. I figured that keeping the work bench
in order has larger effect on the work efficiency than buying expensive soldering stations.
Whoever works there should clean up the table after the work to the state shown on
the right side of the picture (at least).
If you leave the bench for a while (more than 30min) but intend to return later and
resume the work, please write your name and time on a piece of paper and put it on the bench.
Otherwise, your stuff can be taken away anytime.
Attachment 1: Cleanup.jpg
Cleanup.jpg
  1135   Fri Nov 14 17:41:50 2008 JenneOmnistructureElectronicsSweet New Soldering Iron
The fancy new Weller Soldering Iron is now hooked up on the electronics bench.

Accessories for it are in the blue twirly cabinet (spare tips of different types, CD, and USB cable to connect it to a computer, should we ever decide to do so.

Rana: the soldering iron has a USB port?
Attachment 1: newSolderingIron.JPG
newSolderingIron.JPG
  1139   Mon Nov 17 11:01:15 2008 AlbertoHowToElectronicsCalibrating the Frequency Standard of the Marconi
I locked the SRS Rubidium Frequency Standard FS275 to the the 1pps from the GPS. The specs from the manual provide a frequency accuracy of 5x10^-11, that is 5x10-4 @ 10 MHz, since this is the reference signal frequency we're are going to use.
The Marconi internal frequency standard is provided by a TCXO oscillator. The instrument can be set in either one of these ways: 1) Indirect Synchronization, by which the internal TCXO is phase-locked to the external frequency standard (i.e. the SRS FS275 in our case) 2) Direct Sync, in which the internal TCXO is bypassed and the frequency standard is the external one.

I checked the specs of both frequency standards and found:

SRS FS275: 5x10^-11 -> 5x10^-10 Hz @ 10 MHz

Marconi: here what the data sheet says is that "the temperature coefficient is 7 in 10^7 in the temperature range between 0 and 55 C" and so should be also the frequency accuracy.

The SRS FS275 seems more accurate than the TCXO therefore I'm going to set the Marconi on the direct external mode.
Attachment 1: 2023ASeriesOperatingManual.pdf
2023ASeriesOperatingManual.pdf 2023ASeriesOperatingManual.pdf 2023ASeriesOperatingManual.pdf 2023ASeriesOperatingManual.pdf
Attachment 2: SRS_FS275_Rubidium_Frequency_Standard.pdf
SRS_FS275_Rubidium_Frequency_Standard.pdf SRS_FS275_Rubidium_Frequency_Standard.pdf
  1146   Wed Nov 19 10:32:11 2008 AlbertoConfigurationElectronicsAll the Marconi Set to the Rubidium Frequency Standard
I placed the SRS Rubidium FS275 over the PSL rack, next to the frequency counter. This one and the Marconi on the PSL rack have been connected to the 10MHz output of the frequency standard. I set also the first Marconi, the one that used to drive the others, to external, direct frequency reference. Now it reads 166981718 Hz versus 166981725 Hz measured by the frequency counter: 8 Hz difference.
  1147   Wed Nov 19 18:02:18 2008 ranaConfigurationElectronicsAll the Marconi Set to the Rubidium Frequency Standard
Not sure what was going on before. I changed the frequency counter to use an AC coupled input, and had it average
for 50 seconds. The output now agrees with the Marconi front panel to less than 1 Hz. Its still not 0.000 Hz,
but not bad.
  1208   Tue Dec 30 18:51:18 2008 rana,yoichiConfigurationElectronicsIlluminator Power Supply reset
We noticed that none of the illuminators were working.

The switches were off on all the ports. After turning them on it still didn't work.

The +24 V Sorensen power supply which powers all of the illuminators had its OVP light on.
We turned it off, ramped the voltage to zero, turned it back on, and then went back to +24 V.

We then checked the operation of the illuminators; ETMY is still MIA.

Each of the illuminators sucks ~0.6-0.7 A when the (unlabeled) rheostat knob panel is set
to the "25" setting.

It seems pretty unwise, in the EMI sense, to be sending Amps of unshielded, high current,
switching supply outputs for 40m down the arms. This creates a huge antenna for radiating
the switching noise. I hereby assign minus 5 points to whoever designed this system.

Illuminator Upgrade:
- Use LEDs of a wavelength that the OSEMs don't see. LEDs are also cool so that the
  Suspension won't drift in alignment.

- Use 2 power supplies so that the power is balanced.

- Make is +/-12 V twisted AWG 14 wire so that the EMI is contained. Should also
  be shielded cable.
  1252   Sat Jan 24 11:50:24 2009 AlbertoConfigurationElectronicsPhotodiode Filters' Transfer Functions
I found an elog entry by Jenne with the measurement of the transfer functions of the filters of some of our photodetectors. Since it might turn useful to us these days, while I'm thinking about posting them on the wiki sometime, I also copy the link here:
Jenne's on the PD's TF

If we still had the data for those plots, it would be great. Do we have it?
  1315   Mon Feb 16 23:09:52 2009 ranaUpdateElectronicsMC Servo Board offset gone bad!

The attached plot shows that someone broke the MC_SUM_MON channel around 10:30 AM this past Wednesday the 11th. This is the EPICS monitor of the MC error point.

Come forward now with your confession and I promise that I won't let Steve hurt you.

Attachment 1: mcoff.png
mcoff.png
  1326   Thu Feb 19 22:40:33 2009 KiwamuUpdateElectronicsPSL angle QPD

I checked a broken QPD, which was placed for PSL angle monitor, and finally I cocluded one segment of the quadrant diode was broken.

The broken segment has a offset voltage of -0.7V after 1st I-V amplifier. It means the diode segment has a current offset without any injection of light.

Tomorrow I will check a new QPD for replacement.

Kiwamu IZUMI

 

  1345   Mon Mar 2 16:27:40 2009 AlbertoConfigurationElectronicsMC Drum Mode SR560 Preamplifier Replaced

Today I checked out the SR560 around the lab. I confirmed that the one with the label "channel A noisy" is effectively mulfuctioning.

It was coonected to the lock-in amplifier set up for the drum mode peak readout.

I repleaced that with a working one.

  1407   Mon Mar 16 15:19:52 2009 OsamuDAQElectronicsSR785

I borrowed SR785 to measure AA, AI noise and TF.

  1445   Mon Mar 30 15:51:27 2009 steveUpdateElectronicsHP4291A left the lab to be repaired

Eric Gustafson is handling the old HP4291A rehabilitation. Tarac picked both units up today.

March of 2008 Tucker Electronics failed to fix it's intermittent ~25MHz 0.5V oscillation at the swept sine output

See 40m-elog id:398 on 3-24-2008 by Rob Ward

 

  1476   Sun Apr 12 19:31:43 2009 ranaSummaryElectronicsAmphony 2500 Headphones
We bought the Amphony 2500 Digital Wireless headphones recently. The other cheapo headphones we have are OK for control room use, but have a lot of noise
and are, therefore, not useful for noise hunting.

The new digital ones are pretty much noise-free. With the transmitter next to rosalba, you can walk halfway down the east arm and all around the MC area
before the reception goes bad. For real noise hunting, we will want to put the transmitter next to the BS chamber and take an analog pickoff from the DC PDs.

In the OMC diagram, we should put an AUDIO filterbank and wire it to the DAC so that we can do arbitrary IIR filtering on the audio signal.
  1664   Wed Jun 10 01:52:34 2009 AlbertoUpdateElectronicsMC length and Marconis' frequencies

Pete, Rob, Alberto,

yesterday we thought that some of the problems we were having in locking the IFO might be related to a change of the length of the mode cleaner. So today we decided to measure it again.

We followed the Sigg-Frolov technique (see 40m Wiki, Waldman, Fricke). For the record, the MC_AO input corresponds to IN2 on the MC Servo board.

We obtained: L = 27.092 +/- 0.001 m

From the new measurement we reset the frequencies of the Marconis to the following values:

33196450 Hz

132785800 Hz

165982250 Hz

199178700 Hz

 

  1681   Tue Jun 16 20:03:41 2009 AlbertoUpdateElectronicsRequirements on Wenzel Multiplier

For the 40m Upgrade, we plan to eliminate the Mach-Zehnder and replace it with a single EOM driven by all three modulation frequencies that we'll need: f1=11MHz, f2=5*f1=55MHz, fmc=29.5MHz.

A frequency generator will produce the three frequencies and with some other electronics we'll properly combine and feed them to the EOM.

The frequency generator will have two crystals to produce the f1 and fmc signals. The f2 modulation will be obtained by a frequency multiplier (5x) from the f1.

The frequency multiplier, for the way it works, will inevitably introduce some unwanted harmonics into the signals. These will show up as extra modulation frequencies in the EOM.

In order to quantify the effects of such unwanted harmonics on the interferometer and thus to let us set some limits on their amplitude, I ran some simulations with Optickle. The way the EOM is represented is by three RF modulators in series. In order to introduce the unwanted harmonics, I just added an RF modulator in series for each of them. I also made sure not to leave any space in between the modulators, so not to introduce phase shifts.

To check the effect at DC I looked at the sensing matrix and at the error signals. I considered the 3f error signals that we plan to use for the short DOFs and looked at how they depend on the CARM offset. I repeated the simulations for several possible amplitude of the unwanted harmonics. Some results are shown in the plots attached to this entry. 'ga' is the amplitude ratio of the unwanted
harmonics relative to the amplitude of the 11 & 55 MHz modulations.

Comparing to the case where there are no unwanted harmonics (ga = 0), one can see that not considerable effect on the error signals for amplitudes 40dB smaller than that of the main sidebands. Above that value, the REFL31I signals, that we're going to use to control PRCL, will start to be distorted: gain and linearity range change.

So 40 dB of attenuation in the unwanted harmonics is probably the minimum requirement on the frequency multiplier, although 60dB would provide a safer margin.

I'm still thinking how to evaluate any AC effect on the IFO.

 

** TODO: Plot DC sweeps with a wider range (+/- 20 pm). Also plot swept sines to look for changes in TFs out to ~10 kHz.

Attachment 1: SummaryOfResult.pdf
SummaryOfResult.pdf SummaryOfResult.pdf SummaryOfResult.pdf SummaryOfResult.pdf SummaryOfResult.pdf SummaryOfResult.pdf SummaryOfResult.pdf SummaryOfResult.pdf
  1724   Wed Jul 8 18:46:56 2009 DmassAoGElectronicsBeam Scan Funky

The beam scan (which has been living in the bridge subbasement for a bit now) is in a state of imperfection.

I noticed that:

  • The waist reading seems to change by not insignificant amounts as you move the spot across the head, even for just small perturbations about the center.
  • None of the features which require two slits seem to be working (unsure if this is software or hardware related)

I took some pictures to try and illuminate the situation - The inverted images are included to make it easier to see the flecks (?) in the slits

I am not sure how to figure out if any bit of the scan is/has been fried.

 

Pending further investigation, enjoy large error bars in your scan measurements!

 

PICTURES OF BOTH SLITS ON THE BEAMSCAN HEAD:

Attachment 1: beamscanhead3.png
beamscanhead3.png
Attachment 2: beamscanhead6.png
beamscanhead6.png
  2014   Mon Sep 28 23:13:14 2009 JenneConfigurationElectronicsRob is breaking stuff....

Koji and I were looking for an extender card to aid with MZ board testing.  Rob went off on a quest to find one.  He found 2 (in addition to the one in the drawer near the electronics bench which says "15V shorted"), and put them in some empty slots in 1X1 to test them out.  Somehow, this burned a few pins on each board (1 pin on one of them, and 3 pins on the other). We now have 0 functioning extender cards: unfortunately, both extender cards now need fixing.  The 2 slots that were used in 1X1 now have yellow electrical tape covering the connectors so that they do not get used, because the ends of the burnt-off pins may still be in there. 

In other, not-Rob's-fault news, the Martian network is down...we're going to try to reset it so that we have use of the laptops again.

  2019   Tue Sep 29 16:14:44 2009 robConfigurationElectronicsRob is breaking stuff....

Quote:

Koji and I were looking for an extender card to aid with MZ board testing.  Rob went off on a quest to find one.  He found 2 (in addition to the one in the drawer near the electronics bench which says "15V shorted"), and put them in some empty slots in 1X1 to test them out.  Somehow, this burned a few pins on each board (1 pin on one of them, and 3 pins on the other). We now have 0 functioning extender cards: unfortunately, both extender cards now need fixing.  The 2 slots that were used in 1X1 now have yellow electrical tape covering the connectors so that they do not get used, because the ends of the burnt-off pins may still be in there. 

In other, not-Rob's-fault news, the Martian network is down...we're going to try to reset it so that we have use of the laptops again.

 

This happened when I plugged the cards into a crate with computers, which apparently is a no-no.  The extender cards only go in VME crates full of in-house, LIGO-designed electronics.

  2064   Wed Oct 7 11:18:40 2009 kiwamuSummaryElectronicsracks of electronics

 

I took the pictures of all racks of electronics yesterday, and then uploaded these pictures on the wiki.

http://lhocds.ligo-wa.caltech.edu:8000/40m/Electronics

You can see them by clicking "pictures" in the wiki page.

 

  2110   Sun Oct 18 19:55:45 2009 ranaConfigurationElectronicsIP POS is back: ND filter gone, new resistors in

Its back in and re-centered. Our next move on IPPOS should be to replace its steering mirror with something bigger and more rigid.

Electronics changes:

20K -> 3.65 K  (R6, R20, R42, R31) (unused)

20K -> 3.65 K  (R7, R21, R32, R43, R11, R24, R35, R46)

If you look in the schematic (D990272), you see that its an AD797 transimpedance stage with a couple of LT1125 stages set to give some switchable gain. It looks like some of these

switches are on and some are not, but I am not sure where it would be controlled from. I've attached a snapshot of one quadrant of the schematic below.

The schematic shows the switches in the so-called 'normally closed' configuration. This is what the switches do with zero volts applied to the control inputs. As the schematic also shows,

just disconnecting the 'switch' inputs cause the switch's control inputs to go high (normally open configuration, i.e. pins 2-3 connected, pin4 open). For the record, the default positions of the IPPOS switches are:

switch1   high

switch2   low

switch3   low

switch4   high

 


** EDIT (Nov 2, 2009): I forgot to attach the before and after images; here they are:

IMG_0068.JPGIMG_0072.JPG

  2112   Sun Oct 18 22:06:15 2009 ranaConfigurationElectronicsIP POS is back: ND filter gone, new resistors in

I tried to compare the IP_POS time series with the IPANG and MC_TRANS but was foiled at first:

1) The IPANG scan rate was set to 0.5 second, so it doesn't resolve the pendulum motions well. Fixed in the .db file.

2) Someone had used a Windows/DOS editor to edit the .db file and it was filled with "^M" characters. I have removed them all using this command:   tr -d "\r" <ETMXaux.db > new.db

3) The MC_TRANS P/Y channels were on the MC Lock screen but had never been added to the DAQ. Remember, if there's a useful readback on an EPICS screen. its not necessarily in the frames unless you add it to the C0EDCU file. I have done that now and restarted the fb daqd. Channels now exist.

4) Changed the PREC of the IPPOS channels to 3 from 2.

5) changed the sign for the IBQPD (aka IPANG) so that bigger signal is positive on the EPICS screen.

Attachment 1: Untitled.png
Untitled.png
  2118   Mon Oct 19 14:48:15 2009 rana, robSummaryElectronicspiezo jena measuring box
Attached is the schematic of the Piezo Jena driver measuring box made in a Pomona box:
                2.2 uF
In ----o-------- | | --------o-------- Out
       |                     |
       _                     |
       _  1uF                R  7.5 kOhms
       |                     |
       |                     |
      GND                   GND
The 1 uF cap is there to simulate the piezo and the 2.2 uF and 7.5k resistor ac couple the signal for the spectrum analyzer. They give a ~10 Hz corner frequency.
Attachment 1: PA160153.JPG
PA160153.JPG
Attachment 2: PA160151.JPG
PA160151.JPG
  2244   Wed Nov 11 20:57:06 2009 kiwamuUpdateElectronicsMulti-resonant EOM --- LC tank circuit ---

I have been working about multi-resonant EOM since last week.

In order to characterize the behavior of the each components, I have made a simple LC tank circuit.

You can see the picture of the circuit below.

DSCN0160.JPG

Before constructing the circuit, I made an "ideal" calculation of the transfer function without any assumptions by my hand and pen.

Most difficult part in the calculation is the dealing with a transformer analytically. Eventually I found how to deal with it in the analytical calculation.

The comparison of the calculated and measured transfer function is attached.

 It shows the resonant frequency of ~50MHz as I expected. Those are nicely matched below 50MHz !!

For the next step, I will make the model of the circuit with stray capacitors, lead inductors, ... by changing the inductance or something. 

 

Attachment 2: LCtank_complete.png
LCtank_complete.png
  2262   Fri Nov 13 03:38:47 2009 kiwamuUpdateElectronicsmulti-resonant EOM --- impedance of LC tank circuit ----

I have measured the impedance of the LC tank circuit which I referred on my last entry.

The configuration of the circuit is exactly the same as that time.

In order to observe the impedance, by using Koji's technique I injected a RF signal into the output of the resonant circuit.

In addition I left the input opened, therefore the measured impedance does not include the effect of the transformer.

 

- - - - - - - - - - - - results

The measured impedance is attached below; "LCtank_impedance.png"

The peak around 50MHz is the main resonance and it has impedance of ~1500 [Ohm], which should go to infinity in the ideal case (no losses).

In fact the impedance looked from the input of the circuit gets reduced by 1/n^2, where "n" is the turn ratio of the transformer.

By putting the n=4, the input impedance of the circuit should be 93 [Ohm]. This is a moderate value we can easily perform impedance-matching by some technique.

I also fitted the data with a standard model of equivalent circuit (see attachment 2).

In the figure.2 red component and red letter represents the design. All the other black stuff are parasites.

But right now I have no idea the fitted value is reasonable or not.

For the next I should check the input impedance again by the direct way; putting the signal into the input.

 

 

 

Attachment 1: LCtank_impedance.png
LCtank_impedance.png
Attachment 2: LCtank_model.png
LCtank_model.png
  2263   Fri Nov 13 05:03:09 2009 kiwamuUpdateElectronicsmulti-resonant EOM --- input impedance of LC tank ----

I measured the input impedance of the LC tank circuit with the transformer. The result is attached.

It looks interesting because the input impedance is almost dominated

by the primary coil of the transformer with inductance of 75nH (see attachment 1).

The impedance at the resonance is ~100 [Ohm], I think this number is quite reasonable because I expected that as 93 [Ohm]

 

Note that the input impedance can be derived as follower;

(input impedance) = L1 + Z/n^2.

Where L1 is the inductance of the primary coil, Z is the load in the secondary loop and n is the turn ratio.

 

I think now I am getting ready to enter the next phase \(^o^)/

Attachment 1: input_impedance.png
input_impedance.png
Attachment 2: input_impedance2.png
input_impedance2.png
  2286   Tue Nov 17 21:10:35 2009 ranaSummaryElectronicsBusby Low Noise Box: Photos and Upgrades

IMG_0217.JPG

It looked like the Busby Low Noise Box had too much low frequency noise and so I upgraded it. Here is a photo of the inside - I have changed out the 0.8 uF AC coupling cap with a big, white, 20 uF one I found on Rob's desk.

The Busby Box is still working well. The 9V batteries have only run down to 7.8V. The original designer also put a spare AD743 (ultra low current FET amp) and a OP27 (best for ~kOhm source impedances) in there.

Here's the noise after the fix. There's no change in the DC noise, but the AC noise is much lower than before:

busby-noise.png

I think that the AC coupled noise is higher because we are seeing the current noise of the opamp. In the DC coupled case, the impedance to ground from the input pins of the opamp is very low and so the current noise is irrelevant.

The change I implemented, puts in a corner frequency of fc = 1/2/pi/R/C = 1/2/pi/10e3/20e-6 = 0.8 Hz.

Overall, the box is pretty good. Not great in terms of current noise and so it misses getting an A+. But its easily a solid A-.

  2288   Wed Nov 18 00:38:33 2009 ranaSummaryElectronicsVoltage Noise of the SR560's OUTPUTs (the back panel)

I've measured the voltage noise of the SR560's lead acid battery outputs; they're not so bad.

Steve ordered us some replacement lead-acid batteries for our battery powered pre-amps (SR560). In the unit he replaced, I measured the noise using the following setup:

SR560                              Busby Box

(+12V/GND) -------------AC Input      Out  ----------------   SR785

The SR785 was DC coupled and auto-ranged. The input noise of the SR785 was measured via 50 Ohm term to be at least 10x less than the SR560's noise at all frequencies.

sr560.png

Its clear that this measurement was spoiled by the low frequency noise of the Busby box below 10 Hz. Needs a better pre-amp.

  2292   Wed Nov 18 14:55:59 2009 kiwamuUpdateElectronicsmulti-resonant EOM --- circuit design ----

The circuit design of multi-resonant EOM have proceeded.

By using numerical method, I found the some best choice of the parameters (capacitors and inductors).

In fact there are 6 parameters (Lp, L1, L2, Cp, C1, C2) in the circuit to be determined.

whole_circuit.png

In general the less parameter gives the less calculation time with performing the numerical analysis. Of course it looks 6 parameters are little bit large number.

In order to reduce the arbitrary parameters, I put 4 boundary conditions.

Each boundary conditions fixed resonant peaks and valleys; first peak=11MHz, third peak=55MHz, first valley=19MHz, second valley=44MHz.

designed.png

So now the remaining arbitrary parameters successfully get reduced to 2. Only we have to do is optimize the second peak as it to be 29.5MHz.

Then I take C1 and C2 as free parameters seeing how the second peak agree with 29.5MHz by changing the value of the C1 and C2.

mont.png

the red color represents the good agreement with 29.5MHz, in contrast blue contour represents the bad.

 You can see some best choice along the yellow belt. Now what we should do is to examine some of that and to select one of those.

  2294   Wed Nov 18 16:58:36 2009 kiwamuUpdateElectronicsmulti-resonant EOM --- EOM characterization ---

In designing the whole circuit it is better to know the characteristic of the EOM.

I made impedance measurement with the EOM (New Focus model 4064) and I found it has capacitance of 10pF.

This is good agreement with the data sheet which says "5-10pF".

The measured plot is attached below. For comparison there also plotted "open" and "10pF mica".

In the interested band( from 1MHz to 100MHz), EOM looks just a capacitor.

But indeed it has lead inductance of 12nH, resistance of 0.74[Ohm], and parasitic capacitance of 5.5pF.

In some case we have to take account of those parasites in designing.

EOM_impedance.png

 

  2295   Wed Nov 18 22:38:17 2009 KojiUpdateElectronicsmulti-resonant EOM --- EOM characterization ---

How can I get those values from the figure?

Quote:

But indeed it has lead inductance of 12nH, resistance of 0.74[Ohm], and parasitic capacitance of 5.5pF. 

 

  2340   Wed Nov 25 20:44:48 2009 kiwamuUpdateElectronicsMulti-resonant EOM --- Q-factor ----

Now I am studying about the behavior of the Q-factor in the resonant circuit because the Q-factor of the circuit directly determine the performance as the EOM driver.

Here I summarize the fundamental which explains why Q-factor is important.

 --------------------------------------

The EOM driver circuit can be approximately described as shown in figure below

trans.png

Z represents the impedance of a resonant circuit.

In an ideal case, the transformer just raise the voltage level n-times larger.  Rin is the output impedance of the signal source and usually has 50[Ohm].

The transformer also makes the impedance Z 1/n^2 smaller. Therefore this configuration gives a following relation between Vin and Vout.

eq1.png

 Where G is the gain for the voltage. And G goes to a maximum value when Rin=Z/n2. This relation is shown clearly in the following plot.

 

impedance.png

 Note that I put Rin=50 [Ohm] for calculating the plot.

Under the condition  Rin=Z/n2( generally referred as impedance matching ), the maximum gain can be expressed as;

eq2.png

 

It means that larger Z makes more efficient gain. In our case, interested Z is considered as the impedance at a resonance.

So what we should do is making a resonant circuit which has a higher impedance at the resonance (e.g. high Q-resonant circuit).

 

 

  2341   Thu Nov 26 02:08:34 2009 KojiUpdateElectronicsMulti-resonant EOM --- Q-factor ----

The key point of the story is:
"The recipe to exploit maximum benefit from a resonant EOM"
- Make a resonant EOM circuit. Measure the impedance Z at the resonance.
- This Z determines the optimum turn ratio n of the step-up transformer.
 
(n2 = Z/Rin where Rin is 50Ohm in our case.)
- This n gives the maximum gain Gmax (= n/2) that can be obtained with the step up transformer.
  And, the impedance matching is also satisfied in this condition.

OK: The larger Z, the better. The higher Q, the Z larger, thus the better.
(Although the relationship between Z and Q were not described in the original post.)

So, how can we make the Q higher? What is the recipe for the resonant circuit?
=> Choose the components with smaller loss (resistance). The details will be provided by Kiwamu soon??? 


When I was young (3 months ago), I thought...

  • Hey! Let's increase the Q of an EOM! It will increase the modulation!
  • Hey! Let's use the step-up transformer with n as high as possible! It will increase the modulation!
  • Hey! Take the impedance matching! It will increase the modulation!

I was just too thoughtless. In reality, they are closely related each other.

A high Q resonant circuit has a high residual resistance at the resonant frequency. As far as the impedance is higher than the equivalent output impedance of the driving circuit (i.e. Z>Rin n2), we get the benefit of increasing the turn ratio of the transformer. In other words, "the performance of the resonant EOM is limited by the turn ratio of the transformer." (give us more turns!)

OK. So can we increase the turn ratio infinitely? No. Once Rin n2 gets larger than Z, you no longer get the benefit of the impedance transforming. The output impedance of the signal source yields too much voltage drop.

There is an optimum point for n. That is the above recipe. 

So, a low Q resonant EOM has a destiny to be useless. But high Q EOM still needs to be optimized. As far as we use a transformer with a low turn ratio, it only shows ordinary performance.

 

 

  2403   Sat Dec 12 07:36:56 2009 ranaHowToElectronicsHow to Measure the Length of a Cable: Interferometry

Need to measure the length of the cable, but too lazy to use a measuring tape?

Then you too can become an expert cable length measurer by just using an RF signal generator and a scope:

  1. Disconnect or short (not 50 Ohm term) the far side of the cable.
  2. Put a T on the near side of the cable.
  3. Drive the input of the T with your signal source.
  4. Look at the output of the T with the scope while sweeping the signal source's frequency knob.

The T is kind of acting like a beamsplitter in an asymmetric length Michelson in this case. Just as we can use the RF phase shift between the arms to measure the Schnupp asymmetry, we can also use a T to measure the cable length. The speed of light in the cable is documented in the cable catalog, but in most cases its just 66% of the speed of light in the vacuum.

  2436   Mon Dec 21 01:14:08 2009 ranaSummaryElectronicsNoise measurement of the Rai Weiss FET preamp box

 I shorted the input to the box and then put its output into the SR560 (low noise, G = 100, AC). I put the output of the SR560 into the SR785.

*** BTW, the 2nd channel of the SR785 is kind of broken. Its too noisy by a factor of 100. Needs to go back for repair once we get started in the vac.

The attached PNG shows its input-referred noise with the short.

The picture shows the inside of the box before I did anything. The TO-5 package metal can is the meaty super dual-FET that gives this thing all of its low noise power.

Picture_2.pngRWnoise.png

In the spectra on the right are two traces. The BLUE one is the noise of the box as I found it. The BLACK one is the noise after I replaced R1, R6, R7, & R10 with metal film resistors.

The offset at the output of the box with either an open or shorted input is +265 mV.

I think we probably should also replace R2, R3, & R1, but we don't have any metal film resistors lower than 100 Ohms in the kit...but hopefully Steve will read this elog and do the right thing.

Attachment 1: IMG_0242.JPG
IMG_0242.JPG
  2450   Thu Dec 24 01:25:29 2009 kiwamuUpdateElectronicsimpedance analyzing

The validation for high impedance measurement has been well done.

The impedance measurement is one of the keys for designing the EOM circuit.

So far I was very struggling to measure the high impedance ( above several 1000 Ohm) at RF because the EOM circuit has a high impedance at its resonance.

Finally I realized that the measured impedance was suppressed by a parasitic resistance, which especially reduces the impedance at the resonance.

Also I found that we can extract the TRUE impedance data by subtracting the effect of the parasitic resistance from resultant data.

In order to confirm whether this subtraction works correctly or not,  the impedance was directly re-measured with another analyzer for crosscheck.

                The followers are details about the re-measurement.
 

 

(measurement )

The measurement has been performed with help from Peter and Frank. ( Thank you !)

By using  network analyzer AG4395A with the impedance test kit AG43961A (these are at the PSL lab.), the impedance of resonant circuit with EOM was measured.

The picture of setup is attached. This impedance test kit allows to measure typically 0.1 [Ohm]-1M [Ohm] and frequency range of 100kHz-500MHz.

 

(result)
The resultant plot is attached. In the plot the blue curve represents the impedance measured by usual analyzer at 40m.

Note this curve is already subtracted the effect of the parasitic resistance.

( the parasitic resistance is in parallel to the circuit and it has ~7.8k Ohm, which is measured while the probe of the analyzer stays open. )

The red curve is the re-measured data using the impedance test kit.

The important point is that; these two peak values at the resonance around 40MHz show good agreement in 10%.

The resonant frequencies for two data differs a little bit, which might be the effect of a stray capacitance ( ~several [pF] )

The red curve has a structure around 80MHz, I think this comes from the non-coaxial cables, which connect the circuit and analyzing kit.

You can see these cables colored black and red in the picture.

 

( conclusion )

Our measurement with the subtraction of the  parasitic resistance effect is working reliably !

Attachment 1: DSCN0421.JPG
DSCN0421.JPG
Attachment 2: EOM.png
EOM.png
  2454   Sun Dec 27 23:44:59 2009 ranaUpdateElectronicsMCT QPD investigation

Quote:

I found that MCT QPD has dependence of the total output on the position of the spot. Since the QPD needs the supply and bias voltages from the sum/diff amp, I could not separate the problems of the QPD iteself and the sum/diff amplifier by the investigation on Tuesday. On Wednesday, I investigated a generic quad photodiode interface module D990692.

 This is indeed sad. But, we can perhaps bypass all of this by just using the individual segment outputs. According to the circuit diagram and the c1iool0 .db file, we should be able to just do the math on the segments and ignore the VERT/HOR/SUM signals completely. In that case, we can just use high impedance for the sum/diff buffers as Koji says and not suffer from the calibration errors at all I think.

  2455   Mon Dec 28 01:17:01 2009 KojiUpdateElectronicsMCT QPD investigation

Unfortunately, the signals for individual segments also suffer from the voltage drop as all of the low impedance amplifiers are hung from the same input.
In order to utilize the individual channels, we anyway have to remove the resistors for the VERT/HOR/SUM amps.
That is possible. But does it disable some fast channels for future ASC purposes?

 

Quote:

 This is indeed sad. But, we can perhaps bypass all of this by just using the individual segment outputs. According to the circuit diagram and the c1iool0 .db file, we should be able to just do the math on the segments and ignore the VERT/HOR/SUM signals completely. In that case, we can just use high impedance for the sum/diff buffers as Koji says and not suffer from the calibration errors at all I think.

 

  2476   Tue Jan 5 09:18:38 2010 AlbertoOmnistructureElectronicsUniversal PDH Box Stored in the RF Cabinet

FYI: I stored the Universal PDH boxes in the RF cabiner in the Y arm.

  2523   Mon Jan 18 23:44:19 2010 kiwamuUpdateElectronicstriple resonant circuit for EOM

The first design of the triple resonant EOM circuit has been done.

If only EOM has a loss of 4 Ohm, the gain of the circuit is expected to be 11 at 55MHz

So far I've worked on investigation of the single resonant circuit and accumulated the knowledge about resonant circuits.

Then I started the next step, designing the triple resonant circuit.

Here I report the first design of the circuit and the expected gain.

 


( What I did )

At first in order to determine the parameters, such as inductors and capacitors, I have solved some equations with numerical ways (see the past entry).

In the calculation I put 6 boundary conditions as followers;

(first peak=11MHz, second peak=29.5MHz, third peak=55MHz, first valley=22MHz, second valley=33MHz, Cp=18pF)

The valley frequencies of 22MHz and 33MHz are chosen in order to eliminate the higher harmonics of 11MHz, and Cp of 18pF represents the capacitance of the EOM.

Basically the number of parameters to be determined is 6 ( L1, L2, ...,), therefore it is completely solved under 6 boundary conditions. And in this case, only one solution exists.

The point is calculation does not include losses because the loss does not change the resonant frequency.

 

whole_circuit.png

( results )

As a result I obtained the 6 parameters for each components shown in the table below.

Cp [pF] 18.1
C1 [pF]  45.5
C2 [pF] 10.0
Lp [uH] 2.33
L1 [uH] 1.15
L2 [uH] 2.33

Then I inserted the loss into the EOM to see how the impedance looks like. The loss is 4 Ohm and inserted in series to the EOM. This number is based on the past measurement .

Let us recall that the gain of the impedance-matched circuit with a transformer is proportional to square-root of the peak impedance.

It is represented by G = sqrt(Zres/50) where Zres is the impedance at the resonance.

 As you can see in the figure, Zres = 6.4 kOhm at 55MHz, therefore the gain will be G=11 at 55MHz.

Essentially this gain is the same as that of the single resonant circuit that I've been worked with, because its performance was also limited mainly by the EOM loss.

 An interesting thing is that all three peaks are exactly on the EOM limited line (black dash line), which is represented by Zres = L/CR = 1/ (2pi f Cp)^2 R. Where R = 4 Ohm.

 designed_circuit.png

( next plan )

There are other solutions to create the same peaks and valleys because of the similar solution.

 It is easy to understand if you put Cp' = Cp x N, the solutions must be scaled like L1'=L1/N, C1'=C1 x N, ...,  Finally such scaling gives the same resonant frequencies.

So the next plan is to study the effect of losses in each components while changing the similar solution.

After the study of the loss I will select an optimum similar solution.

  2524   Tue Jan 19 00:10:44 2010 ranaUpdateElectronicstriple resonant circuit for EOM

Very cool.         

  2525   Tue Jan 19 02:39:57 2010 kiwamuUpdateElectronicsdesign complete --- triple resonant circuit for EOM ---

The design of the triple resonant circuit has been fixed.

I found the optimum configuration, whose gain is still 11 at 55MHz even if there are realistic losses.

As I mentioned in the last entry, there are infinite number of the similar solutions to create the same resonant frequencies.

However owing to the effect of the losses, the resultant gain varies if the similar solution changes

The aim of this study is to select the optimum solution which has a maximum gain ( = the highest impedance at the resonance ).

In order to handle the losses in the calculation, I modeled the loss for both inductors and the capacitors.

Then I put them into the circuit, and calculated the impedance while changing the solutions.

 


 

(method)

1). put the scaling parameter as k in order to create the similar solution.

2). scale the all electrical parameters (L1, L2,...) by using k, so that C1'=C1 x k, L1'=L1/k ,...

3). Insert the losses into all the electrical components

4). Draw the impedance curve in frequency domain.

5). See how the height of the impedance at the resonance change

6). Repeat many time this procedure with another k.

7). Find and select the optimum k

scaling.png

There is a trick in the calculation.

I put a capacitor named Cpp in parallel to the EOM in order to scale the capacitance of the EOM (see the schematic).

For example if we choose k=2, this means all the capacitor has to be 2-times larger.

For the EOM, we have to put Cpp with the same capacitance as Cp (EOM). As a result these two capacitors can be dealt together as 2 x Cp.

So that Cpp should be Cpp = (k-1) Cp, and Cpp vanishes when we choose k=1.

 

The important point is that the scaling parameter k must be greater than unity, that is k > 1.

This restriction directly comes from Cp, the capacitance of the EOM, because we can not go to less than Cp.

If you want to put k < 1, it means you have to reduce the capacitance of the EOM somehow (like cutting the EO crystal ??)

 

(loss model)

I've modeled the loss for both the inductors and the capacitors in order to calculate the realistic impedance.

The model is based on the past measurements I've performed and the data sheet.

   Loss for Capacitor :  R(C) = 0.5 (C / 10pF)^{-0.3} Ohm

   Loss for Inductor :    R(L) = 0.1 ( L / 1uH) Ohm

Of course this seems to be dirty and rough treatment.

But I think it's enough to express the tendency that the loss  increase / decrease monotonically as  L / C get increased.

These losses are inserted in series to every electrical components.

( Note that: this model depends on both the company and the product model. Here I assume use of Coilcraft inductors and mica capacitors scattered around 40m )

 

( results )

The optimum configuration is found when k=1, there is no scaling. This is the same configuration listed in last entry

Therefore we don't need to insert the parallel capacitor Cpp in order to achieve the optimum gain.

The figure below shows the some examples of the calculated impedance. You can see the peak height decrease by increasing the scale factor k.

realistic.png

The black dash line represents the EOM-loss limit, which only contains the loss of the EOM.

The impedance at the resonance of 55MHz is 6.2 kOhm, which decreased by 3% from the EOM-loss limit. This corresponds to gain of G = 11.

The other two peaks, 11MHz and 29.5MHz dramatically get decreased from EOM-loss limit.

I guess this is because the structure below 50MHz is mainly composed by L1, L2, C1, C2.

In fact these components have big inductance and small capacitance, so that it makes lossy.

 

( next step )

The next step is to choose the appropriate transformer and to solder the circuit.

  2526   Tue Jan 19 02:40:38 2010 KojiUpdateElectronicstriple resonant circuit for EOM

The design looks very good. I have some questions.

1. As far as I remember, you've got the gain of slightly worse than 10 for a 55MHz single resonant case. Why your expectation of the gain (G=11) for the highest resonance better than this?

Supposing the loss exists only on the EOM, the other part of the LC network for the triple work as an inductor at the resonant frequency. This is just equivalent as the single resonant case. So the expected gain at 55MHz should coincides with what we already have. Probably, the resistance of 4 Ohm that is used here had too rough precision???

2. How can you adjust the resonances precisely?

Do we need any variable components for Cs and Ls, that may have worse quality than the fixed one, generally speaking.
I myself has no experience that I had to tune the commercial EOM because of a drift or whatever. I hope if you can adjust the resonance with a fixed component it should be fine.

3. Changing Cp. What does it mean?

Do you put additional cap for Cp?

4. The resonances for the lower two look very narrow. Is that fine?

This will show up in a better shape if we look at the transfer function for the gain. Is this right?

If we have BW>100kHz, it is sufficient.

5. Impedance matching for the lower two resonances.

Yep. You know this problem already.

 

  2527   Tue Jan 19 03:04:14 2010 KojiUpdateElectronicstriple resonant circuit for EOM

Self-follow:

I got the answer of Q3 from the follow-up entry.

For Q4, once you get the impedance of the LC network lower than n^2*50, the EOM gain will be quite low. This means that the resonance is anyway narrow.
I did some simple calculation and it shows that the width of the resonance will be 100kHz~500kHz. So it maybe OK.

Quote:

The design looks very good. I have some questions.

1. As far as I remember, you've got the gain of slightly worse than 10 for a 55MHz single resonant case. Why your expectation of the gain (G=11) for the highest resonance better than this?

Supposing the loss exists only on the EOM, the other part of the LC network for the triple work as an inductor at the resonant frequency. This is just equivalent as the single resonant case. So the expected gain at 55MHz should coincides with what we already have. Probably, the resistance of 4 Ohm that is used here had too rough precision???

2. How can you adjust the resonances precisely?

Do we need any variable components for Cs and Ls, that may have worse quality than the fixed one, generally speaking.
I myself has no experience that I had to tune the commercial EOM because of a drift or whatever. I hope if you can adjust the resonance with a fixed component it should be fine.

3. Changing Cp. What does it mean?

Do you put additional cap for Cp?

4. The resonances for the lower two look very narrow. Is that fine?

This will show up in a better shape if we look at the transfer function for the gain. Is this right?

If we have BW>100kHz, it is sufficient.

5. Impedance matching for the lower two resonances.

Yep. You know this problem already. 

 

  2528   Tue Jan 19 03:20:28 2010 KojiUpdateElectronicsdesign complete --- triple resonant circuit for EOM ---

First I was confused, but now I think I understood.

My confusion:
If the k get bigger, L get smaller, C get bigger. This makes R(L) smaller and R(C) smaller. This sounds very nice. But why smaller k is preferable in the Kiwamu's result?

Explanation:
The resultant impedance of the network at a resonance is determined by Zres = L/(R C) or something like that. Here R = R(L)+R(C). (I hope this is right.)

Here larger Zres is preferable. So smaller R is nice.

But If the speed of reduction for R is slower than that of L/C (which is proportional to k^-2), increasing k does not help us to increase of Zres. And that's the case.

This means "if we can put the LC network in the box of EOM, we can do better job!" as we can reduce Cp.

Quote:

scaling.png


   Loss for Capacitor :  R(C) = 0.5 (C / 10pF)^{-0.3} Ohm

   Loss for Inductor :    R(L) = 0.1 ( L / 1uH) Ohm

  2529   Tue Jan 19 03:27:47 2010 kiwamuUpdateElectronicsRe: triple resonant circuit for EOM

1. You are right, the gain for the single resonant circuit was about 9.3 in my measurement.

But the reason why the triple is better than the single resonant circuit comes from the transformer.

The impedance can be degraded by a loss of the transformer, because it got worse after applying the transformer in the past measurement.

Also I definitely confirmed that the circuit had the impedance of 7.2 kOhm at the resonance of 52.9MHz without the transformer.

So it shall give the gain of 12, but did not after applying the transformer.

 

2.  Yes, I think we need some variable components just in case.

 

5.  For the impedance matching, I will select a transformer so that 55MHz is matched. In contrast I will leave two lower resonances as they are.

This is because 11MHz and 29.5MHz usually tend to have higher impedance than 55MHz. In this case, even if the impedance is mismatched, the gain for these can be kept higher than 11.

I will post the detail for this mismatched case tomorrow.

 

Quote:

The design looks very good. I have some questions.

1. As far as I remember, you've got the gain of slightly worse than 10 for a 55MHz single resonant case. Why your expectation of the gain (G=11) for the highest resonance better than this?

Supposing the loss exists only on the EOM, the other part of the LC network for the triple work as an inductor at the resonant frequency. This is just equivalent as the single resonant case. So the expected gain at 55MHz should coincides with what we already have. Probably, the resistance of 4 Ohm that is used here had too rough precision???

2. How can you adjust the resonances precisely?

Do we need any variable components for Cs and Ls, that may have worse quality than the fixed one, generally speaking.
I myself has no experience that I had to tune the commercial EOM because of a drift or whatever. I hope if you can adjust the resonance with a fixed component it should be fine.

3. Changing Cp. What does it mean?

Do you put additional cap for Cp?

4. The resonances for the lower two look very narrow. Is that fine?

This will show up in a better shape if we look at the transfer function for the gain. Is this right?

If we have BW>100kHz, it is sufficient.

5. Impedance matching for the lower two resonances.

Yep. You know this problem already.

 

 

  2533   Tue Jan 19 23:26:07 2010 kiwamuUpdateElectronicsRe:Re: triple resonant circuit for EOM

Quote:

5.  For the impedance matching, I will select a transformer so that 55MHz is matched. In contrast I will leave two lower resonances as they are.

This is because 11MHz and 29.5MHz usually tend to have higher impedance than 55MHz. In this case, even if the impedance is mismatched, the gain for these can be kept higher than 11.

I will post the detail for this mismatched case tomorrow.

 

Here the technique of the impedance matching for the triple resonant circuit are explained.

In our case, the transformer should be chosen so that the impedance of the resonance at 55MHz is matched.

We are going to use the transformer to step up the voltage applied onto the EOM.

To obtain the maximum step-up-gain, it is better to think about the behavior of the transformer.

When using the transformer there are two different cases practically. And each case requires different optimization technique. This is the key point.

By considering these two cases, we can finally select the most appropriate transformer and obtain the maximum gain.

 

 


( how to maximize the gain ?)

Let us consider about the transformer. The gain of the circuit by using the transformer is represented by

eq1.png         (1)

Where ZL is the impedance of the load (i.e. impedance of the circuit without the transformer ) and n is the turn ratio.

It is apparent that G is the function of two parameters, ZL and n.  This leads to two different solutions for maximizing the gain practically. 

 

matching_edit.png

 

  - case.1 : The turn ratio n is fixed.

In this case, the tunable parameter is the impedance ZL.  The gain as a function of ZL is shown in the left figure above.

In order to maximize the gain, Z must be as high as possible.  The gain G get close to 2n when the impedance ZL goes to infinity.

There also is another important thing; If the impedance ZL is bigger than the matched impedance (i.e. ZL = 50 * n^2 ), the gain can get higher than n.

 

  - case.2 : The impedance ZL is fixed.

In contrast to case1, once the impedance ZL is fixed, the tunable parameter is n. The gain as a function of n is shown in the right figure above.

In this case the impedance matched condition is the best solution, where ZL=50*n^2. ( indicated as red arrow in the figure )

The gain can not go higher than n somehow. This is clearly different from case1.
 

 

( Application to the triple resonant circuit )

Here we can define the goal as "all three resonances have gain of more than n, while n is set to be as high as possible"

According to consideration of case1, if each resonance has an impedance of greater than 50*n^2 (matched condition) it looks fine, but not enough in fact.

For example if we choose n=2, it corresponds to the matched impedance of 50*n^2 = 200 Ohm. Typically every three resonance has several kOhm which is clearly bigger than the matched impedance successfully.

However no matter how big impedance we try to make,  the gains can not be greater than G=2n=4 for all the three resonance. This is ridiculous.

What we have to do is to choose n so that it matches the impedance of the resonance which has the smallest impedance.

In our case, usually the resonance at 55MHz tends to have the smallest impedance in those three. According to this if we choose n correctly, the other two is mismatched.

However they can still have the gain of more than n, because their impedance is bigger than the matching impedance. This can be easily understand by recalling the case1.

 

(expected optimum gain of designed circuit)

 By using the equation (1), the expected gain of the triple resonant circuit including the losses is calculated. The parameters can be found in last entry.

designed_response.png

The turn ratio is set as n=11, which matches the impedance of the resonance at 55MHz. Therefore 55MHz has the gain of 11.

The gain at 11MHz is bigger than n=11, this corresponds to the case1. Thus the impedance at 11MHz can go close to gain of 22, if we can make the impedance much big.

 

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