I will be in the Clean and Bake Lab today from 9am to 4pm.

For the first pass, it's probably easiest to use the existing DCPD amplifier. Looking at the gain and noise performance in Attachment #1, seems totally fine, the electronics noise will not be limiting if we have ~10mW of LO power. I assumed a transimpedance resistor of 1 kohm, and all other numbers as on the schematic (though who knows if the schematic is accurate). The noise should be measured to confirm that the box is performing as expected...

Summary:

Koji and I had a discussion last Friday about the suspension electronics. I think there are still a few open questions - see Attachment #1. We should probably make a decision on these soon.

Other useful links:

1. High-voltage coil driver circuit - D1900163. 
   • This board is ready to be fabricated and tested on the bench.
   • The way the connectors J2 and J3 are designed currently is meant to interface with the existing coil driver electronics.
   • Depending on the eventual coil driver we choose for the fast path, it may be benificial to change the signals on the connectors J2 and J3, to avoid the need for a custom interface board.
2. HAM-A coil driver noise analysis.
   • The linked attachment evaluates the noise for the design value of the fast path series resistor, which is 1.2 kohms.
   • Iff we still have ambitions of measuring ponderomotive squeezing, we will need the resistance to be much higher, ~10 kohms (in the linked noise budget, only the Johnson noise of the series resistor is considered, but in reality, the OpAmp voltage and current noises also matter). 
   • This corresponds to a maximum current of 10V/10kohms = 100uA. 
   • Looking at signals to the ETMs from the current lock acquisition sequence, the RMS current to a single coil is approximately _____ (to be filled in later).
   • So we may need a version of the fast coil driver that supports a low noise mode (with large series resistance) and a high-range mode (with lower series resistance for lock acquisition).
3. You can follow the links to DCC entries for other parts from Attachment #1.

From Attachment #1, looks like the phasing and gain for CARM on POX11 is nearly the same as CARM of REFL11, which is probably why I was able to execute a partial transition last night. The response in POY11 is ~10 times greater than POX11, as expected - though the two photodiodes have similar RF transimpedance, there is a ZFL-500-HLN at the POY11 output. The actual numerical values are 2.5e10 cts/m for CARM-->REFL11_I, 2.6e10 cts/m for CARM-->POX11_I, and 3.2e11 cts/m for CARM-->POY11_I.

So I think I'll just have to fiddle around with the transition settings a little more tonight. 

One possible concern is that the POX and POY signals are digitized without preamplificatio, maybe this explains the larger uncertainty ellipse for the POX and POY photodiodes relative to the REFL11 photodiode? Maybe the high frequency noise is worse and is injecting junk in the AO path? I think it's valid to directly compare the POX and REFL spectra in Attachment #2, without correcting for any loops, because this signal is digitized from the LSC demodulator board output (not the preamplified one, which is what goes to the CM board, and hence, is suppressed by the CARM loop). Hard to be sure though, because while the heads are supposed to have similar transimpedance, and the POX photodiode has +12dB more whitening gain than REFL11, and I don't know what the relative light levels on these photodiodes are in lock.

I started my attempt on noise budgeting of ALS by going back to how Kiwamu did it and adding as many sources as I could find up till now. This calculation is present in ALS_Noise_Budget notebook. I intend to collect data for noise sources and all future work on ALS in the ALS repo.

The noise budget runs simulink through matlab.engine inside python and remaining calculations including the pygwinc ones are done in python. Please point out any errors that I might have done here. I still need to add noise due to DFD and the ADC after it. For the residual frequency noise of AUX laser, I have currently used an upper limit of 1kHz/rt Hz at 10 Hz free-running frequency noise of an NPRO laser.

I forgot about the pointing - probably we will need another actuator to control the pointing of the AS beam onto the DCPDs. I found a few old PI PZTs (model number is S-320, which is a retired part), one is labelled broken but the others don't indicate a-priori that they are broken. I'll post a more detailed hardware survey later.

You can activate all 3axis

The emergency lamps above the exit sign on the NW entrance to the control room are on. I tried opening and closing the door, but it remains on. Probably nothing to worry about, but noting here anyway.

It happened before too. Doesn't it say it has occasional self-testing or something?

True - it is now not on anymore.

To facilitate this investigation, I've DQed the 4 face coil outputs for the two ETMs. EX is currently running with 5 times the series resistance of EY, so it'll be a nice consistency check. Compilation, installation etc went smooth. But when restarting the c1scx model, there was a weird issue - the foton file, C1SCX.txt, got completely wiped (all filter coefficients were empty, even though the filter module names themselves existed). I just copied the chiara backup version, restarted the model, and all was well again.

This corresponds to 8 additional channels, recorded at 16k as float 32 numbers, so in the worst case (neglecting any clever compression algorithms), we are using disk space at a rate of ~4 MB/s more. Seems okay, but anyway, I will remove these DQ channels in a few days, once we're happy we have enough info to inform the coil driver design.

spoke too soon - there was an RFM error for the TRX channel, and restarting that model on c1sus took down all the vertex FEs. Anyways, now, things are back to normal I think. The remaining red light in c1lsc is from the DNN model not running - I forgot to remove those channels, this would've been a good chance! Anyways, given that there is an MLTI in construction, I'm removing these channels from the c1lsc model, so the next time we restart, the changes will be propagated.

For whatever reason, my usual locking scripts aren't able to get me to the PRFPMI locked state - some EPICS channel value must not have been set correctly after the model reboot 😞. I'll debug in the coming days.

Fun times lie ahead for getting the new BHD FEs installed I guess 🤡 ....

Attachment #1 - The 80mW pickoff was getting clipped on a BNC cable, and not making it to the doubling oven. 😢 .

• Since the PSL doubled beam isn't used for locking these days, I just didn't notice.
• I blame the ringdown team, this crazy tee arrangement wasn't the case before.
• I fixed the situation by changing the cabling such that the beam clears the cables comfortably.

Attachment #2 - PSL green shutter removed. Alignment into the doubling oven is extremely tedious, and so I opted to preserve the capability of recovering the green beam by simply removing a single mirror.

Attachment #3 - The beam path for coupling the LO beam into a fiber.

• Primary goal was to have easy access to some steering mirrors so that I can optimize alignment into the fiber collimator.
• I opted to use the NW corner of the PSL table - that's where most of our existing fiber hardware is anyways, and there was sufficient space and easy access over there.
• 3 Y1 mirrors were installed, using the preferred Polaris mounts and 3/4" post + baseplate hardware. They were labelled Y1-1037-45P so that future workers need not be un-necessarily tortured. The third mirror is not visible in this photograph.
• Once the collimator arrives, I will mode match this beam into the fiber. Plan is to use the fiber originally used for the mode spectroscopy project. It needs to be moved to the NW corner of the PSL table, and the other end needs to be routed to the AP table (it was brought back to the PSL table to facilitate Anjali's fiber MZ experiment). 
• There is plenty of space in the beam path for mode-matching lens(es) and polarization control optics.

Attachment #4 shows the BHD photodiodes taken from QIL. 

• Unfortunately, we could not find the readout electronics. 
• In the worst case, we can just interface these PDs with the existing Satellite box (associated with the copper OMC).
• It might be that the OMC cavity can simply be placed on this breadboard, making the whole setup nice and portable.
• We may want to consider having an OFI between the OMC and the IFO AS beam at some point...

The (masked) tech accessed all areas in the lab (office area, control room, VEA) between ~230pm-3pm. The laser safety goggles he used have been kept aside for appropriate sanitaiton.

1. I found that an EPICS channel wasn't reset to the correct value by burtrestore after the FE bootfest yesterday.
   • This cost me the whole of last night, found it finally tonight. 
   • I'll try and modify the locking scripts to better capture such errors, but ideally, we should just use Guardian or something since it's made for this purpose already.
   • Anyways, tonight I was able to re-acquire the PRFPMI lock in a completely scripted way.
2. Locking CARM on POX remains out of reach.
   • I think this has to do with the fact that the zero-crossing of the CARM and REFL error signals are dependent on the 3f PRCL/MICH error point offsets.
   • So even if the DC gain is right, the fact that we use POX for the digital AO path and REFL for the analog AO path is leading to some conflict I think.
   • Ran out of energy tonight, I'll try again tomorrow.

The DQ channels of the ETM coils were active tonight, so I'll make the coil driver actuation budget over the next couple of days.

I will be in the Clean and Bake lab today from 9am to 4pm.

Attachment #1 shows the proposed wiring and CDS topology for the in air BHD setup. The PDF document has hyperlinks you can follow to the DCC entries. Main points:

1. I think we should run the realtime model on c1lsc. This will negate the need for any IPC between c1ioo and c1lsc machines.
2. I think we have most of the electronics we need already, though I am still in the process of testing the various boards, especially the HV ones.
3. We may choose to use the switchable whitening feature for the M2 ISS board
   • This would require some BIO channels
   • There are plenty spare in c1lsc, so it's not going to be a show stopper
   • This is why I've not explicitly included a whitening board for now...
4. The main job seems to be to make a whole bunch of custom cables. For the most part, I think we have the long (~20m) long D9 cables, so I propose just snipping off the connector at one end, and soldering on the appropriate connectors to the correct conductors.
5. For the homodyne phase control - the proposal is to use a PI PZT with 3 piezoelectric elements. We would drive the 3 elements with the same voltage, by shorting the conductors together (at least that's how I understood Koji's comment), so we'd only need a single DAC channel for this purpose.
   • Need to confirm that the parallel PZT capacitances (each element is ~300 nF so 3 in parallel would be ~900 nF) still allows sufficient actuation bandwidth.
   • If the relative actuation strength of the 3 elements needs to be individually tuned, we may have to use three DAC channels. The D980323 board will allow the driving of 3 independent channels. I have one of these boards in hand, but need to check if it works, and also implement the changes outlined here.
6. The alignment control has not yet been accounted for
   • We could consider using the in-vacuum PZTs, these were verified to be working ~2018.
   • If we use only 1 steering PZT mirror, we have sufficient free DAC channels available in c1lsc. But if we need both (to avoid clipping for example), then we need more DAC channels - we can either free up one DAFI channel, or install a DAC in the c1lsc expansion chassis. 
7. We may want to expand to have a second OMC at some point. In which case we'd need, at the very least
   • 1 more DAC card
   • A HV driver for the second OMC length (could use the Trek driver if we use D980323 for the homodyne phase control).

Please comment if I've overlooked something.

Summary:

Looking at the signals to the test mass coils, it seems borderline to me that we will be able to acquire lock and run in a low noise configuration with the same series resistor in the coil driver circuit. The way I see it, options are:

1. Use a moderately high series resistance (e.g. 5 kohms) for the time being, and go ahead with the HAM-A coil driver.
   • This will mean a current noise of ~3pA/rtHz, which translates to ~3e-18 m/rtHz @ 100 Hz in DARM displacement noise (assuming the ITMs have much higher series resistance than the ETMs).
   • If the lock acquisiton looks smooth, double the resistance to 10 kohms.
   • With 5 kohm series resistance, there is negligible possibility of measuring ponderomotive squeezing for any of the input powers we consider feasible, but this is under the assumption that we will expose coil driver noise, which is very optimistic imho.
2. Re-design a new coil driver that allows switchable impedance, so we can have a higher noise acquisition mode for acquiring and holding the ALS lock, then transition to a lower noise, lower range config once the RF / BHD lock has been acquired.
   • On paper, this solves all the problems, but the design of such a circuit is probably pretty non-trivial and time consuming.

Details:

I only looked at the ETMs for this study. The assumption is that we will have no length actuation on the ITMs, only local damping and Oplev loops (and maybe some ASC actuation?), which can be sufficiently low-pass filtered such that even with coil de-whitening, we won't have any range issues.

Attachment #1 shows the time-domain traces of the coil driver signals as we transition from POX/POY lock to the ALS lock. There are some transients, but I think we will be able to hold the lock even with a 5 kohm resistor (~twice what is on ETMX right now). From just these numbers, it would seem we can even go up to 10 kohms right away and still be able to acquire lock, especially if we re-design the digital feedback loop to have better low-pass filtering of the high-frequency ALS noise, see the next attachment.

Attachment #2 shows the f-domain picture, once the arm lengths are fully under ALS control (~25 seconds onwards in Attachment #1). The RMS is dominated by high frequency ALS length loop noise, which we can possibly improve with better design of the digital control loop.

Finally, Attachment #3 shows the situation once DARM control has been transitioned over to AS55_Q. Note that the vertex DoFs are still under 3f control, so there is the possibility that we can make this even lower noise. However, one thing that is not factored in here is that we will have to de-whiten these signals to low-pass filter the DAC noise (unless there is some demonstrated clever technique with noise-mons or something to subtract the DAC noise digitally). Nevertheless, it seems like we can run safely with 5 kohms on each ETM coil and still only use ~2000 cts RMS, which is ~1/10th the DAC range (to allow for dealing with spurious transients etc). 

The LO pickoff has been coupled into a fiber with ~90% MM (8 mW / 9 mW input). While I wait for the DCPD electronics to be found in the Cryo lab, I want to monitor the stability of the pointing, polarization etc, so I'd like to clear some space on the AP table that was occupied for the mode spectroscopy project. If there are no objections before 2pm tomorrow July 21 2020, I will commence this work.

I've added 4 proposed schemes for implementing ALS in voyager. Major thing to figure out is what AUX laser would be and how we would compare the different PSL and AUX lasers to create an error signal for ALS. Everywhere below, 2um would mean wavelengths near 2 um including the proposed 2128nm. Since it is not fixed, I'm using a categorical name. Same is the case for 1um which actually would mean half of whatever 2 um carries.

Higher Harmonic Generation:

• We can follow the current system of ALS with using 1.5 times PSL frequency as AUX instead of second harmonic as 1 um is strongly absorbed in Si.
• To generate 1.5 times PSL frequency, three stages would be required.
  • SHG: Second Harmonic Generation mode matched to convert 2um to 1um. If we are instead making 2 um from 1um to start with, this stage will not be required.
  • SFG: Sum Frequency Generation mode matched to sum 2um photon and 1um photon to give 0.65 um photon.
  • DPDC: Degenerate Parametric Down Conversion mode matched to convert 0.65 um to 1.3 um (which would be 1.5 times PSL frequency).
• To compare, we can either convert pick-off from PSL to AUX frequency by doing the above 3 stages (Scheme II).
• Or we can just do SHG and SFG at PSL pick-off and do another SHG at AUX end (Scheme I) to compare the AUX and PSL both converted to 0.65 um (which would be 2 times AUX and 3 times PSL frequency).
• This method would have added noise from SHG, SFG and DPDC processes along with issues to be inefficiency of conversion.

Arbitrary AUX frequency:

• We can get away with using some standard laser near 1.5 um region directly as AUX. Most probably this would be 1550 nm.
• What's left is to devise a method of comparing 1.5 um and 2um frequencies. Following are two possible ways I could think of:

Using a frequency comb:

• Good stable frequency combs covering the wavelength region from 1.5 um to 2 um are available of the shelf.
• We would beat PSL and transmitted AUX separately with the frequency comb. The two beat note frequencies would be:
  
  
• Here, m1 and m2 represent the nearest modes (comb teeth) of frequency comb to PSL and AUX respectively.
• Carrier Envelope Offset frequency () can be easily generated by using an SHG crystal in front of the Frequency comb. This step is not really required since most of the modern frequency combs now comb with inbuilt zero stabilization.
• Mixing above beatnotes with would remove from them along with any noise associated with .
• Further, a Direct Digital Synthesis IC is required to multiply the AUX side RF signal by m1/m2. This finally makes the two RF signals to be:
  
  
• Which on mixing would give desired error signal for DFD as :
  
• This method is described in Stenger et al. PRL. 88, 073601 and is useful in comparing two different optical frequencies with a frequency comb with effective cancellation of all noise due to the frequency comb itself. Only extra noise is from the DDS IC which is minimal.
• This method, however, might be an overkill and expensive. But in case (for whatever reason) we want to send in another AUX at another frequency down the 40m cavity, this method allows the same setup to be used for multiple AUX frequencies at once.

Using a Transfer Cavity:

• We can make another more easily controlled and higher finesse cavity with a PZT actuator on one of the mirrors.
• In the schematic, I have imagined it has a triangular cavity with a back end mirror driven by PZT.
• Shining PSL from one side of the transfer cavity and employing the usual PDH, we can lock the cavity to PSL.
• This lock would require to be strong and wide bandwidth. If PZT can't provide enough bandwidth, we can also put an EOM inside the cavity! (See this poster from Simon group at UChicago)
• Another laser at AUX frequency, called AUX2 would be sent from the other side of the cavity and usual PDH is employed to lock AUX2 to the transfer cavity.
• So clearly, this cavity also requires coatings and coarse length such that it is resonant with both PSL and AUX frequencies simultaneously.
• And, the FSS for AUX2 needs to be good and high bandwidth as well.
• The transmitted AUX2 from the transfer cavity now would carry stability of PSL at the frequency of AUX and can be directly beaten with transmitted AUX from the 40m cavity to generate an error signal for DFD.
• I believe this is a more doable and cheaper option. Even if we want to do a frequency comb scheme, this could be a precursor to it.

_________________________

EditTue Jul 21 17:24:09 2020: (Jamie's suggestion)

Using Mode Cleaner cavity as Transfer Cavity:

• If we coat the mode cleaner cavity mirrors appropriately, we can use it to lock AUX2 laser (mentioned above).
• This will get rid of all extra optics. The only requirement is for FSS to be good on AUX2 to transfer PSL (MC) stability to AUX frequency.
• I've added suggested schematic for this scheme at the bottom.

Attachment #1 shows the RIN of the local oscillator beam delivered to the AP table via fiber. I used a PDA520 to make this measurement, while the electronics for the DCPDs are pending. I don't really have an explanation for the difference between the locked IFO trace vs the not locked trace - we don't have an ISS running (but this first test suggests we should) and the beam is picked off before any cavities etc, so this is a reflection of the state of the FSS servo at the times of measurement?

Tried locking CARM using the hybrid REFL (for AO path) and POX 11 (for MCL path) scheme a bunch of times today, but I had no luck. When the CARM offset is zeroed, the PRMI lock is lost almost immediately. Maybe this is indicative of some excess noise in the POX data stream relative to the REFL signal? The one thing I haven't tried is to take the IFO all the way to the locked state, and then transition the MCL actuation from CM_SLOW to POX11_I.

An SR785 is sitting on the North side of the AP table in the walkway - I will clear it tomorrow.

I decided to use the old EY auxiliary optics table, which is now stored along the east arm about 10 m from the end, as a workspace for assembling the little PMCs. I wiped everything down with isopropanol for general cleanliness, removed the metal plate on the south edge of the table enclosure to allow access, covered the table with some clean Aluminium foil, and then moved the plastic box with PMC parts to the table - see Attachment #1. I haven't actually done any assembly just yet, waiting for more info (if available) on the procedure and implements available...

This year we've struggled with vacuum controls unreliability (e.g., spurious interlock triggers) caused by decaying hardware. Here are details of the vacuum refurbishment plan I described on the 40m call this week.

☑ Refurbish TP2 and TP3 dry pumps. Completed [ELOG 15417].

☑ Automated notifications of interlock-trigger events. Email to 40m list and a new interlock flag channel. Completed [ELOG 15424].

☐ Replace failing UPS.

• Two new Tripp Lite units on order, 110V and 230V [ELOG 15465].
• Jordan will install them in the vacuum rack once received.
• Once installed, Jon will come test the new units, set up communications, and integrate them into the interlock system following this plan [ELOG 15446].
• Jon will move the pumps and other equipment to the new UPS units only after completing the above step.

☐ Remove interlock dependencies on TP2/TP3 serial readbacks. Due to persistent glitching [ELOG 15140, ELOG 15392].

Unlike TP2 and TP3, the TP1 readbacks are real analog signals routed to Acromags. As these have caused us no issues at all, the plan is to eliminate dependence on the TP2/3 digital readbacks in favor of the analog controller outputs. All the digital readback channels will continue to exist, but the interlock system will no longer depend on them. This will require adding 2 new sinking BI channels each for TP2 and TP3 (for a total of 4 new channels). We have 8 open Acromag XT1111 channels in the c1vac system [ELOG 14493], so the new channels can be accommodated. The below table summarizes the proposed changes.

I installed the Tripp Lite SMX1000RT2U and Tripp Lite Smart1000LCD at the bottom of the 1x8 electronics rack. These are plugged in to power, and are ready for testing. All other cables (serial, usb, etc.) have been left on the table next to the 1x8 rack.

To carry out the next steps of the vac refurbishment plan [ELOG 15499], I've ordered parts necessary for interfacing the UPS units and the analog TP2/3 controller outputs with c1vac. The purchase list is appended to the main BHD list and is located here. Some parts we already had in the boxes of Acromag materials. Jordan is gathering what we do already have and staging it on the vacuum controls console table - please don't move them or put them away.

This afternoon Jordan is going to carry out a test of the V4 and V5 hardware interlocks. To inform the interlock improvement plan [15499], we need to characterize exactly how these work (they pre-date the 2018 upgrade). I have provided him a sequence of steps for each test and will also be backing him up on Zoom.

We will close V1 as a precaution but there should be no other impact to the IFO. The tests are expected to take <1 hour. We will advise when they are completed.

We explore bilinear SRCL to DARM noise coupling mechanisms, and show two cases that by doing BHD readout the noise performance can be improved. In the first case, the bilinear piece is due to residual DHARD motion (see also LHO:45823), and it matters mostly for the low-frequency (<100 Hz) part, and in the second piece the bilinear piece is due to residual SRCL fluctuation and it matters mostly for the a few x 100 Hz part. Details are below:

=================================================

General Model:

We can write the SRCL to DARM transfer function as (Evan Hall's thesis, eq. 2.29)

Z_s2d(f) = C_lf(f) * F^2 * x_D + C_hf(f) * F * dphi_S * x_D    ---- (1)

where

C_lf ~ 1/f^2 and C_hf ~ f are constants at each frequency unless there are major upgrades to the IFO,

F is the finesse of the arm cavity which depends on the alignment, spot position on the TMs, etc., 

dphi_S is the SRCL detuning (wrt the nominal 90 deg value), 

x_D is the DC DARM offset. 

The linear part of this can be removed with feedforward subtractions and it is the bilinear piece that matters, which reads

dZ_s2d = C_lf * <F>^2 * dx_D + C_hf * <F> * <dphi_S> * dx_D

             + 2C_lf * <F> * <x_D>  * dF + C_hf * <dphi_S> * <x_D> * dF

             + C_hf  * <F> * <x_D> * d(dphi_S).     ---- (2)

The first term in (2) is due to residual DARM motion dx_D. This term does not depends on the DC value of DARM offset <x_D> and thus does not depend on doing BHD or DC readout. On the other hand, the typical residual DARM motion is 1 fm << 1 pm of DARM offset. Since the current feedforward reduction factor is about 10 (see both Den Martynov's thesis and Evan Hall's thesis), clearly we are not limited by the residual DARM motion. 

The second term is due to the change in the arm finesse, which can be affected by, e.g., the alignment fluctuation (both increasing the loss due to scattering into 01/10 modes and affecting the spot positon and hence changing the losses), and is likely to be the reason why we see the effect being modulated by DHARD. 

The last term in (2) is due to the residual SRCL fluctuation and is important for the ~ a few x 100 Hz band.

=================================================

DHARD effects. 

As argued above, the DHARD affects the SRCL -> DARM coupling as it changes the finesse in the arm cavity (through scattering into 01/10 modes; in finesse we cannot directly simulate the effects due to spot hitting a rougher location). 

Since in the second term of eq. (2) the LF part depends on the DARM DC offset <x_D>, this effect can be improved by going from DC readout to BHD. 

To simulate it in finesse, at a fixed DARM DC offset, we compute the SRCL->DARM transfer functions at different DHARD offsets, and then numerically compute the derivative \partial Z_s2d / \partial \theta_{DH}. Then multiplying this derivative with the rms value of DHARD fluctuation \theta_{DH} we then know the expected bilinear coupling piece. 

The result is shown in the first attached plot. Here we have assumed a flat SRCL noise of 5e-16 m/rtHz for simplicity (see PRD 93, 112004, 2016). We do not account for the loop effects which further reduces the high frequency components for now. The residual DHARD RMS is assumed to be 1 nrad. 

In the first plot, from top to bottom we show the SRCL noise projection at different DARM DC offsets of (0.1, 1, 10) pm. Since the DHARD alignment only affects the arm finesse starting at quadratic order, it thus matters what DC offset in DHARD we assume. In each pannel, the blue trace is for no DC offset in DHARD and the orange one for a 5 nrad DC offset. As a reference, the A+ sensitivity is shown in grey trace in each plot as a reference. 

We can see if there is a large DC offset in DHARD (a few nrad) and we still do DC readout with a few pm of DARM offset, then the bilinear piece of SRCL can still contaminate the sensitivity in the 10-100 Hz band (bottom panel; orange trace). On the other hand, if we do BHD, then the SRCL noise should be down by ~ x100  even compared to with the top panel. 

(A 5 nrad of DC offset in DHARD coupled with 1 nrad RMS would cause about 0.5% RIN in the arms. This is somewhat greater than the typically measured RIN which is more like <~ 0.2%. See the second plot). 

=================================================

SRCL effect. 

Similarly we can consider the SRCL->DARM coupling due to residual SRCL rms. The approach is very similar to what we did above for DHARD. I.e., we compute Z_s2d at fixed DARM offset and for different SRCL offsets, then we numerically evaluate \partial Z_s2d / \partial dphi_S. A residual SRCL rms of 0.1 nm is then used to generate the projection shown in the third figure. 

Unlike the DHARD effect, the bilinear SRCL piece does not depend on the DC SRCL detuning (for the 50-500 Hz part). It does still depends on the DARM DC offset and therefore could be improved by BHD.

Since we do not include the LP of the SRCL loop in this plot, the HF noise at 1 kHz is artifical as it can be easily filtered out. However, the LP will not be very strong around 100-300 Hz for a SRCL UGF ~ 30 Hz, and thus doing BHD could still have some small improvements for this effect. 

This test has been completed. The IFO configuration has been reverted to nominal.

For future reference: yes, both the V4 and V5 hardware interlocks were found to still be connected and work. A TTL signal from the analog output port of each pump controller (TP2 and TP3) is connected to an auxiliary relay inside the main valve relay box. These serve the purpose of interupting the (Acromag) control signal to the primary V4/5 relay. This interrupt is triggered by each pump's R1 setpoint signal, which is programmed to go low when the rotation speed falls below 80% of the low-speed setting.

3. I agree - this whitening will be handy to have for diagnostics.

4. I think in principle, we can ask a company to make the custom cables for us to save us some hand labor. Rich/Chub probably know the right companies to do small numbers of dirty cables.

5. Can't we just a single Noliac PZT in the same way that the OMC does? Or is the lead time too long?

6. Do we need active steering for this in-air test? I'm not even sure how we would get the alignment signal, so maybe that's a good reason to figure this out.

This earthquake and friends had tripped all watchdogs. I used the scripted watchdog re-enabler, and released the stuck ITMX (this operation is still requires a human and hasn't been scripted yet). IMC is locked again and all Oplevs report healthy optic alignment.

I've pushed an MCMC simulation to the A+ BHD repo (filename MCMC_TFs.ipynb). The idea is to show how random offsets around ideal IFO change the noise couplings of different DOFs to readout.

At each step of the simulation:

1. Random offsets for the different DOFs are generated from a normal distribution. The RMSs are taken from experimental data and some guesses and can be changed later. The laser frequency is tuned to match the CARM offset.

These are the current RMS detunings I use:

2. A transfer function is computed for the noisy DOFs.

3. Projected noise is calculated.

These are the noise level for the DOFs:

The attachments show the projected noise levels for the noisy DOFs. Each curve is a different instance of random offsets. The ideal case - "zero offsets" is also shown.

OMC Comm and OMC diff refer to the common and differential length change of the OMCs.

Be aware that there is now a KEPCO HV supply that is energized, sitting on the floor immediately adjacent to the OMC rack, east of the AP table. It is currently set to 100 V DC, and a PI PZT installed on the AP table has its 3 PZTs energized by said supply (via an OMC piezo driver). I will post pictures etc of the work from the last 10 days over the weekend.

that's great. I think we would like to figure out how to present this so that its clear what the distribution of TFs is. Maybe we can plot the most likely curve as well as a shaded region indicating the 5% and 95% values?

and then we add the loops

Summary : 

I have been working on analyzing the seismic data obtained from the 3 seismometers present in the lab. I noticed while looking at the combined time series and the gain plots of the 3 seismometers that there is some error in the calibration of the BS seismometer. The EX and the EY seismometers seem to be well-calibrated as opposed to the BS seismometer.

The calibration factors have been determined to be :

BS-X Channel: 

BS-Y Channel: 

BS-Z Channel:

Details :

The seismometers each have 3 channels i.e X, Y, and Z for measuring the displacements in all the 3 directions. The X channels of the three seismometers should more or less be coherent in the absence of any seismic excitation with the gain amongst all the similar channels being 1. So is the case with the Y and Z channels. After analyzing multiple datasets, it was observed that the values of all the three channels of the BS seismometer differed very significantly from their corresponding channels in the EX and the EY seismometers and they were not calibrated in the region that they were found to be coherent as well. 

Method :

Note: All the frequency domain plots that have been calculated are for a sampling rate of 32 Hz. The plots were found to be extremely coherent in a certain frequency range i.e ~0.1 Hz to 2 Hz so this frequency range is used to understand the relative calibration errors. The spread around the function is because of the error caused by coherence values differing from unity and the averages performed for the Welch function. 9 averages have been performed for the following analysis keeping in mind the needed frequency resolution(~0.01Hz) and the accuracy of the power calculated at every frequency. 

1. I first analyzed the regions in which the similar channels were found to be coherent to have a proper gain analysis. The EY seismometer was found to be the most stable one so it has been used as a reference. I saw the coherence between similar channels of the 2 seismometers and the bode plots together. A transfer function estimator was used to analyze the relative calibration in between all 3 pairs of seismometers. In the given frequency range EX and EY have a gain of 1 so their relative calibration is proper. The relative calibration in between the BS and the EY seismometers is not proper as the resultant gain is not 1. The attached plots show the discrepancies clearly : 

• BS-X & EY-X Transfer Function : Attachment #1
• BS-Y & EY-Y Transfer Function : Attachment #2

          The gain in the given frequency range is ~3. The phase plotting also shows a 180-degree phase as opposed to 0 so a negative sign would also be required in the calibration factor. Thus the calibration factor for the Y channel of the BS seismometer should be around ~3. 

• BS-Z & EY-Z Transfer Function : Attachment #3

The mean value of the gain in the given frequency range is the desired calibration factor and the error would be the mean of the error for the gain dataset chosen which is caused due to factors mentioned above.

Note: The standard error envelope plotted in the attached graphs is calculated as follows :

         1. Divide the data into n segments according to the resolution wanted for the Welch averaging to be performed later. 

         2. Calculate PSD for every segment (no averaging).

         3. Calculate the standard error for every value in the data segment by looking at distribution formed by the n number values we obtain by taking that respective value from every segment.

Discussions :

The BS seismometer is a different model than the EX and the EY seismometers which might be a major cause as to why we need special calibration for the BS seismometer while EX and EY are fine. The sign flip in the BS-Y seismometer may cause a lot of errors in future data acquisitions. The time series plots in Attachment #4 shows an evident DC offset present in the data. All of the information mentioned above indicates that there is some electrical or mechanical defect present in the seismometer and may require a reset. Kindly let me know if and when the seismometer is reset so that I can calibrate it again. 

I fixed some stuff in the MCMC simulation:

1. Results are now plotted as shades from minimum to maximum. I tried making the shade the STD around a mean but it doesn't look good on a log scale when the STD is bigger than the mean.

2. Added comparison with aLigo. The OMCL diff and comm motions in A+ are both compared to the single OMCL DOF of aLigo.

3. I fixed a serious error in the code that produced incorrect results.

4. Imbalances in the IFO such as differential arm loss are generated randomly at the beginning and stay fixed for the rest of the simulation instead of being treated as an offset.

5. The simulation now runs with maxtem=2. That is, TEM modes up to 2nd order are considered.

The results are attached.

All the details are in E2000436, and documents linked from there, I think an elog would be much too verbose. In summary, a workable setup consisting of

• 2 DCPDs interfaced with the realtime CDS system. Note that because this circuit is single-ended, while the AA and ADC are differential receiving, there is an overall gain of 0.5. Explicitly, for the 300 ohm DC transimpedance, the conversion is ~350 cts/mW.
• A local oscillator beam delivered via fiber that is mode-matched (roughly) with the IFO AS beam.
• A PZT mounted mirror to control the homodyne phase. The PZT (S320) is an obsolete part and it's hard to find a datasheet for it, but if its specs are comparable to the more modern S330, the full stroke is 10 um, for a max applied voltage of 100 V DC, so 100nm/V. c.f. 200V for 3um full stroke of the Noliac.

was prepared.

Last night, I locked the PRMI with the carrier resonant, and convinced myself that the DCPD null stream was sensing the MICH degree of freedom (while it was locked on AS55_Q) with good SNR below ~60 Hz. Above ~60 Hz, in this configuration, the ADC noise was dominating, but by next week, I'll have a whitening board installed that will solve this particular issue. With the optical gain of MICH in this configuration, the ADC noise level was equivalent to ~500 nrad/rtHz of phase noise above ~60 Hz (plots later).

Now, I can think about how to commission this setup interferometrically.

Some tests done today:

All of these tests were done with the PRMI locked with carrier resonant in the recycling cavity (i.e. sidebands rejected to REFL port). I then actuated the BS length DOF with a sine wave at 311.1 Hz, 40 cts amplitude (corresponding to ~8 pm of peak-to-peak displacement).

1. Attempt to balance the DCPDs
   • I tried to tune the digital gains of the two DCPDs so as to minimize the appearance of this line in the SUM channel
   • but no matter how I tuned the gains, I couldn't make the line in the SUM channel disappear entirely - in fact, the best I could do was to make the line height in SUM and NULL channels (yes I recognize the poor channel name choice, I'll change "NULL" to "DIFF" at the next model recompile) the same. See Attachment #1.
   • The lobes around the main peak are indicative of some scattering?
   • Attachment #2 shows a wider frequency range. The homodyne phase isn't controlled, so the "NULL" channel is not necessarily measuring the correct quadrature to be sensing MICH motion.
   • I think I can back out something about the contrast defect from this fact, but I need to go back to some modeling.
2. A simple test of the homodyne phase actuator
   • I wanted to check that this PI S320 piezo actually allows me to actuate the optical path length of the local oscillator.
   • I'm using the OMC HV driver to drive said PZT - so there are two DAC channels available, one to dither the optic and one to apply a control signal. I think mainly this is to avoid using up DAC range for the dither signal, the overall dynamic range is still limited by the HV supply.
   • I can't find the maximum voltage that can be applied on the datasheet - so conservatively, I limited the HV output to saturate at 100 V DC, as this is the maximum for the S330 piezos used for green steering, for which there is a manual.
   • The S320 manual does say the full stroke of each PZT element is 10 um - so the actuation coefficient is ~100 nm/V. I then drove this actuator with a sine wave of 500 cts amplitude, at 314.1 Hz (corresponding to 15 nm of motion). With only the LO beam incident on the PDs, I saw no signal in either DCPD - as expected, so this was good.
   • Then, with the PRMI locked, I repeated the test. If there is no DC light field (as expected for the PRMI in this configuration), I wouldn't expect this drive signal to show up in the DCPDs. But in fact, I do. Again, this supports the presence of some (for now unquantified) contrast defect.

While it would seem from these graphs that the RIN of the LO beam at these frequencies is rather high, it is because of the ADC noise. More whitening (to be installed in the coming days) will allow us to get a better estimate, should be ~1e-6 I think.

I was just playing today, still need to setup some more screens, DTT templates etc to do more tests in a convenient way.

See Attachment #1. J8 was connected to a "LASTI timing slave" sitting in the rack that Chiara lives in - we don't use this for anything and I confirmed that there was no effect on the RTCDS when I pulled that fiber out. The LASTI timing slave also had a blinky that was blinking when the fiber was plugged in - which I take to believe that the slot works. 

Can we get away with just using these two available slots, J8 and J13? Do we really need three new expansion chassis?

See Attachments #1 and #2. We don't have any Q3000 QPDs in hand, at least not in the photodiode box stored in the clean optics cabinet at the south end. I also checked a cabinet along the east arm where we store some photodiodes - but didn't find any there either. The only QPDs we have in hand are the YAG-444-4AH, which I believe is what is used in the iLIGO WFS heads.

So how many do we want to get?

The "source" output of the SR785 has a DC offset of -6.66 V. I couldn't make this up.

Upshot is, this SR785 is basically not usable for TF measurements. I was using the unit to characterize the newly stuffed ISC whitening board. The initial set of measurements were sensible, and at some point, I started getting garbage data. Unclear what the cause of this is. AFAIK, we don't have any knob to tune the offset - adjusting the "offset" in the source menu, I can change the level of the offset, but only by ~1 V even if I apply an offset of 10 V. I also tried connecting the ground connection on the rear of the SR785 to the bench power supply ground, no change.

Do we have to send this in for repair?

I believe we will use two new chassis at most. We'll replace c1ioo from Sun to Supermicro, but we recycle the existing timing system.

Grrr. Let's repair the unit. Let's get a help from Chub & Jordan.

Do you have a second unit in the lab to survive for a while?

When I tested Q3000 for aLIGO, the failure rate was pretty high. Let's get 10pcs.

That's great. I wonder if we can also get away with not adding new Dolphin infrastructure. I'd really like to avoid changing any IPC drivers.

The new dolphin eventually helps us. But the installation is an invasive change to the existing system and should be done at the installation stage of the 40m BHD.

There was a power outage ~30 mins ago that knocked out CDS, PSL etc. The lights in the office area also flickered briefly. Working on recovery now. The elog was also down (since nodus presumably rebooted), I restarted the service just now. Vacuum status seems okay, even though the status string reads "Unrecognized".

The recovery was complete at 1830 local time. Curiously, the EX NPRO and the doubling oven temp controllers stayed on, usually they are taken out as well. Also, all the slow machines and associated Acromag crates survived. I guess the interruption was so fleeting that some devices survived.

The control room workstation, zita, which is responsible for the IFO status StripTool display on the large TV screen, has some display driver issues I think - it crashed twice when I tried to change the default display arrangement (large TV + small monitor). It also wants to update to Ubuntu 18.04 LTS, but I decided not to for the time being (it is running Ubuntu 16.04 LTS). Anyways, after a couple of power cycles, the wall StripTools are up once again.

I added the EPCIS channels for the c1omc model (gains, matrix elements etc) to the autoburt such that we have a record of these, since we expect these models to be running somewhat regularly now, and I also expect many CDS crashes.

That's great news we won't have to worry about a new timing fanout for the two new machines, c1bhd and c1sus2. And there's no plan to change Dolphin IPC drivers. The plan is only to install the same (older) version of the driver on the two new machines and plug into free slots in the existing switch.

The vac system is going down now for planned repairs [ELOG 15499]. It will likely take most of the day. Will advise when it's back up.

Vacuum work is completed. The TP2 and TP3 interlocks have been overhauled as proposed in ELOG 15499 and seem to be performing reliably. We're now back in the nominal system state, with TP2 again backing for TP1 and TP3 pumping the annuli. I'll post the full implementation details in the morning.

I did not get to setting up the new UPS units. That will have to be scheduled for another day.

Summary

Yesterday I completed the switchover of small turbo pump interlocks as proposed in ELOG 15499. This overhaul altogether eliminates the dependency on RS232 readbacks, which had become unreliable (glitchy) in both controllers. In their place, the V4(5) valve-close interlocks are now predicated on an analog controller output whose voltage goes high when the rotation speed is >= 80% of the nominal setpoint. The critical speed is 52.8 krpm for TP2 and 40 krpm for TP3. There already exist hardware interlocks of V4(5) using the same signals, which I have also tested.

Interlock signal

Unlike the TP1 controller, which exposes simple relays whose open/closed states are sensed by Acromags, what the TP2(3) controllers output is an energized 24V signal for controlling such a relay (output circuit pictured below). I hadn't appreciated this difference and it cost me time yesterday. The ultimate solution was to route the signals through a set of new 24V Phoenix Contact relays installed inside the Acromag chassis. However, this required removing the chassis from the rack and bringing it to the electronics bench (rather than doing the work in situ, as I had planned). The relays are mounted to the second DIN rail opposite the Acromags. Each TP2(3) signal controls the state of a relay, which in turn is sensed using an Acromag XT1111.

Signal routing

The TP2(3) "normal-speed" signals are already in use by hardware interlocks of V4(5). Each signal is routed into the main AC relay box, where it controls an "interrupter" relay through which the Acromag control signal for the main V4(5) relay is passed. These signals are now shared with the digital controls system using a passive DB15 Y-splitter. The signal routing is shown below.

Interlock conditions

The new turbo-pump-related interlock conditions and their channel predicates are listed below. The full up-to-date channel list and wiring assignments for c1vac are maintained here.

There are two new channels, both of which provide a binary indication of whether the pump speed is outside its nominal range. I did not have enough 24V relays to also add the C1:Vac-TP2(3)_fail channels listed in ELOG 15499. However, these signals are redundant with the existing interlocks, and the existing serial "Status" readback will already print failure messages to the MEDM screens. All of the TP2(3) serial readback channels remain, which monitor voltage, current, operational status, and temperature. The pump on/off and low-speed mode on/off controls remain implemented with serial signals as well.

The new analog readbacks have been added to the MEDM controls screens, circled below:

Other incidental repairs

• I replaced the (dead) LED monitor at the vac controls console. In the process of finding a replacement, I came across another dead spare monitor as well. Both have been labeled "DEAD" and moved to Jordan's desk for disposal.
• I found the current TP3 Varian V70D controller to be just as glitchy in the analog outputs as well. That likely indicates there is a problem with the microprocessor itself, not just the serial communications card as I thought might be the case. I replaced the controller with the spare unit which was mounted right next to it in the rack [ELOG 13143]. The new unit has not glitched since the time I installed it around 10 pm last night.