I recalibrated the QPD today as I had shifted its position a little. I then identified the linear range of the QPD and performed a preliminary calibration of the Piezo tip-tilt within this range.
-I recalibrated the QPD as I had shifted it around a little in order to see if I could move it to a position such that I could get the full dynamic range of the piezo tilt within the linear regime of the QPD. This proved difficult because there are two reflections from the mirror (seeing as it is AR coated for 532nm and I am using a red laser). At a larger separation, these diverge and the stray spot does not bother me. But it does become a problem when I move the QPD closer to the mirror (in an effort to cut down the range in which the spot on the QPD moves). In any case, I had moved the QPD till it was practically touching the mirror, and even then, could not get the spot motion over the full range of the PZTs motion to stay within the QPD's linear regime (as verified by applying a 20Vpp 1Hz sine wave to the PZT driver board and looking at the X and Y outputs from the QPD amplifier.
-So I reverted to a configuration in which the QPD was ~40cm away from the mirror (measured using a measuring tape).
-The new calibration constants are as follows (see attached plots):
X-Coordinate: -3.43 V/mm
Y-Coordinate: -3.41 V/mm
-I then determined the linear range of the QPD to be when the output was in the range [-0.5V 0.5V].
-Next, at Jenne's suggestion, I decided to do a preliminary calibration of the PZT within this linear range. I used an SR function generator to supply an input voltage to the PZT driver board's input (connected to Channel 1 of the piezo). In order to supply a DC voltage, I set a DC offset, and set the signal amplitude to 0V. I then noted the X and Y-coordinate outputs, being sure to run through the input voltages in a cyclic fashion as one would expect some hysteresis.
-I did this for both the pitch and yaw inputs, but have only superficially analysed the latter case (I will put up results for the former later).
-There is indeed some hysteresis, though the tilt seems to vary linearly with the input voltage. I have not yet included a calibration constant as I wish to perform this calibration over the entire dynamic range of the PZT.
-There is some residual coupling between the pitch and yaw motion of the tip tilt, possibly due to its imperfect orientation in the holder (I have yet to account for the QPD's tilt).
-I have not included a graphical representation here, but there is significantly more pitch to yaw coupling when my input signal is applied to the tip-tilts pitch input (Channel 2), as compared to when it is input to channel 1. It is not clear to me why this is so.
-I have to think of some smart way of calibrating the PZT over its entire range of motion, keeping the spot in the QPD's linear regime throughout. One idea is to start at one extreme (say with input voltage -10V), and then perform the calibration, re-centering the spot to 0 on the QPD each time the QPD amp output reaches the end of its linear regime. I am not sure if this will work, but it is worth a shot. The other option is to replace the red laser with a green laser (from one of the laser pointers) in the hope that multiple reflections will be avoided from the mirror. Then I will have to recalibrate the set up, and see if I can get the QPD close enough to the mirror such that the spot stays within the linear regime of the QPD. More investigation needs to be done.
QPD Calibration Plots:
Piezo tilt vs input voltage plots:
Yaw Tilt Pitch Tilt
In light of recent events and the decision to test the piezo tip-tilts for green beam steering on the X-end table, I have set up 8 excitation points to channels 8 through 15 of the DAC on c1scx (as was done earlier for the DAC at 1Y4 with Jenne's help) in order to verify that the pin-outs of the DAC interface board. I have not yet compiled the model or restarted the computer, and will do these tomorrow, after which I will do the test. The channels are named YYY_CHAN9 etc.
I just compiled and installed the model with the excitation points on c1scx and then restarted framebuilder. The channels I set up are now showing up in the awggui dropdown menu. I will do the tests on the DAC channels shortly.
Just to keep things on record, these are the steps I followed:
I just finished carrying out the same checks for the DAC at 1X9 (with channels 9 through 16 that are unused as of now) as those I had done for the DAC at 1Y4, as the hardware prep up till now was done with the characterisation of the DAC at 1Y4. Conclusions:
I will now proceed to install various pieces of hardware (AI Board, PZT driver board, HV Power Supply and cabling) at 1X9, while not making the connection to the PZTs till I receive the go ahead.
Given that the green beam is to be used as the reference during the vent, it was decided to first test the PZT mounted mirrors at the X-endtable rather than the Y-endtable as originally planned. Yesterday, I prepared a second PZT mounted mirror, completed the full range calibration, and with Manasa, installed the mirrors on the X-endtable as mentioned in this elog. The calibration constants have been determined to be (see attached plots for aproximate range of actuation):
M1-pitch: 0.1106 mrad/V
M1-yaw: 0.143 mrad/V
M2-pitch: 0.197 mrad/V
M2-yaw: 0.27 mrad/V
Second 2-inch mirror glued to tip-tilt and mounted:
Full range calibration of PZT:
Having prepared the two steering mirrors, I calibrated them for the full range of input voltages, to get a rough idea of whether the tilt varied linearly and also the range of actuation.
Analysis and remarks:
PZT Calibration Plots
The circles are datapoints for the degree of freedom to which the input is applied, while the 'x's are for the other degree of freedom. Different colours correspond to data measured with the position of the translational stage at some value.
M1 Pitch M1 Yaw
M2 Pitch M2 Yaw
Installation of the mirrors at the X-endtable:
The calibrated mirrors were taken to the X-endtable for installation. The steering mirrors in place were swapped out for the PZT mounted pair. Manasa managed (after considerable tweaking) to mode-match the green beam to the cavity with the new steering mirror configuration. In order to fine tune the alignment, Koji moved ITMx and ETMx in pitch and yaw so as to maximise green TRX. We then got an idea of which way the input pointing had to be moved in order to maximise the green transmission.
I have updated the schematic of the D980323 PZT driver boards to reflect the changes made. The following changes were made (highlighted in red on the schematic):
I have also changed the routing of the 100V from the HV power supply onto the board, it is now done using an SMA T-connector and two short lengths of RG58 cable with SMA connectors crimped on.
The boards are functional (output swings between 0 and 100V as verified with a multimeter for input voltages in the range -10V to +10V applied using a function generator.
The following hardware has been installed on rack 1X9;
I have also verified that the AI board is functional in the eurocrate by using the LEMO monitoring points on the front panel.
The driver boards remain to be verified, but this cannot be done until we connect the HV supply to the board.
The signal chain from the DAC output to the output of the PZT driver board (including the HV supply) has been verified.
I had installed the two boards in the eurocrate yesterday and laid out the cables from 1X9 to the endtable. The output of the AI board had been verified using the monitor port on the front panel, but the output from the PZT driver board was yet to be checked because I had not connected the HV supply yesterday.
When I tried this initially today, I was not getting the expected output from the monitor channels on the front panel of the PZT driver board, even though the board was verified to be working. Alex helped debug the problem, which was identified as the -15V supply voltage not making it onto the board.
I changed the slot the board was sitting in, and used a long screw to bolt the board to the crate. Both the AI board and the PZT driver board seem to be slightly odd-sized, and hence, will not work unless firmly pushed into the eurocrate and bolted down. This would be the first thing to check if a problem is detected with this system.
In any case, I have bolted both boards to the eurocrate, and the output from the PZT driver board is as expected when I sent a 10Vp sine wave out from the DAC. I think the cables can now be hooked up to the PZTs once we are pumped down.
I have glued a fourth mirror to a PZT (using superglue) and inserted it into a modified mount. This is to be used together with the 1-inch Laseroptik mirror I had glued a couple of weeks back at the Y-endtable. I will be calibrating both these mirrors tonight such that these are ready to put in as soon as we are pumped down.
The mirror was one of those removed from the X-endtable during the switch of the steering mirrors. It is a CVI 2-inch mirror, with HR and AR coatings for 532 nm.
I have made a new model for the endtable PZT servo, and have put it in c1iscex. Model name is c1asx. Yesterday, Koji helped me start the model up. The model seems to be running fine now (there were some problems initially, I will post a more detailed elog about this in a bit) but some channels, which are computer generated, don't seem to exist (they show up as white blocks on the MEDM GDS_TP screen). I am attaching a screenshot of the said screen and the names of the channels. More detailed elog about what was done in making the model to follow.
C1:DAQ-DC0_C1ASX_STATUS (this is the channel name for the two leftmost white blocks)
These are roughly the steps I followed in setting up the new model for the endtable PZT servo - C1ASX.
I made a SIMULINK model of the servo, using MATLAB R2013a. The path to the model is /opt/rtcds/caltech/c1/userapps/release/isc/c1/models/c1asx.mdl. I am listing the parameters set on the CDS_PARAMETERS block:
Making, Compiling and Installing the Model:
After saving the model, I ssh-ed into c1iscex and ran the following commands:
rtcds make c1asx - this gave me a whole bunch of errors initially, which I tracked down to a naming problem in some of the from and goto flags: there should not be any spaces.
rtcds install c1asx
rtcds start c1asx - this gave me an error which said something like 'can't start/stop model.' Koji pointed out that given that a new model is being started, there is an additional step involved, which is to add the model name to the rtsystab file (this is located at /diskless/root/etc/rtsystab on framebuilder, and is mirrored in the various computers. It would be advisable to make sure that the changes are mirrored in the corresponding file on the computer in which the new model is being installed).
After adding the model name to the rtsystab file, I tried running rtcds start c1asx again. This time, no errors were output, but the model was not up and running as verified by looking at the C1:ASX_GDS_TP medm screen.
Koji suggested making a simple model (1 CDS parameters block, 1 ADC block and 2 filter modules, appropriately terminated) and see if that starts up, which it did. I then tried adding my servo minus the DAC block and recompiled and restarted the model. This too worked fine. I figured that the next logical step would be to add the DAC block to the model, and restart the model. But when I tried this, c1iscex crashed .
Jenne helped in restoring things to a working state (we reverted the c1asx model to just 2 filter modules, and went to the X-end and restarted the computer. This did not work the first time so I went back in and restarted it again, at which point we were able to ssh into c1iscex again and restart the four models running on it).
Since Manasa and Koji were working on getting things set up for the pumpdown,I did not try anything again till later in the evening, when Koji helped in debugging the problem further. In the meantime, at Jenne' suggestion, I made the model once again in MATLAB R2010b. In the evening, when I tried restarting the model, Koji suggested that the DAC channels in c1asx may be used by other models, at which point I realised I had set up excitation points on channels 8 through 15 of the DAC in c1scx (detailed here) in order to test the hardware at 1X9. I removed the excitation points from channels 8-11 of the DAC block in c1scx (these are the ones used in c1asx), and recompiled and restarted c1asx (using the above sequence of commands). I then tried recompiling and starting c1asx once more, and this time, it worked . At least, the GDS_TP screen suggests that the model is running alright, except for the fact that some computer generated channels seem to be missing. This problem is unresolved for now, and probably has something to do with the fact that C1ASX channels do not appear in Dataviewer.
I do not think this has to do with restarting framebuilder (I did the usual telnel fb 8088 followed by shutdown). In any case, I have added the new model to the CDS_FE_STATUS screen, and will continue to debug the same. I have also got a template medm screen (work in progress) which I will elog about soon as I get it done.
Note to self: There are 4 more excitation channels still hooked up to the DAC (channels 12-15) in the c1scx model. I plan to remove these and put them in c1asx.
I don't know what's going on here (why the channels are white), and I don't yet have a suggestion of where to look to fix it but...
Is there a reason that you're making a new model for this? You could just use and existing model at c1iscex, like the c1scx, and put your stuff in a top-names block. Then you wouldn't have to worry about all of the issues with adding and integrating a new model.
Koji just fixed this.
It seems that the new model's channels were not automatically added to the master file in the framebuilder (/opt/rtcds/caltech/c1/target/master). Adding the following two lines to the master file fixed the problem;
The box is now green. It looks like C1ASX.ini is created automatically in /opt/rtcds/caltech/c1/chans but the master file needs to be manually edited. The channels are now showing up on dataviewer etc. I have updated the information on the wiki page.
I have made some minor changes to the model, made all the MEDM screens, and linked monitors on these to the appropriate channels. I have borrowed heavily from the C1ASS MEDM screens (particularly for the small filter modules-it was convenient to just copy and paste an existing module, and edit the channel names using EMACS/GEDIT), and have edited these to suit the needs of this servo. Some features:
I think I am now ready to take some measurements and try and optimize this servo. There is no green transmission at the PSL table at the moment, so not much can be done, though the first step would be to take the power spectrum of the error signal, which would help me decide the appropriate frequencies for the LOs. I would then have to add the appropriate filters to the model. The last, and most difficult step, would be the measurement of the output matrix, though Koji has given me some ideas about how this measurement can be done. I also have a template script ready, though I will only finalise this after optimising the servo and running it a couple of times manually.
Attached are screenshots of the MEDM screens.
The following slow channels have been added and are now being recorded by FB.
In order to integrate the data collected by the Raspberry-Pi from the Y-end doubling oven temperature controller and also the data from the frequency counter which will be hooked up to monitor the beat frequency, Koji helped me set up some slow EPICS record channels (in ALS as we felt this was most appropriate). The procedure for setting up slow channels was as follows (virtually identical to what is detailed in this elog:
I will now integrate these channels into my scripts, and make some simple MEDM screens.
In order to decide what frequencies to dither the 4 degrees of freedom (M1-pitch&yaw, M2-pitch&yaw) at, I took the power spectrum of the X and Y-arm green transmission (C1:ALS-TRX_OUT, C1:ALS-TRY_OUT). Plots showing the power spectra are attached. Looking at the power spectra, I would think that for the X-arm, it would be okay to dither at 40, 50, 60 and 70 Hz. In order to check if the piezos could respond to these frequencies, I used my QPD setup and shook the PZTs with a 100Hz, 1Vpp sinusoid, and saw that the spot moved smoothly on the QPD.
As for choosing the modulation amplitude, I did a simplistic approximation assuming that the misalignment only rotates the beam axis relative to the cavity axis, and determined what angle coupled 10% of the power into the next eigenmode. Assuming that this is small enough such that if we are already locked to TEM00, the dither won't kick it up to some higher-order mode, the LO amplitude should be in the range of 30-60 digital counts (determined using the PZT calibration constants determined here. This corresponds to a sine-wave of ~50mV amplitude reaching the PZTs (after HV amplification). I am not sure if this is too small, but according to the PZT datasheet, these platforms are supposed to have a resolution of 0.02 urad, which would correspond to the input signal changing by ~0.1 mV, so this signal should be capable of dithering the tip-tilt.
I have already added band-pass filters centered at these frequencies to the model (with a passband of 5Hz, 2Hz on either side), and low-pass filters to pull out the DC component of the output of the lock-in amplifiers. It remains to tune the gains of the filter stages. These parameters (frequency, amplitude of the LOs) may also have to be changed after tests). Hopefully the PZTs can be plugged in tomorrow, and I can try and make a measurement of the output matrix.
Koji also suggested that it may be good to have a path in the model that feeds back to the PZTs by dithering the cavity mirrors as opposed to the PZT mounted mirrors. I will work on incorporating this into the SIMULINK model (c1asx.mdl) and also into the master medm screen.
Power Spectra of Arm Green Transmission:
I had prepared two more PZT mounted mirrors for the Y-end some time back. These are:
I used the same QPD set-up and the methodology described here to do a full-range calibration of these PZTs. Plots attached. The calibration constants have been determined to be:
CVI-pitch: 0.316 mrad/V
CVI-yaw: 0.4018 mrad/V
Laseroptik pitch: 0.2447 mrad/V
Laseroptik yaw: 0.2822 mrad/V
CVI YAW CVI PITCH
Laseroptik YAW Laseroptik PITCH
I redid the power spectrum measurement for the X-arm green transmission after aligning the arm to green using the ITMX/ETMX Pitch and Yaw sliders on IFOalign.
The Y-axis now reflects the relative intensity noise (RIN), which I obtained by taking the average value of the X-arm green transmission using tdsavg. Based on this measurement, I have now picked four new frequencies at which to try and modulate the PZT mirrors: 10, 19, 34 and 39 Hz. Bandpass filters in the LIA stage have been appropriately modified.
I have done some preliminary testing of the X-End Green ASS Servo. I will write a more detailed elog about this soon, but I thought I'd note down the important stuff here.
Yesterday, while we were venting, I aligned the X-arm to the green using the sliders on IFOalign, maximizing the transmission. Then I retook a power spectrum so as to determine the LO frequencies. Jenne pointed out that LO frequencies should not be integers (it usually suffices to append a .098725 or something to the frequency) so I made the necessary changes.
I did a first run of the servo yesterday, and more runs today. Notable points:
More details to follow.
Over the last three days, I've had the interferometer to test and optimize the ASX Servo. Based on what I have seen, I think the conclusion is that with the current parameters, the servo does its job provided the input pointing set up at the endtable with the coarse adjustment knobs is reasonably good. Once the cavity is aligned and IR transmission maximized using ASS, I have been able to get the green transmission up to 0.8 which is close to the best we had pre-vent. I have not been elogging regularly over the last few days, so this one is going to be a longish one.
Major changes made:
Details of tests runs:
For the most part, I have been trying to optimize the PZT mirror dither servo. To this end, I did the following:
Attempt to measure transfer function:
One of the things that came up during my presentation was how fast the loop was capable of responding. I was able to get a quantitative idea of this by playing around with the overall servo gain. Initially, it took ~30 seconds for the servo to get the transmission up to its peak value, with a servo gain of 1. When I ramped this up to 5, the response was much faster, with the peak transmission being reached in ~5seconds.
I wanted to get a more quantitative picture, and hence tried to measure the transfer function by first injecting an excitation into the 'SIG' filter-bank in the demodulation stage. However, coherence between the IN1 and IN2 signals was very poor for all the amplitude configurations I tried. At Jenne's suggestion, I tried injecting the excitation at the control-filters stage, but found no improvement. Perhaps the amplitude envelope was wrong or the measurement technique has to be rethought.
New MEDM screen:
We fixed the problematic DIN connectors on 1Y2, by swapping out the 3 DIN connector blocks that were of the wrong type (see attached image for the difference between the types appropriate for "Live" and "Ground").
Before doing anything, Eric turned the Wenzel multiplier off. We have not turned this back on.
Then we turned off the power supply unit at the base of 1Y2, removed the connectors from the rail, swapped out the connectors, reinstalled them on the rail, and turned the power supply back on. After swapping these out, we verified with a multimeter that between each pair of "Live" and "Ground" blocks, there was ~15V. We could now use the third unused pair of blocks to power the delay line phase shifter box, though for the moment, it remains powered by the bench power supply.
1. POP110 RF amps are powered from the cross connect. But that +15V block has GND connections that are not connected to the ground.
i.e. The ground potential is given by the signal ground. (Attachment 1)
This is caused by the misuse of the DIN connector blocks. The hod side uses an isolated block assuming a fuse is inserted.
However, the ground sides also have the isolated blocks
2. One of the POP110 RF cable has a suspicious shiled. The rigidity of the cable is low, suggesting the broken shield. (Attachment 2)
I made some changes to the c1tst model running on c1iscey in order to test my algorithm for frequency counting. I followed the steps listed in elog 8909 to make, install and start the model.
I need to debug a few things and run some more diagnostics so I am leaving the model in its edited version (Eric had committed it to the svn before I made any changes).
I have been working on setting up a frequency counting module that can give us a readout of the beat frequency, divided by a factor of 2^14 using the Wenzel frequency dividers as described here. This is a summary of what I have thus far.
The algorithm, and simulink model
The basic idea is to pass the digitized signal through a Schmitt trigger (existing RCG module), which provides some noise immunity, and should in theory output a clean square wave with the same frequency as the input. The output of the Schmitt trigger module is either 0 (for input < lower threshold value) and 1 (for input greater than the high threshold value). By differencing this between successive samples, we can detect a "zero-crossing", and by measuring the time interval between successive zero crossings, we can take the reciprocal to get the frequency. The last bit of this operation (i.e. measuring the interval) is done using a piece of custom C code. Initially, I was trying to use the part "GPS" from CDS_PARTS to get the current GPS time and hence measure intervals between successive zero-crossings, but this didn't work out because the output of GPS is in seconds, and that doesn't give me the required precision to count frequency. I tried implementing some more precision timing using the clock_gettime() function, which is capable of giving nanosecond precision, but this didn't work for me. So I am now using a more crude way of measuring the interval, by using a counter variable that is incremented each time a zero-crossing is NOT detected, and then converting this to time using the FE_RATE macro (=16384). In any case, the ADC sampling rate limits the resolution of frequency counting using zero-crossing detection (more on this later). Attachment 1 shows the SIMULINK block diagram for this entire procedure.
Testing the model
I implemented all of this on c1tst, and followed the steps listed here to get the model up and running. I then used one of the DB37 breakout boards to send a signal to the ADC using the DS345 function generator. Attachment 2 shows some diagnostic plots - input signal was a 2.5Vpp (chosen to match the output from the Wenzel dividers) square wave at 2kHz:
The right column pointed me to the limitations of frequency counting using this method - even though the input frequency was constant (2kHz), the counter variable, and hence the frequency readout, was neither accurate nor precise. But this was to be expected given the limitations imposed by ADC sampling? We only get information of the state of the input signal once within each sampling interval, and hence, we cannot know if a zero crossing has occurred until the next sampling interval. Moreover, we can only count frequency in discrete steps. In attachments 3 and 4, I've plotted these discrete frequencies which can be measured - the error bars indicate the error in the frequency readout if the counter variable is 1 more or less than the "true" value - this can (and does) happen if the high and low times of the Schmitt trigger are not equal over time (see top left plot in Attachment 2, its not very obvious, but all the "low" times are not equal, and so, the interval between detected zero crossings is not equal). This becomes a problem for small values of the counter variable, i.e. at high input frequencies. I was having a look at the elogs Aidan wrote some years ago for a different digital frequency counting approach, and I guess the conclusion there was similar - for high input frequencies, the error is large.
I further did two frequency sweeps using the DS345, to see if I could recover this in the frequency readout. Attachments 5 and 6 show the results of these sweeps. For low frequencies, i.e. 100-500 Hz, the jitter in the readout is small (though this will be multiplied by a factor of 2^14), but by the time the input frequency gets up to 2kHz, the jitter in the readout is pretty bad (and gets worse for even higher frequencies.
Some refinements can be made to the algorithm, perhaps by introducing some averaging (i.e. not reading out frequency for every pair of zero crossings, but every 5) which may improve the jitter in the readout, but I would think that the current approach is not very useful above 2kHz (corresponding to ~30MHz of pre-divider frequency), because of the limitations shown in attachments 3 and 4.
Earlier today, the front panels for the 1U chassis I obtained to house the Wenzel dividers + RF amplifiers arrived, which meant that finally I had everything needed to complete the assembly. Pictures of the finished arrangement attached.
Summary of the arrangement:
Once I figure out the problem with this amplifier/replace it, the box is ready to be installed.
The new 2x2 fiber couplers arrived today so Eric gave me an overview on the changes to be made to the existing configuration of the FOL fiber box. I removed the box from the table after ensuring that the PDs were powered OFF and removing and capping all fiber leads on the front panel. Here is a summary of the changes made.
I then ran a quick check to see what the power levels were at the input to the PDs, using the fiber coupled power meter. However, I found that there was no light in the fiber marked "PSL light in" (the power meter read out "Sig. Low"). The X arm Aux light had an input power of 1.12 mW, which after the various coupling losses etc went down to 63 uW just before the PD. The corresponding figures for the Y arm are 200 uW and 2.2 uW. I am not too sure of how the AUX light is coupled into fibers so I am not trying to tweak the alignment to see if I get more power.
Eric pointed out that the 1x2 couplers that were used in the previous arrangement and which I recycled, were in fact NOT appropriate - they are not 50-50 couplers but 90-10 couplers, which explains the measured power levels I quoted here.
I switched out these for a pair of the newly arrived 2x2 couplers, and have also replaced the datasheets on the inside of the top cover. I then redid the power level measurements, and got some sensible values this time (see Attachment #1 for revised layout and measured power levels, numbers in red are powers for PSL light, numbers in green are for AUX laser light, and all numbers are in mW). I did find that the 90-10 splitter in the PSL+Y path was not working (though the one in the PSL+X path seems to be working fine), and hence, have not quoted power levels at the output of these splitters. For now, I guess we can bypass the splitters and take the PSL+AUX light from the 2x2 couplers directly to the PDs.
We had a look at the IR beat (PSL+Xarm) today using the new FOL fiber box, and compared it to the green beat signal for the same combination. We first switched out the green Y beat input into the RF amplifiers on the PSL table with the PSL+Xarm IR beat input (so in all the plots, the BEATY channels really correspond to the IR beat for PSL+X). The IR and green beat notes were found without much difficulty, and we compared the beat signal PSDs for the green and IR signals (see Attachment #1 - arms were locked to green and the X slow control was turned on). The pink trace (labeled REF1) corresponds to the green beat signal, and was in good agreement with an earlier reference trace Eric had saved for the same signal. The teal trace (labeled REF0) corresponds to the the IR beat signal monitored simultaneously.
We then went back to the PSL table to check the amplitude of the signal from the broadband fiber PDs using the Agilent network analyzer. An initial measurement yielded a beat note (@~50MHz) at ~-22dbm (17mV rms). We figured that by bypassing the 90-10 splitter in this path, we could get a stronger signal. But after switching out the fiber connections we found that the signal amplitude had fallen to ~-27dbm (10mV rms). As per my earlier measurements here, we expect ~600uW of light on the PD, and a quick calculation suggested the signal should be more like 60mV, so we used the fiber power meter to check the power levels after each of the couplers again. We then found that the fiber connector on the front panel of the box for the PSL input wasnt ideal (the laser power after the first 50-50 coupler was only ~250 uW, though the input was ~1.2 mW). The power after the first coupler also fluctuated unpredictably (<100 uW to 350 uW) in response to slightly tightening/loosening the fiber connections on the front panel. I then switched the PSL input to one of the two unused fiber connectors on the front panel (meant for the 10% of the beat signal for the DC readout), and found that this input behaved much better, with ~450 uW of power available after the first 50-50 coupler. The power going into the beat PD was also measured to be ~550uW, closer to what was expected. The beat signal peak now was ~-14dbm (~30mV rms).
We then once again repeated the comparison between green and IR beat signals - but while in the control room, I noticed that the beat signal amplitude on the network analyzer in the control room was fluctuating by nearly 1.5 divisions on the vertical scale - not sure what the reason for this is. A look at the PSD of the IR beat with higher power incident on the PD was also not encouraging (see blue trace in Attachment #1), it seems to have gotten worse in the 10-30 Hz range. We also looked at the coherence between the beat spectrum and the beat note amplitude in order to look for any linear coupling between the two, but from Attachment #2, we cannot explain the disparity between the green and IR beat spectra. This warrants further investigation.
Everything on the PSL table has now been restored to the configurations before these investigations (i.e. the Y+PSL green beat cable has been reconnected to the RF amplifier, and both green beat PDs have been powered back ON. The fiber PDs are powered OFF)
The new ZKL-1R5 RF amplifier that Steve ordered arrived yesterday. I installed this in the frequency divider box and did a quick check using the Fluke RF signal generator and an oscilloscope to verify that both the X and Y paths were working.
I've now installed the box in the 1X2 rack where the olf "RF amplifiers for ALS and FOL" box used to sit (I swapped that out as I needed the L brackets on that chassis to mount mine, see Attachment #1 for the new layout). The power cable that used to power the old chassis was available, but the connector was of the wrong gender, so I had to switch this out. After verifying that I was getting the correct voltage (+15V), I connected it to the chassis.
I then did a quick check with the Fluke generator to make sure that all was working as expected - Eric had set up some ADC channels for me earlier today in the C1ALS model, and I copied over my frequency counting module from C1TST into C1ALS, and recompiled the model. The RF generator was set to generate a 25MHz signal at -20dBm, which I then split using an RF power splitter between the X and Y arms. I then checked the output using dataviewer - I recovered an output frequency of ~27.64 MHz with a jitter of ~0.02 MHz with a 20Hz low-pass filter in place (see Attachment #2), which looks consistent with the systematic error inherent in the zero-crossing counting algorithm and random fluctuations I had observed in my earlier trials, discussed here. But a more systematic investigation needs to be carried out in this regard. The interfacing between the hardware and software seems to be working alright though. I've left the RF generator near the 1x2 rack for now, though its powered off.
The mode cleaner unlocked quite a few times while I was working but looks stable now.
I carried out some more tests on the digital frequency counting system today, mainly to see if the actual performance mirrors the expected systematic errors I had calculated here.
Setup and measurement details:
I used the Fluke 6061A RF signal generator to output an RF signal at various frequencies, one at a time, between 10 and 70 MHz. I split the signal (at -15 dBm) into two parts, one for the X-channel and one for the Y-channel using a mini-circuits splitter. I then looked at the input signal using testpoints I had set up within the model, to decide what thresholds to set for the Scmitt trigger. Finally, I averaged the outputs of the X and Y channels using z avg -s 10 C1:ALS-FC_X_FREQUENCY_OUT and also looked at the standard deviation as a measure of the fluctuations in the output (these averages were taken after a low-pass filter stage with two poles at 20Hz, chosen arbitrarily).
I carried out some further diagnostics and found some ways in which I could optimize the zero-crossing-counting algorithm, such that the error in the measured frequency is now entirely within the expected range (due to a +-1 clock cycle error in the counting). We can now determine frequencies up to ~60 MHz with less than 1 MHz systematic error and <10 kHz statistical error (fluctuations after the 20 Hz lowpass). This should be sufficient for slow control of the end-laser temperatures.
The conclusion from my earleir tests was that there was possibly an improvement that could be made to setting the thresholds for the Schmitt trigger stage in the model. In order to investigate this, I wanted to have a look at the 64K sampled raw input to the ADCs. Yesterday Eric helped me edit the appropriate .par file for viewing these channels for c1x03, and for an input frequency of 70MHz (after division, ~4.3 kHz square wave), the signal looked as expected (top left plot, attachment #1). This prompted me to check the counting algorithm again with the help of various test points I had setup within the model. I found that there was a tendency to under-count the number of clock-cycles between zero-crossings by more than 1 clock cycle, due to the way my code was organized. I fixed this and found that the performance improved dramatically, compared to my previous trials. With the revised counting algortihm, there was at most a +-1 clock cycle error in the counting, and the systematic error between the measured and requested RF frequencies is now completely accounted for taking this consideration into account. The origin of this residual error can be understood by looking at the top right plot in Attachment #1 - presumably because of the effects of some downsampling filter, the input signal to the Schmitt trigger isnt a clean square wave (even at 4kHz) - specifically, the time spent in the LOW and HIGH states of the Schmitt trigger can vary between successive zero crossings because of the shape of the input waveform. As a result, there can be a +-1 clock cycle error in the counting process. Attachment #2 shows this - the red and blue lines envelope the measured frequency for the whole range investigated: 10-70MHz. Attachment #3 shows the systematic error as a function of the requested frequency.
If there was some way to bypass the downsampling filter, perhaps the high-frequency performance could be improved a little.
Steve pointed out that in the aftermath of the Nitrogen running out a couple of times last week, the RGA had shut itself off thinking that there was a leak and so it was not performing the scheduled scans once a day. So the data files from the scheduled scans were empty in the /opt/rtcds/caltech/c1/scripts/RGA/logs directory. The wiki page for getting it up and running again is up-to-date, but the script RGAset.py did not exist on the c0rga machine, which the RGA is communicating with via serial port. I copied over the script RGAset.py from rossa to c0rga and ran the script on that machine - but the error flags it returned were not all 0 (indicating some error according to the manual) - so I edited the script to send just the initialize command ('IN0') and commented out the other commands, after which I got error flags which were all 0. After this, I ran a manual scan using 'RGAlogger.py', and it appears that the RGA is now able to take scans again - I'm attaching a plot of the scan results. We've saved this scan as a reference to compare against after a few days.
I'm working on setting up a moving-average in the custom C code block that counts the zero crossings to see if this approach is able to mitigate the glitchy frequency readout due to mis-counting by one clock cycle between successive zero crossings. I'm storing an array the size of the moving average window of frequency readouts at each clock cycle, and then taking the arithmetic mean over the window. By keeping a summing variable that updates itself each clock cycle, the actual moving average process isnt very intensive in terms of computational time. The array does take up some memory, but even if I perform the moving average over 1 second with 16384 double precision numbers stored in the array, its still only 130 kB so I don't think it is a concern. Some tests I've been doing while tuning the code suggest that with a moving average over 16384 samples (i.e. 1 second), I can eliminate glitches at the 1Hz level in the frequency readout for frequencies up to 5 kHz (generated digitally using an oscillator block). Some tuning still needs to be done, and the window could possibly be shortened. I also need to take a look at the systematic errors in this revised counting scheme, preferably with an analog source, but this is overall, I think, an improvement.
On a side note, I noticed some strange behaviour while running the cds average command - even though my signal had zero fluctuations, using z avg 10 -s C1:TST-FC_FREQUENCY_OUT gave me a standard deviation of ~1 kHz. I'm not sure what the problem is here, but all the calibration data I took in earlier trials were obtained using this so it would be useful to perform the calibration again.
I've made quite a few changes in the software as well as the hardware of the digital frequency counting setup.
Eric helped me test the new setup by doing an arm scan through an IR resonance by ramping the ALS offset from -3 to +3 with a ramp time of 45 seconds. The data was acquired with the window size of the moving average set to 4096 clock cycles, and a 2 Hz low pass IIR filter before the frequency readout. Attachment 1 shows a plot of the data, and a fit with a function of the form trans = a/(1+((x-b)/c)^2), where a = normalization, b = center of lorentzian, and c = linewidth (FWHM) of the peak (the fitted parameter values, along with 95% confidence bounds are also quoted on the plot). In terms of the data acquisition, comparing this dataset to one from an earlier scan Eric did (elog11111) suggests that the frequency counting setup is working reasonably well - at any rate, I think the data is a lot cleaner than before implementing the moving average and having a 20Hz lowpass IIR filter. In any case, we plan to repeat this measurement sometime next week during a nighttime locking session. It remains to calculate the arm loss from these numbers analogous to what was done earlier for the X arm.
Calculation of loss:
Fitted linewidth = 10.884 kHz +/- 11Hz (95% C.I.)
FSR of Y arm (from elog 9804) = 3.9468 MHz +/- 1.1 kHz
=> Y arm Finesse = FSR/fitted linewidth = 362.6 +/- 0.5
Total round trip loss = 2*pi/Finesse = 0.0173
Sorry for the confusion - I did mean Green beat frequency, and I had neglected the factor of 2 in my earlier calculations. However, the fit parameter "c" in my fit was actually the half-width at half maximum and not the full width at half maximum. After correcting for both these errors (new fit is Attachment #1, where I have now accounted for the factor of 2, and the X axis is the IR beat frequency), I don't think the numbers change too much. It could be that the frequency counter wasn't reading out the frequency correctly, but looking at a time series plot of the frequency counter readout (Attachment #2), and my earlier trials, I don't think this is the case (38 MHz is a frequency at which I don't expect much systematic error - also, the offset was stepped from -3 to 3 over 45 seconds).
The revised numbers:
Fitted linewidth = 2*c = 10.884 kHz +/- 2 Hz (95% C.I.)
I found (an old) 10 dB coupler in the RF component shelves near MC2 - it has BNC connectors and not SMA connectors, but I thought it would be worth it to switch out the 20dB coupler currently on the X green beat PD on the PSL table with it. I used some BNC to SMA adaptors for this purpose. It appears that the coupler works, because I am now able to register an input signal on the X arm channel of the digital frequency counter (i.e. the coupled output from the green beat PD). I thought it may be useful to have this in place and do an IR transmissions arm scan using ALS for the X arm as well, in order to compare the results with those discussed here. However, the beat note amplitude on the analyzer in the control room looks noticeably lower - I am not sure if the coupler is responsible for this or if it has to do with the problems we have been having with the X end laser (the green transmission doesn't look glitchy on striptool though, and the transmission itself is ~0.45). In any case, we could always remove the coupler if this is hindering locking efforts tonight.
I began my attempts to characterize the PDH loops at the X end today. My goal was to make the following measurements:
which I can then put into my simulink noise-budget scheme for the proposed IR beat setup.
I've made an Optickle model of a simple FP cavity and intend to match the measured PDH error signal from the X end to the simulated error signal to get the Hz/V calibration. I'll put the plots up for these shortly.
With regards to the other measurements, I was slowed down by remote data-acquisition from the SR785 - I've only managed to collect the analyzer noise floor data, and I plan to continue these measurements during the day tomorrow.
Over the course of my investigations into the systematic errors in the frequency readout using digital frequency counting, I noticed that my counter variable that keeps track of the number of clock cycles between successive zero crossings was NOT oscillating between 2 values as I would have expected (because of there being a +/- 1 clock cycle difference between successive zero crossings due to the fixed sampling time of 1/16384 seconds), but that there were occassional excursions to values that were +/- 3 clock cycles away. I then checked the output of the SCHMITTTRIGGER CDS library part (which I was using to provide some noise immunity), and noticed that it wasn't triggering on every zero crossing at higher frequencies. I tested this by hooking up a digital oscillator to the SCHMITTTRIGGER part, and looked at its output for different frequencies. The parameters used were as follows:
SCHMITTTRIGGER lower threshold: -1.0
SCHMITTTRIGGER upper threshold: +1.0
I am attaching plots for two frequencies, 3000Hz and 4628Hz (Attachments #1 and #2) . I would have expected a flip in the state of the output of the schmitt trigger between every pair of horizontal red lines in this plot, but at 4628 Hz, it looks like the schmitt trigger isn't catching some of the zero crossings? Come to think of it, I am not even sure why the output of the schmitt trigger takes on any values other than 0 or 1 (could this be an artefact of some sort of interpolation in the visualization of these plots? But this would not affect the conclusion about the schmitt trigger missing some of the zero crossings?)
As an interim measure, I implemented a Schmitt trigger in my C code block - it was just a couple of extra lines anyways - I have designated the schmitt trigger output as a static variable that should hold its value in successive execution cycles, unless it is updated by comparing the input value to the thresholds (code attached for reference). Attachments #3 and #4 show the output of this implementation of a Schmitt trigger at the same two frequencies, and I am seeing the expected flip in the state between successive zero crossings as expected (though I'm still not sure why it takes on values other than 0 and 1?). Anyways this warrants further investigation. An elog regarding the implications of this on the systematic error in the frequency counter readout to follow.
I was trying to take a few more IR transmission scans with ALS when the ETMX got kicked again. I'm not sure how to fix this, so for the time being, I'm leaving the Oplev servo and the LSC turned off. The oplev spot looks really far off center especially in yaw, the yaw error is ~ -80.
The oplev and the LSC are off.
While the ETMx issues are being investigated - with Eric's help, I took some data from arm scans of the Y arm through ~2FSRs using ALS. I've also collected the data from the frequency counter readout during these scans but since they were done rather fast (over 60seconds), I am not sure how accurate this data will be. The idea however is to use the frequency readout from the phase tracker - this has to be linearized though, which I will do during the daytime tomorrow. The plan is to use our GPS timing unit to synchronize the following chain :
GPS timing unit 1PPS out --> FS725 Rb Clock 1PPS in (I recovered one which was borrowed from the 40m some time ago from the ATF lab yesterday evening, waiting for it to lock to the Rb clock now)
FS725 Rb Clock 10 MHz out --> Fluke 6061A 10MHz reference in
FS725 Rb Clock 10 MHz out-->agilent network analyzer 10MHz reference in (for measurement of the frequency of the signal output from the Fluke RF signal generator independent of its front panel display)
Then I plan to look at the phase tracker output as a function of the driving frequency (which will also be measured, offline, using the digital frequency counter setup) over a range of 20 MHz - 50 MHz in steps of 1 MHz. Results to follow.
Earlier tonight, Eric and I tweaked the PMC alignment (the mode cleaner was not staying locked for more than a couple of minutes, for almost an hour).
I've made a few more changes to the frequency counting code - these are mostly details and the algorithm is essentially unchanged.
The other thing that came up in the meeting last week was this issue of the systematic errors in the measured frequency, and how it was always over-estimating the 'actual' frequency. I've been investigating the origin of this over the last few days, and think I've found an explanation. But first, Attachment #1 shows why there is a systematic error in the first place - because we are counting the period of the input signal in terms of clock cycles, which can only take on discrete, integer values, we expect this number to fluctuate between the two integers bounding the 'true value'. So, if I'm trying to measure an input signal of 3000Hz, I would measure its period as either 5 or 6 clock cycles, while the "true" value should be 5.4613 clock cycles. In attachment #1, I've plotted the actual measured frequency and the measured frequency if we always undercounted/overcounted to the nearest integer clock cycles, as functions of the requested frequency. So the observed systematic error is consistent with what is to be expected.
The reason why this doesn't average out to zero is shown in Attachment #2. In order to investigate this further, I recorded some additional diagnostic variables. If I were to average the period (in terms of clock cycles - i.e. I look for the peaks in the blue cuve, add them up, and divide by the number of peaks), I find that I can recover the expected period in terms of clock cycles pretty accurately. However, the way the code is set up at the moment, the c code block outputs a value every 1/16384 seconds (red curve) - but this is only updated each time I detect a zero crossing - and as a result, if I average this, I am in effect performing some sort of weighted average that distorts the true ratio of the number of times each integer clock-cycle-period is observed. This is the origin on the systematic error, and is a function of the relative frequency each of the two integer values of the clock-cycle-period occurs, which explains why the systematic error was a function of the requested frequency as seen in Attachment #1, and not a constant offset.
At the frequencies I investigated (10-70MHz in 5MHz steps), the maximum systematic error was ~1%.
Is there a fix?
I've been reading up a bit on the two approaches to frequency counting - direct and reciprocal. My algorithm is the latter, which is generally regarded as the more precise of the two. However, in both these approaches, there is a parameter known as the 'gate-time': this is effectively how long a frequency counter measures for before outputting a value. In the current approach, the gate time is effectively 1/16384 seconds. I would think that it is perhaps possible to eliminate the systematic error by setting the gate time to something like 0.25 seconds, and within the gate time, do an average of the total number of periods measured. Something like 0.25 seconds should be long enough that if, within the window, we do the averaging, and between windows, we hold the averaged value, the systematic error could be eliminated. I will give this a try tomorrow. This would be different from the moving average approach already explored in that within the gate-time, I would perform the average only using those datapoints where the 'running counter variable' shown in Attachment #2 is reset to zero - this way, I avoid the artificial weighting that is an artefact of spitting out a value every clock cycle.
I performed a preliminary calibration of the X and Y phase trackers, and found that the slopes of a linear fit of phase tracker output as a function of driven frequency (as measured with digital frequency counter) are 0.7886 +/- 0.0016 and 0.9630 +/- 0.0012 respectively (see Attachments #1 and #2). Based on this, the EPICS calibration constants have been updated. The data used for calibration has also been uploaded (Attachment #4).
I found that by adopting the approach I suggested as a fix in elog 11736, and setting a gate time of 1second, I could eliminate the systematic bias in measured frequency I had been seeing, the origin of which is also discussed in elog 11736. This was verified by using a digital oscillator to supply the input to the frequency counting block, and verifying that I could recover the driving frequency without any systematic bias. Therefore, I used this as a measure of the driving frequency independent of the front panel display of the Fluke 6061A.
The actual calibration was done as follows:
Y-arm transmission scan
I used the information from Attachment #2 to calibrate the X-axis of the Y-arm transmission data I collected on Wednesday evening. Looking at the beat frequency on the analyzer in the control room, between 24 and 47 MHz (green beat frequency, within the range the calibration was done over), we saw three IR resonances. I've marked these peaks, and also the 11MHz sideband resonances, in Attachment #3. It remains to fit the various peaks. I did a quick calculation of the FSR, and the number I got using these 3 peaks is 3.9703 +/- 0.0049 MHz. This value is ~23 kHz greater than that reported in elog 9804, but the error is also ~4 times greater (6 IR resonances were scanned in elog 9804) so I think these measurements are consistent.
I had brought an FS725 Rubidium clock back from W Bridge - the idea was to hook this up to the GPS 1PPS output, and use the 10MHz output from the FS725 as the reference for the fluke 6061A. However, the FS725 has not locked to the Rb frequency even though it has been left powered on for ~2days now. Do I have to do something else to get it to lock? The manual says that it should lock within 7 minutes of being powered on. Once this is locked, I can repeat the calibration with an 'absolute' frequency source...
There are two modulation frequencies that make it to the arm cavities, at ~11MHz and ~55MHz. Each of these will have their own modulation depth indepedent of each other. Bundling them together into one number doesn't tell us what's really going on.
As an update to Yutaro's earlier post - I've done an independent study of this data, doing the fitting with MATLAB, and trying to estimate (i) the FSR, (ii) the mode matching efficienct, and (iii) the modulation depths at 11MHz and 55MHz.
The values I've obtained are as follows:
FSR = 3.9704 MHz +/- 17 kHz
Mode matching efficiency = 92.59 % (TEM00 = 1, TEM10 = 0.0325, TEM20 = 0.0475)
Modulation depth at 11MHz = 0.179
Modulation depth at 55MHz = 0.131
Misc Remarks and Conclusions:
After thinking about the interpretation of the various peaks seen in the scan through 2 FSRs, I have revised the information presented in the previous elog. Yutaro pointed out that the modulation frequency isn't exactly 11MHz, but according to this elog, is 11.066209 MHz. So instead of using mod(11e6,FSR), I really should have been using mod(11.066209,FSR) and mod(5*11.066209,FSR) to locate the positions of the 11MHz and 55MHz sidebands relative to the carrier resonances. With this correction, the 'unknown' peaks identified in Attachment #1 in elog 11743 are in fact the 55MHz sideband resonances.
However, this means that the peaks which were previously identified as 55MHz sideband resonances have to be interpreted now - I'm having trouble identifying these. If we assume that the types of peaks present in the scan are 11 MHz sideband, 55MHz sideband, and the TEM00, TEM10, TEM20, TEM30, and TEM40 mode resonances, then the peaks marked in grey in Attachment #1 to this elog can be interpreted as TEM30 (right of a carrier resonance) and TEM40 (left of a carrier resonance) mode resonances - however, the fitted center frequencies differ from the expected center frequencies (determined using the same method as elog 469) by ~3% (for TEM30) and ~20% (for TEM40) - therefore I am skeptical about these peaks, particularly the 4th HOM resonances. In any case, they are the smallest of all the peaks, and any correction due to them will be small.
The updated modulation depths are as follows (computed using the same method as described in elog 11743, the updated plot showing the ratio of bessel functions as a function of the modulation depth is Attachment #2 in this elog):
@11.066209 MHz ---- 0.179
@5*11.066209 MHz --- 0.226
These numbers are now reasonably consistent with those reported in elog10211.
As for the mode-matching efficiency, the overall number is almost unchanged if I assume the TEM30 peaks are accurately interpreted: 92.11%. But the dominant HOM contribution comes from the first HOM resonance: (TEM00 = 1, TEM20 = 0.0325, TEM10 = 0.0475, TEM30 = 0.0056). These numbers may change slightly if the 4th HOM resonances are also correctly identified.
ETMx is still not well behaved and the mode cleaner isnt too happy either, so I think we will save the measurement of the round trip arm loss for daytime tomorrow.
In the last few days, with Koji's help, I have recovered both the FS725 Rubidium references from W. Bridge, one from the ATF lab, and one from the CTN lab. Both are back at the 40m at the moment.
However, the one that was recovered from the ATF lab is no longer locking to the Rubidium reference frequency, although it was locked at the time we disconnected it from the ATF lab. I emailed the support staff at SRS, who seem to think that either the internal oscillator has drifted too far, or the Rb lamp is dead. Either ways, it needs to be repaired. They suggested that I run a check by issuing some serial commands to the unit to determine which of these is actually the problem, but I've been having some trouble setting up the serial link - I will try this again tomorrow. I'm also having trouble generating an RMA number that is needed to start the repair/maintenance process, but I've emailed SRS support again and hope to hear back from them soon.
The other FS725, recovered from the CTN lab earlier today, seems to work fine and is locked to the Rb reference at the moment. I plan to redo the calibration of the phase tracker with an 'absolute' frequency reference with the help of the FS725 and out GPS timing unit tomorrow. Once that is done, the working unit can be returned to the CTN lab.
In order to synchronise the FS725 Rb clock with our GPS timing signals, I laid out a longish cable running from 1X7 to the IOO rack via the overhead cable guide. There was a T-connector attached to the 1pps output of the GPS timing unit, with one of the outputs unused - I have connected one end of the cable I laid out to this output, with the other end going to the 1pps input of the FS725. I am now waiting for the FS725 to sync to the external reference, before running the calibration of the phase tracker once again using the same method detailed here, using the 10MHz output from the FS725 to serve as a reference for the Fluke RF signal generator...
Having obtained a working FS725 Rubidium standard and syncing it to out GPS timing unit, I wanted to have one more pass at calibrating the phase tracker output, with the RF signal generator calibrated relative to an 'absolute' source. I also extended the range of frequencies swept over to 15MHz to 110MHz. We found that the phase tracker output appears linear over the entire range scanned, but taking a closer look at the residuals suggested some quadratic structure. Restricting the fitted range to [31MHz 89MHz] yields the following calibration constants for the X and Y arm respectively: 0.9904 +/- 0.0008 and 0.9984 +/- 0.0005. This suggests that out previous calibration was pretty accurate, and that it is valid over a wider range of frequencies, so we could plausibly fit in more FSRs in future scans if necessary. I have not updated these values on the EPICS screens (though judging by how close they are to 1, I wonder if this is even necessary)...
The principle change in the setup compared to that used to collect the data presented in elog 11738 was the addition of the FS725 rubidium standard. As detailed here, I synced the Rubidium standard to our GPS timing unit (this took a while - the manual suggests it should only take minutes, but it took about 10 hours - the two photos in Attachment #1 show the status of the front panel before and after it synced to the external 1PPS input). I then took 10 MHz outputs from the FS725, and ran one to the Fluke 6061A, and the other to the AG4395A. The Fluke 6061 A has a small switch at the back which has to be set to "EXT" in order for it to use the external reference (it has now been returned to the "INT" state). We then connected the output of the signal generator via a 3-way minicircuits splitter to the AG4395A, and the two beat channels.
I cleared the phase history on the MEDM screen, and set the phase tracker UGF. We then swept through frequencies from 15MHz to 110MHz (using the AG4395 to verify the frequency at each step). I used the following command to record the average value (over 10 seconds) and the standard deviation: z avg 10 -s C1:ALS-BEATX_FINE_PHASE_OUT_HZ >> 20151113_PT_X.dat and so on.. The amplitude of the signal generated (i.e. before the splitter) was -18dBm (chosen such that the Q outputs of either phase tracker was between 1000 and 3000), while the gains were ~100 (X) and 50 (Y). I then downloaded the data and fitted it.
The output of the phase tracker looks roughly linear over the entire range of frequencies scanned - but looking at the residuals, one could say there was some quadratic structure to it (see residual plots in Attachment #2). By looking at the shapes of the residuals, I judged that if we fit in the range [31MHz 89MHz] (for both X and Y), we should see negligible structure in the residuals. Attachment #3 contains the fits and residuals for these fits. One could argue that there is still some structure in the residuals, but is markedly less than over the entire range, and, I think, small enough to be neglected. The calibration constants quoted at the beginning of the elog are from the fits over this range. In principle, we could always break this down into smaller pieces and do a linear fit over that range. But this should allow us to scan through >5 FSRs.
Since the beat signal also goes to the frequency counter via the couplers, I was also collecting the readouts of the frequency counter. Attachment #5 contains the data collected. It is interesting to note that the FCs fail at ~101 MHz (corresponding to ~6146 Hz after the dividers).
Also, we had taken another dataset last night, but found that there was an anomalous kink in the X phase tracker output at (coincidentally?) 89 MHz (I've attached the data in Attachment #6). I'm not sure why this happened, but this is what led me to take another dataset earlier today (Attachment #4).
Summary of Attachments:
ROC_ETMY = 59.3 +/- 0.1 m.
I followed a slightly different fitting approach to Yutaro's in an attempt to determine the g-factor of the Y arm cavity (details of which are below), from which I determined the FSR to be 3.932 +/- 0.005 MHz (which would mean the cavity length is 38.12 +/- 0.05 m) and the RoC of ETMY to be 60.5 +/- 0.2 m. This is roughly consistent (within 2 error bars) of the ATF measurement of the RoC of ETMY quoted here.
I set up the problem as follows: we have a bunch of peaks that have been identified as TEM00, TEM10... etc, and from the fitting, we have a bunch of central frequencies for the Lorentzian shapes. The equation governing the spacing of the HOM's from the TEM00 peaks is:
The main differences in my approach are the following:
A Caltech maintenance staff dropped by at around noon today, and told me that he had seen a small puddle of water on the other side of the door along the Y-arm that is kept locked (about 10m from the end-table, on the south side of the arm). He suspected a leak in the lab. Koji and I went down to the said door and observed that there was indeed a small puddle of water accumulated there. There isn't any obvious source of a leak on our side of the door, although the walls tiles in the area suggest that there could be a leak in one of the pipes running through the wall/under the floor. In any case, the leak doesn't seem too dramatic, and we have decided to consult Steve as to what is to be done about this once he is back on Wednesday.
I noticed that all the models running on C1LSC had crashed when I came in earlier today. I restarted all of them by ssh-ing into C1LSC and running rtcds restart all. The models seem to be running fine now.
We've been talking about putting in BLRMS filters for several channels - it would be a pain to manually copy over the correct bandpass and lowpass filter coefficients into the newly created filter banks, and so I've set up a script (attached) that can do the job. As template filters, I'vm using the filters rana detailed here. Essentially, what the script does is identify the (empty on creation) block of text for a given filter: e.g. RMS_STS1Z_BP_0p01_0p03 for STS1Z), and appends the template filter coefficients. To test my script, I first backed up the original C1PEM.txt file from /opt/rtcds/caltech/c1/chans, removed all the filter coefficients for the STS1Z BLRMS filters, and then replaced it with one generated using my script. I then loaded the coefficients for all the filters in the C1PEM modules, without any obvious error messages being generated. I also checked that foton could read the new file, and checked tmake sure that sensible filter shapes were seen for some channels. Since this seems to be working, I'm going to start putting in BLRMS blocks into the models tomorrow.