I consulted with Margot about the cleaning of the optics

• Optics are considered as a clean object. Large dusts can be removed by ionized N2 flow etc.
• Barrel of optics can be wiped with Acetone.
• Optical surfaces are best to be cleaned by First Contact.
• A peek mesh should be embedded in the first contact so that the First Contact sheet can be easily removed.
• When peeling a F.C. sheet from a mirror surface, ionized N2 should be brown for discharging.
• If there are residuals visible on the mirror surface, it should be removed by Acetone. Don't use alchols.
• Use paper lens tissue for wiping as the lint free wipe can be eaten by Acetone.
• In fact, All of the procedure is described in a certain document.
• For a small amount, Margot can provide us a bottle of F.C. and some PEEK meshes.

Details of the Ionized N2 system

• This N2 should have higher purity than 4N (UHP - Ultra High Purity). This means we should use 4N - UHP or 5N - Research Grade.
• The ionized gun used in the clean room at Downs: made by Terra Universal.com
• Flow path: N2 cylinder - Filter - Gun

Floor wiped with a wet wiper (Aug 13, 2012)
Floor wiped with a wet wiper (Aug 15, 2012)
Floor wiped with a wet wiper (Sep 25, 2012)
Air conditioning prefilter replaced (Sep 25, 2012)
Floor wiped with a wet wiper (Oct 01, 2012)
Floor wiped with a wet wiper (Nov 06, 2012) / ATF too
Floor wiped with a wet wiper (Jan 04, 2013)
Floor wiped with a wet wiper (Mar 23, 2013)
Floor wiped with a wet wiper (Apr 17, 2013)
Air conditioning prefilter replaced (Apr 17, 2013)
Floor wiped with a wet wiper (Jun 24, 2013)
Removing Vladimir's mess. Floor swept with a broom (Jun 26, 2013)
Completed removing Vladimir's mess. Floor swept with a wet wiper (Jun 27, 2013)
Air conditioning prefilter replaced (Sep 12, 2013)
Floor wiper head replaced. (Dec 10, 2013)
Floor wiped with a wet wiper (Dec 10, 2013)
Floor wiped with a wet wiper (Apr 1, 2014)

Air conditioning prefilter replaced (Dec 30, 2014)
Air conditioning prefilter replaced (some time in 2015...)

Floor wiped with a wet wiper (Dec 1, 2015)
Floor wiped with a wet wiper (Aug 23, 2016)
Air conditioning prefilter replaced (Aug 8, 2017) = 1 stock remains
Air conditioning prefilter replaced (Unkniwn) = no stock remains
Air conditioning prefilter replaced (Jul 25, 2022) = 5 stock remains
Floor wiped with a wet wiper (Mar 7, 2023)

Aug 13, 2012 / 0.5um 1000~2000/(0.1 cu ft) / 0.7um   400-600/(0.1 cu ft) by ATF particle counter (MET ONE 227A)

They are counts/(0.1 ft^3)! These numbers should be multiplied by 10 to know the particle "CLASS".

Particle counts

Before the prefilter is installed: 0.5um 1191cnts, 0.7um 346cnts

2:20 prefilter installed
2:25 0.5um 650 / 0.7um 255
3:00 0.5um 578 / 0.7um 99
4:00 0.5um 480 / 0.7um 102
5:00 0.5um 426 / 0.7um 76

They are counts/(0.1 ft^3)! These numbers should be multiplied by 10 to know the particle "CLASS".

1. It turned out that the particle counter MET ONE 227A at ATF shows
(particle count)/(0.1 ft^3)

This means that the numbers I saw previously should be multiplied by 10.
So the nominal class of the room was 5000.

2. As our GT-321s have no diffuser, I borrowed a diffuser from 227A.
The diffuser actually increases the count. We need to buy them.
All the measurments below are performed with the diffuser and calibrated in Count/ft^3.

3. Measured the particle level without the HEPA running.

With diffuser: [cnt/ft^3]

Over Class 10000

4. The two HEPA fans are turned on at the speed "MED".

Basically no particles are detected in the HEPA booth.

With diffuser, inside of the HEPA booth:

The particle level in the room (outside of the HEPA booth) is also improved

With diffuser, outside of the HEPA booth GT-321 #1:
0.3 um 18612
0.5 um   1728

5. The two HEPA fans are turned on at the speed "LOW".

Particle levels are still zero inside.

With diffuser, inside of the HEPA booth, GT-321 #1:
0.3 um 0
0.5 um 0

The particle level in the room (outside of the HEPA booth) is also improved
but the cleaning power for 0.3um seems degraded.

With diffuser, outside of the HEPA booth, GT-321 #1:
0.3 um 34488
0.5 um   1386

Steve and I visited Margot to have a training session for application of First Contact on optics.

- Make "thick" layer of first contact. It becomes thin when it gets dried.

- Apply more FC once a peek sheet is placed on the FC

- Wait for drying (~15min)

- Rip off the FC layer by pulling a peek tab. Make sure the ionized N2 is applied during ripping.

- Margot has a Dark Field Microscope. We checked how the dusts are removed from the surface.
There are many dusts on the mirror even if they are invisible. First Contact actually removes
these dusts very efficiently. Margot told us that even carbonhydrates (like finger prints) can be removed by FC.

The OMC is in the air bake oven now.

Suddenly something dirty emerged in the lab. What is this? It looks like an insulation foam or similar, but is quite degraded and emits a lot of particulates.

This does not belong to the lab. I don't see piping above this area which shows broken insulation or anything. All the pipes in the room are painted white.

The only possibility is that it comes from the hole between the next lab (CRIME Lab). I found that the A.C. today is much stronger and colder than last week. And there is a positive pressure from CRIME Lab. Maybe the foam was pushed out from the hole due to the differential pressure (or any RF cable action).

https://www.airscience.com/purair-flow-laminar-flow-cabinets

Rich saids:

I have ordered a small roll of solder for the OMC piezos. 
The alloy is: Sn96.5 Ag3.0 Cu0.5

Diode testing

o Purpose of the measurement

- Test Si QPDs (C30845EH) for ISC QPDs Qty 30 (i.e. 120 elements)

- Test InGaAs PDs (C30665GH) for OMC Qty 10 (i.e. 10 elements)

o Measurement Kit

- Inherited from Frank.

- Has relays in it.

- D0 and D1 switches the measurement instrument connected to an element

- D2 and D3 switches the element of the QPDs

- Digital switch summary

d0 d1 0 0 - ln preamp
d0 d1 1 0 - dark c
d0 d1 0 1 - omc preamp
d0 d1 1 1 - impedance

d2 d3 0 0 - A x x x
d2 d3 1 0 - C x o x
d2 d3 0 1 - B o x o
d2 d3 1 1 - D o o o

- The universal board in the box is currently configured for C30845.
  Pin1 - Elem A. Pin3 - B, Pin7 - C, Pin9 - D, Pin 12 - Case&Bias

o Labview interface

- Controls NI-USB-6009 USB DAQ interface and Agilent 82357B USB-GPIB interface

o Dark current measurement

- Borrowed Peter's source meter KEITHLEY 2635A

- For C30845GH the maxmum reverse bias is set to -20V. This drops the voltage of the each element to the bias voltage.

o Spectrum measurement

- The elements are connected to FEMTO LN current amp DLPCA-200.

- Bias voltage is set to +10V. This lifts up the outside of the amplifier input to +10V.

o Impedance measurement

- Agilent 4395A at PSL lab with impedance measurement kit

- For C30845GH the maxmum reverse bias is set to -15V. This drops the voltage of the each element to the bias voltage.

- Calibration: open - unplug the diode from the socket, short - use a piece of resister lead, 50Ohm - a thin metal resister 51Ohm

- Freq range: 30-50MHz where the response of the cables in the setup is mostly flat.

- Labview VI is configured to read the equivalent circuit parameters in the configuration "D" (series LCR).

- Labview fails to read the series resistance. This was solved by first read the equiv circuit param and then read it with Sim F-CHRST.
  F-CHRST does nothing on the parameters so the second request successfully acquires the first ones.

- For the dark noise measurement, the lid of the die-cast case should also contact to the box for better shielding. This made the 60Hz lines almost completely removed, although unknown 1kHz harmonics remains.

- The precise impedance of the setup can not be obtained from the measurement box; the cable in between is too long. The diode impedance should be measured with the impedance measurement kit.

- With the impedance measurement kit, the bias voltage of +5V should be used, in stead of -5V.

- diode characteristics measured at 10-100MHz

- Typical impedance characteristics of the diodes

Excelitas (Perkin-Elmer) C30665GH Rs=9Ohm, Cd=220pF, L=0~1nH (Vr=5V)

Excelitas (Perkin-Elmer) C30642G Rs=12Ohm, Cd=100pF, L=~5nH (Vr=5V) longer thin wire in a can?

Excelitas (Perkin-Elmer) C30641GH Rs=8Ohm, Cd=26pF, L=12nH (Vr=5V) leg inductance? (leg ~30mm)

- PD serial

C30665GH, Ls ~ 1nH

  1 - 0782 from PK, Rs=8.3Ohm, Cd=219.9pF
  2 - 1139 from PK, Rs=9.9Ohm, Cd=214.3pF
  3 - 0793 from PK, Rs=8.5Ohm, Cd=212.8pF
  4 - 0732 from PK, Rs=7.4Ohm, Cd=214.1pF
  5 - 0791 from PK, Rs=8.4Ohm, Cd=209.9pF
  6 - 0792 from PK, Rs=8.0Ohm, Cd=219.0pF
  7 - 0787 from PK, Rs=9.0Ohm, Cd=197.1pF
  8 - 0790 from PK, Rs=8.4Ohm, Cd=213.1pF
  9 - 0781 from PK, Rs=8.2Ohm, Cd=216.9pF
10 - 0784 from PK, Rs=8.2Ohm, Cd=220.0pF
11 - 1213 from the 40m, Rs=10.0Ohm, Cd=212.9pF
12 - 1208 from the 40m, Rs=9.9Ohm, Cd=216.8pF
13 - 1209 from the 40m, Rs=10.0Ohm, Cd=217.5pF

C30642G, Ls ~ 12nH

20 - 2484 from the 40m EG&G, Rs=12.0Ohm, Cd=99.1pF
21 - 2487 from the 40m EG&G, Rs=14.2Ohm, Cd=109.1pF
22 - 2475 from the 40m EG&G glass crack, Rs=13.5Ohm, Cd=91.6pF
23 - 6367 from the 40m ?, Rs=9.99Ohm, Cd=134.7pF
24 - 1559 from the 40m Perkin-Elmer GH, Rs=8.37Ohm, Cd=94.5pF
25 - 1564 from the 40m Perkin-Elmer GH, Rs=7.73Ohm, Cd=94.5pF
26 - 1565 from the 40m Perkin-Elmer GH, Rs=8.22Ohm, Cd=95.6pF
27 - 1566 from the 40m Perkin-Elmer GH, Rs=8.25Ohm, Cd=94.9pF
28 - 1568 from the 40m Perkin-Elmer GH, Rs=7.83Ohm, Cd=94.9pF
29 - 1575 from the 40m Perkin-Elmer GH, Rs=8.32Ohm, Cd=100.5pF

C30641GH, Perkin Elmer, Ls ~ 12nH

30 - 8983 from the 40m Perkin-Elmer, Rs=8.19Ohm, Cd=25.8pF
31 - 8984 from the 40m Perkin-Elmer, Rs=8.39Ohm, Cd=25.7pF
32 - 8985 from the 40m Perkin-Elmer, Rs=8.60Ohm, Cd=25.2pF
33 - 8996 from the 40m Perkin-Elmer, Rs=8.02Ohm, Cd=25.7pF
34 - 8997 from the 40m Perkin-Elmer, Rs=8.35Ohm, Cd=25.8pF
35 - 8998 from the 40m Perkin-Elmer, Rs=7.89Ohm, Cd=25.5pF
36 - 9000 from the 40m Perkin-Elmer, Rs=8.17Ohm, Cd=25.7pF

Note:
  1mm Au wire with dia. 10um -> 1nH, 0.3 Ohm
20mm BeCu wire with dia. 460um -> 18nH, 0.01 Ohm

Quantum efficiencies of the C30665GH diodes were measured. 

- The diode was biased by the FEMTO preamplifier.

- Diode Pin 1 Signal, Pin 2 +5V, Pin 3 open

- Preamp gain 10^3 V/A

- Beam power was measured by the thorlabs power meter.

PD #1
Incident: 12.82 +/- 0.02 mW
Vout: 9.161 +/- 0.0005 V
PD Reflection (Prompt): 0.404 mW
PD Reflection (Total): 1.168 mW

PD #2
Incident: 12.73 +/- 0.02 mW
Vout: 9.457 +/- 0.0005 V
PD Reflection (Prompt): 0.364 mW
PD Reflection (Total): 0.937 mW

PD #3
Incident: 12.67 +/- 0.02 mW
Vout: 9.1139 +/- 0.01 V
PD Reflection (Prompt): 0.383 mW
PD Reflection (Total): 1.272 mW

PD #4
Incident: 12.71 +/- 0.02 mW
Vout: 9.3065 +/- 0.0005 V
PD Reflection (Prompt): 0.393 mW
PD Reflection (Total): 1.033 mW

PD #5
Incident: 12.69 +/- 0.02 mW
Vout: 9.1071 +/- 0.005 V
PD Reflection (Prompt): 0.401 mW
PD Reflection (Total): 1.183 mW

PD #6
Incident: 12.65 +/- 0.02 mW
Vout: 9.0310 +/- 0.01 V
PD Reflection (Prompt): 0.395 mW
PD Reflection (Total): 1.306 mW

PD #7
Incident: 12.67 +/- 0.02 mW
Vout: 9.0590 +/- 0.0005 V
PD Reflection (Prompt): 0.411 mW
PD Reflection (Total): 1.376 mW

PD #8
Incident: 12.63 +/- 0.01 mW
Vout: 9.0790 +/- 0.0005 V
PD Reflection (Prompt): 0.420 mW
PD Reflection (Total): 1.295 mW

PD #9
Incident: 12.67 +/- 0.02 mW
Vout: 9.2075 +/- 0.0005 V
PD Reflection (Prompt): 0.384 mW
PD Reflection (Total): 1.091 mW

PD #10
Incident: 12.70 +/- 0.01 mW
Vout: 9.0880 +/- 0.001 V
PD Reflection (Prompt): 0.414 mW
PD Reflection (Total): 1.304 mW

PD #11
Incident: 12.64 +/- 0.01 mW
Vout: 9.2861 +/- 0.0005 V
PD Reflection (Prompt): 0.416 mW
PD Reflection (Total): 1.152 mW

PD #12
Incident: 12.68 +/- 0.02 mW
Vout: 9.3650 +/- 0.001 V
PD Reflection (Prompt): 0.419 mW
PD Reflection (Total): 1.057 mW

PD #13
Incident: 12.89 +/- 0.01 mW
Vout: 9.3861 +/- 0.001 V
PD Reflection (Prompt): 0.410 mW
PD Reflection (Total): 1.047 mW

PD serial number
 1 - 0782
 2 - 1139
 3 - 0793
 4 - 0732
 5 - 0791
 6 - 0792
 7 - 0787
 8 - 0790
 9 - 0781
10 - 0784
11 - 1213
12 - 1208
13 - 1209

{
  {1, 12.82, 9.161, 0.404, 1.168},
  {2, 12.73 , 9.457, 0.364 , 0.937} ,
  {3, 12.67 , 9.1139, 0.383 , 1.272 },
  {4, 12.71 , 9.3065, 0.393 , 1.033 },
  {5, 12.69, 9.1071, 0.401 , 1.183 },
  {6, 12.65, 9.0310, 0.395 , 1.306} ,
  {7, 12.67, 9.0590, 0.411 , 1.376} ,
  {8, 12.63 , 9.0790, 0.420 , 1.295} ,
  {9, 12.67 , 9.2075, 0.384 , 1.091} ,
  {10, 12.70, 9.0880, 0.414 , 1.304 },
  {11, 12.64 , 9.2861, 0.416 , 1.152} ,
  {12, 12.68 , 9.3650, 0.419 , 1.057} ,
  {13, 12.89 , 9.3861, 0.410 , 1.047}
};

Yesterday Jeff and Chub worked on the cabling of the OMC. It turned out that the gender of the cable connectors
going from the cavity side to the connector bracket on top of the OMC were opposite from what is needed. 
This way, the connectors can't fixed on the cable harness, thus they are free during the shipping.

We considered several ideas to mitigate this issue and decided to swap the gender of the Mighty Mouse connectors.

In order to check this operation may cause the shortage of the cable length, we made the fitting of the cables.
They seem all long enough for Chub to replace the Mighty Mouse connectors with the proper gender. 

We also checked the polarity of the PZT wires. We marked the positive side of the PZT by a knot at the wire end.

[Koji, Jeff]

Background

In response to the failure of one of the PZTs on L1OMC (LLO:8366), we have been taking place an endurance test of
the four PZT sub-assemblies in prior to their being glued on the glass breadboard.

According to the technical note by Noliac, the common mode of PZT failure is degradation of the impedance
due to cyclic actuation (like 10^7 times) with over voltage. Therefore our procedure of the test to actuate the PZTs
at least 10^7 times with half voltage of the nominal operating voltage (i.e. nominal 200V) and check the degradation
of the impedance.

Driving signal

For the driving of the PZT, a thorlabs HV amp is used. A source signal of 3.5Vpp with an offset of 1.7V is produced
by DS345 function generator. This signal turns to a sinusoidal signal between 0 and 100V in conjunction with the gain
of 15 at the HV amp.

The maximum driving frequency is determined by the current supply limit of the HV amp (60mA). The capacitance
of each PZT is 0.47uF. If we decide to cycle the signal for 4 PZTs in parallel, the maximum frequency achievable
without inducing voltage drop is 100Hz. This yields the test period of 28hours in order to achive 10^7 cycles.

Initial impedance diagnosis

To check the initial state of the PZTs, a DC voltage of 100V was applied via 1kOhm output resistance.
(Note that this output resistance is used only for the impedance test.)
For each PZTs, both side of the resister showed 99.1V for all measurement by a digital multimeter.
Assuming the minimum resolution (0.1V) of the multimeter, the resistance of each PZT was more than 1MOhm before
the cycling test.

Failure detection

In order to detect any impedance drop of the PZTs, the driving signal is monitored on the oscilloscope via a 1:10 probe.
If there is any significant impedance drop, the driver can't provide the driving current correctly. This can be found
by the deviation of the driving voltage from the reference trace on the oscilloscope (below).

Temperature rise

Because of the loss angle of the PZT capacitance, heating of the PZTs is expected. In order to check the temperature rise,
an IR Viewer (FLIR) was used. We did not take care of careful calibration for the PZT emissibity as what we want was a
rough estimation of the temperature.

Before the driving (LEFT) and at the equilibrium (RIGHT)

The temperature change of the PZT was tracked for an hour (below). Fitting of the points indicated that the temperature rise is 2.3degC and the
time constant of 446 sec. This level of temperature rise is totally OK. (Note that the fitting function was T = 27.55 - 2.31 Exp[-t/446.])

Results

DAY1:

Start driving
20:27 25.2 degC, status OK
20:33 26.7 degC, status OK
20:41 26.9 degC, status OK
20:48 27.6 degC, status OK
20:54 27.4 degC, status OK
21:10 27.4 degC, status OK
21:37 status OK
Stop driving

70 minutes of driving (i.e. 4.2x10^5 cycles) => no sign of degradation

DAY2:

Start driving
14:15, 24.5 degC, status OK
14:17, 26.0 degC, status OK
14:24, 27.0 degC, status OK
14:40, 26.8 degC, status OK
14:50, 26.8 degC, status OK
15:30, 26.8 degC, status OK
15:55 status OK
17:40 status OK
21:00 status OK (2.43Mcycles + 0.42Mcycles = 2.85Mcycles)
1d+12:00 status OK (7.83Mcycles + 0.42Mcycles = 8.25Mcycles)
1d+15:00 status OK (8.91Mcycles + 0.42Mcycles = 9.33Mcycles)
1d+18:40 status OK (10.23Mcycles + 0.42Mcycles = 10.65Mcycles)
Stop Driving

After 10.65Mcycles no sign of degradationwas found.

The PZT tests were finished with the conclusion that the PZT won't be damaged with our expected usage.

This is another test of the PZTs to make sure small (~10V) reverse voltage does not break the PZTs.

Background

At the site, we decided to use one of the PZT, which is still alive, for the HV and LV actuation.
The HV actuation is limited to 0 to 100V while the LV actuation is 10Vdc with 1Vpp fast dithering.
This means that a reverse voltage upto 10.5V will be applied to the PZT at the worst case.

From the technical note this level of reverse voltage does not induce polarization of the PZT.
The test is to ensure the PZT is not damaged or degraded by this small reverse voltage.

Method

HV drive: Thorlabs HV amp (G=15) driven with DS345 function generator (3.5Vpp+1.7Vdc, 0.1Hz)
=> 0-100V @0.1Hz
=> The hot side of the potential is connected to the positive side of the PZT

LV drive: Phillips function generator (1Vpp+9.5Vdc@1kHz)
The driving frequency is limited by the current output of the function generator.
=> The hot side of the potential is connected to the negative side of the PZT

These drives shares the common ground.

Tests

Testing with spare PZTs 

Started @19:23 (Aug 23)
Stopped @20:15+2d (Aug 25, duration 48h52m)
17600cycles for the 0.1Hz drive.
176Mcycles for the 1kHz drive.

Checked the impedances of PZT1 and PZT2.

Apply 100Vdc via a 1kOhm resister, 0V detected across the 1kOhm resister
This is equivalent to the resistance of 1MOhm.

Testing with the PZT subassemblies

Started shaking of the four PZT assemblies @20:20 (Aug 25)
No impedance change observed @11:10+1d
No impedance change observed @15:30+1d
Stopped shaking of the four PZT assemblies @XXXX (Aug 26)

Wiring for the test

Rich came to the OMC lab. Pins for the mighty mouse connector were crimped on the 4 PZT wires.
We found the male 4pin mighty mouse connector in the C&B area.

The cable inventory was checked with ICS/DCC combo. It turned out that most of the on-board cables
are at LHO. We decided to send the OMC there and then the cables are installed at the site.

This is not an OMC related and even not happening in the OMC lab (happening at the 40m), but I needed somewhere to elog...

E1400445 first look

LIGO DCC E1400445

Attachment 1: Record of the original state

Attachment 2: Found one of the SMA cable has no shield soldering

Power Supply Board D0900848 was oscillating. Here is the procedure how the issue was fixed.

PCB schematic: LIGO DCC D0900848

0. Extracting the power board.

The top lid and the front panel were removed. Top two modules were removed from the inter-board connection.
Some of the SMA cables were necessary to be removed to allow me to access to the botttom power board.

1. D1~D4 protection diodes

Daniel asked me to remove D1, D2, D3, D4 as the power supply sequence is controlled by the relays.
This was done.

2. Power supply oscillation
Since the power supply systems are entagnled, the oscillation of the transister boosted amps had to be checked one by one.

2.1 VREFP (U5)

First of all, the buffering stage of the positive voltage reference (U5) was oscillating. Attachment 1 is the observed voltage at "VREFP" at D13.
The oscillation was at 580kHz with 400mVpp. This was solved by replacing C20 with 1.2nF. (0805 SMD Cap)

2.2 VREFN (U6)

Then the buffering stage of the negative voltage reference (U6) was checked. Attachment 2 is the observed voltage at "VREFN" at D16.
The oscillation was at 26MHz with 400mVpp. This seemed to have a different mechanism from the U5 oscillation. This oscillation frequency is
higher than the GBW of OP27. So there must be some spurious path to the transister stage. This amplifier stage is a bit unique.
The input is VREFP, but the positive supply is also VREFP. And the feedback path between R31 and C24 is very long. I was afraid that this oscillation
was caused by some combination of L and stray C by the long feedback path and the output to power VREFP coupling, although I could not reproduce
the oscillation on LTSpice. 

After some struggles, adding a 100pF cap between the output of U6 op27 (PIN6) and VREFP (PIN7) stopped the oscillation.
I think this changes the loop function and fullfills the stability condition. I confirmed by a LTSpice model that additional cap does not
screw up the original function of the stage at audio frequencies when everything is functioning as designed.

2.3 Positive supply systems (U10, U11, U12)

Even after fixing the oscillations of U5 and U6, I kept observing the oscllative component of ~600kHz at U10 (+21V), U11 (+15V), and U12 (+5V) stages.
Among them, U11 had the biggest oscillation of 400mVpp at the opamp out (Attachment 3). The other two had small oscillation like 20mVpp at the opamp outputs.
The solution was the same as 2.1. C50, C51, and C52 were replaced to 1.2nF. After the modification I still had the 600kHz component with 2mVpp.
I wanted to check other channels and come back to this.

2.3 Negative supply systems (U7, U8, U9)

Similarly the outputs of U7, U8, and U9 had the oscillation at 600kHz with 40~80mVpp. Once C35, C36, and C37 were replaced with 1.2nF,
I no longer could see any 600kHz anywhere, including U10~U12.

2.4 -24V system (U13)

Last modification was U13. It had a noise of 50mVpp due to piled-up random pulses (Attachment 4). I just tried to replace C63 with 1.2nF
and remove a soldering jumber of W1. There still looks random glitches there. But it's no longer the round shaped pulses but a sharp gliches
and the amplitude is 20mV each (Attachment 5). In fact, later I noticed that Q9 is not stuffed and W2 is closed. This means that the +24V external supply is
directly connected to +24AMP. Therefore U13 has no effect to the 24V suppy system.

3. Restoring all connections / final check of the voltages

Restore the middle and top PCBs to the intra-PCB connector board. Attach the front panel. Restore the SMA connections.

The missing soldering of the SMA cable (reported in the previous entry) was soldered.

Once all the circuits are connected again, the power supply voltages were checked again. There was no sign of oscillation.

All the above modifications are depicted in Attachment 6.

Serial Number of the unit: S1500117
Tester: Koji Arai
Test Date: Jul 22, 2015

1) Verify the proper current draw with the output switch off:

+24 Volt current: 0.08 A Nom.
-24 Volt current: 0.07 A Nom.
+18 Volt current: 0.29 A Nom.
-18 Volt current: 0.24 A Nom.

2) Verify the proper current draw with the output switch on:
+24 Volt current: 0.53 A Nom.
-24 Volt current: 0.07 A Nom.
+18 Volt current: 0.21 A Nom.
-18  Volt current: 0.26 A Nom.

3) Verify the internal supply voltages:

All look good.

TP13 -5.001V
TP12 -15.00
TP11 -21.05
TP10 -10.00
TP5  -18.19
TP6  -24.22
TP2  +24.15
TP3  +18.22
TP9  + 9.99
TP17 +24.15
TP14 +21.04
TP15 +15.00
TP16 + 4.998

4) Verify supply OK logic:
All look good. This required re-disassembling of the PCBs...

Check then pin 5 on U1 (connected to R11) and U4 (connected to R23):

U1 3.68V (=Logic high)
U4 3.68V (=Logic high)

5) Verify the relays for the power supply sequencing: OK

Turn off +/-24 V. Confirm Pin 5 of K1 and K2 are not energized to +/-18V. => OK
Turn on +/-24 V again. Confirm Pin 5 of K1 and K2 are now energized to +/-18V. => OK

6) Verify noise levels of the internal power supply voltages:

TP13 (- 5V) 13nV/rtHz@140Hz
TP12 (-15V) 22nV/rtHz@140Hz
TP11 (-21V) 32nV/rtHz@140Hz
TP10 (-10V) 16nV/rtHz@140Hz
TP9  (+10V)  9nV/rtHz@140Hz
TP14 (+21V) 21nV/rtHz@140Hz
TP15 (+15V) 13nV/rtHz@140Hz
TP16 (+ 5V) 11nV/rtHz@140Hz

Note that the input noise of SR785 is 9~10nV/rtHz@140Hz with -50dBbpk input (AC)

7) Make sure the on/off RF button works,
=> OK

8) Make sure the power output doesn't oscillate,

Connect the RF output to an oscilloscope (50Ohm)
=> RF output: there is no obvious oscillation

Connect the TP1 connector to an oscilloscope
=> check the oscillation with an oscilloscope and SR785 => OK

Connect the CTRL connector to an oscilloscope
=> check the oscillation with an oscilloscope and SR785 => OK

9) EXC check
Connect a function generator to the exc port.
Set the FG output to 1kHz 2Vpk. Check the signal TP1
Turn off the exc switch -> no output
Turn on the exc switch -> nominally 200mVpk

=> OK

10) Openloop transfer function

Connect SR785 FG->EXC TP2->CHA TP1->CHB
EXC 300mV 100Hz-100kHz 200 line

Network Analyzer (AG4395A)
EXC 0dBm TP1->CHA TP2->CHB, measure A/B
801 line
CHA: 0dBatt CHB: 0dBatt
1kHz~2MHz
UGF 133kHz, phase -133.19deg = PM 47deg

11) Calibrate the output with the trimmer on the front panel

13dB setting -> 12.89dBm (maximum setting)

12) Check MON, BIAS and CTRL outputs,
CTRL:    2.95V
MON(L):    6.5mV
BIAS(L):    1.81V
MON(R):    10.7mV
BIAS(R):    1.85V

13) Output check
4+0dB    3.99dBm
6    5.89
8    7.87
10    9.87
12    11.88
14    13.89
16    15.89
18    17.92
20    19.94
22    21.95
24    24.00
26    26.06

4dB+
0.0    3.99
0.2    4.17
0.4    4.36
0.6    4.56
0.8    4.75
1.0    4.94
1.2    5.13
1.4    5.32
1.6    5.53
1.8    5.73
2.0    5.92
2.2    6.10

16dB+
0.0    15.82
0.2    16.11
0.4    16.31
0.6    16.51
0.8    16.72
1.0    16.92
1.2    17.12
1.4    17.32
1.6    17.53
1.8    17.72
2.0    17.92
2.2    18.13

26dB+
0.0    26.06
0.2    26.27
0.4    26.46
0.6    26.58
0.8    26.68
1.0    26.69
1.2    26.70
1.4    3.98
1.6    3.99
1.8    3.99
2.0    3.99
2.2    3.99

(Calibration for Attachment 5 corrected Aug 27, 2015)

Now the test procedure fo the unit is written in the document https://dcc.ligo.org/LIGO-T1500404

And the test result of the first unit (S1500117) has also been uploaded to DCC https://dcc.ligo.org/LIGO-S1500117

Here are some supplimental information with plots

Attachment 1: OLTF of the AM amplitude stabilization servo.

Attachment 2: CLTF/OLTF of the 2nd AM detector self bias adj servo

The secondary RF AM detector provides us the out-of-loop measurement. The secondary loop has an internal control loop to adjust the DC bias.
This loop supresses the RF AM error signal below the control bandwidth. This has been tested by injecting the random noise to the exc and taking
the transfer function between the primary RF AM detector error (MON1) and the secondary one (MON2).

Then the closed loop TF was converted to open loop TF to see where the UGF is. The UGF is 1Hz and the phase margin is 60deg.

Above 10Hz, the residual control gain is <3%. Therefore we practically don't need any compensation of MON2 output above 10Hz.

Attachment 3: Comparison between the power setting and the output power

Attachment 4: Raw power spectra of the monitor channels

Attachment 5: Calibrated in-loop and out-of-loop AM noise spectra

Attachment 6: TFs between BNC monitor ports and DAQ differential signals

BIAS2 and CTRL look just fine. BIAS2 has a gain of two due to the differential output. The TF for CTRL has a HPF shape, but in fact the DC gain is two.
This frequency response comesfro that the actual CTRLis taken after the final stage that has LPF feature while the CTRL DAQ was taken before this final stage.

MON1 and MON2 have some riddle. I could not justify why they have the gain of 10 instead of 20. I looked into the issue (next entry)

Attachment 7: TF between the signals for the CTRL monitor (main unit) and the CTRL monitor on the remote control test rig

The CTRL monitor for the test rig is taken from the CTRL SLOW signal. There fore there is a LPF feature together with the HPF feature described above.
This TF can be used as a reference.

Final Test Result of S1500117: https://dcc.ligo.org/LIGO-S1500117

After some staring the schematic and checking some TFs, I found that the DAQ channels for MON2 have a mistake in the circuit.
Differential driver U14 and U15 of D0900848 are intended to have the gain of +10 and -10 for the pos and neg outputs.
However, the positive output has the gain of +1.

Daniel suggested to shift R66 and R68 by one pad, replace with them by ~5.5K and add a small wire from the now "in air" pad to
the GND near pad 4.

The actual modification can be seen on Attachment 1. The resulting gain was +10.1 as the resisters of 5.49k were used.

The resulting transfer function is found in the Attachment 2. ow the nominal magnitude is ~x20.

You may wonder why the transfer function for MON1 is noisy and lower at low freq (f<1kHz). This is because the input noise of the FFT analizer
contributed to the BNC MON1 signal. High frequency part dominated the RMS of the signal and the FFT analyzer could not have proper range
for the floor noise. The actual voltage noise comparison between the BNC and DAQ signals for MON1 and MON2 can be found in Attachment 3.
 

This is an entry for the work on Aug 3rd.

LIGO DCC E1500151

Power supply check

- Removed the RF AM detector board and exposed the D0900848 power board. The board revision is Rev. A.

- The power supply voltage of +30.2V and -30.5V were connected as +/-31V supplies. These were the maximum I could produce with the bench power supply I had. +17.2V and -17.1V were supplied as +/-17V supplies.

- Voltage reference: The reference voltages were not +/-10V but +/-17V. The cause was tracked down to the voltage reference chip LT1021-10. It was found that the chip was mechanically destroyed (Attachment 1, the legs were cut by me) and unluckily producing +17V. The chip was removed from the board. Since I didn't have any spare LT1021-10, a 8pin DIP socket and an AD587 was used instead. Indeed AD587 has similar performance or even better in some aspects. This fixed the reference voltage.

- -5V supply: After the fix of the reference voltage, I still did not have correct the output voltage of -5V at TP12. It was found that the backpanel had some mechanical stress and caused a leg of the current boost transister cut and a peeling of the PCB pattern on the component lalyer (Attachment 2). I could find some spare of the transister at the 40m. The transister was replaced, and the pattern was fixed by a wire. This fixed the DC values of the power supply voltages. In fact, +/-24V pins had +/-23.7V. But this was as expected. (1+2.74k/2k)*10V = 23.7V .

- VREFP Oscillation: Similarly to the EOM/AOM driver power supply board (http://nodus.ligo.caltech.edu:8080/OMC_Lab/225), the buffer stage for the +10V has an oscillation at 762kHz with 400mVpp at VREFP. This was fixed by replacing C20 (100pF) with 1.2nF. The cap of 680pF was tried at first, but this was not enough to completely elliminate the oscilation.

- VREFN Oscillation: Then, similarly to the EOM/AOM driver power supply board (http://nodus.ligo.caltech.edu:8080/OMC_Lab/225), the amplifier and buffer stage for the -10V has an oscillation at 18MHz with 60mVpp at VREFN. This was fixed by soldering 100pF between pin6 and pin7 of U6.

- Voltage "OK" signal: Checked the voltage of pin5 of U1 and U4 (they are connected). Nominally the OK voltage had +2.78V. The OK voltage turned to "Low (~0V)" when:
The +31V were lowered below +27.5V.
The +31V were lowered below -25.2V.
The +17V were lowered below +15.2V.
The +17V were lowered below -15.4V.

- Current draw: The voltage and current supply on the bench top supplies are listed below

+30.2V 0.09A, -30.5V 0.08A, +17.2V 0.21A, -17.1V 0.10A

- Testpoint voltages:

TP12(-5V) -5.00V
TP11(-15) -14.99V
TP10(-24V) -23.69V
TP9(-10V) -9.99V
TP5(-17V) -17.15V
TP6(-31V) -30.69

TP2(+31V) +30.37V
TP3(+17V) +17.24V
TP8(+10V) +10.00V
TP16(+28V) +28.00V
TP13(+24V) +23.70V
TP14(+15V) +15.00V
TP15(+5V) +5.00V

Entry for Aug 6th, 2015

I faced with difficulties to operate the RF AM detectors.

I tried to operate the RF AM detector. In short I could not as I could not remove the saturation of the MON outputs, no matter how jiggle the power select rotary switches. The input power was 10~15dBm.

D0900761 Rev.A
https://dcc.ligo.org/LIGO-D0900761-v1

I've measured the bias voltage at TP1. Is the bias such high? And does it show this inversion of the slope at high dBm settings?

Setting Vbias
 [dBm]   [V]
   0    21.4
   2    21.0
   4    20.5
   6    19.8
   8    18.8
  10    17.7
  12    16.3
  14    14.2
  16    11.9
  18    10.6
  20    11.8
  22    15.0

The SURF report (https://dcc.ligo.org/LIGO-T1000574) shows monotonic dependence of Vbias from 0.6V to 10V (That is supposed to be the half of the voltage at TP1). 
I wonder I need to reprogram FPGA?

But if this is the issue, the second detector should still work as it has the internal loop to adjust the bias by itself.
TP3 (Page 1 of D0900761 Rev.A) was railed. But still MON2 was saturated.
I didn't see TP2 was also railed. It was ~1V (not sure any more about the polarity). But TP2 should also railed.

Needs further investigation

Spending some days to figure out how to program CPLD (Xilinx CoolRunner II XC2C384).

I learned that the JTAG cable which Daniel sent to me (Altium JTAG USB adapter) was not compatible with Xilinx ISE's iMPACT.

I need to use Altium to program the CPLD. However I'm stuck there. Altium recognizes the JTAG cable but does not see CPLD. (Attachment 1)

Upon the trials, I followed the instruction on awiki as Daniel suggested.
http://here https://awiki.ligo-wa.caltech.edu/aLIGO/TimingFpgaCode

Altium version is 15.1. Xilinx ISE Version is 14.7

Still suffering from a power supply issue!

I have been tracking the issues I'm having with the RF AM detector board.

I found that the -5V test point did not show -5V but ~+5V! It seemed that this pin was not connected to -5V but was passive.

I removed the RF AM detector board and exposed the power board again. Pin 11 of P3 interboard connector indeed was not connected to TP12 (-5V). What the hell?

As seen in the attached photo, the PCB pattern for the Pin 11 is missing at the label "!?" and not driven. I soldered a piece of wire there and now Pin11 is at -5V.

This fix actually changed several things. Now the bias setting by the rotary switches works.

Setting BIAS1
 [dBm]   [V]
   0    0.585
   2    0.720
   4    0.897
   6    1.12
   8    1.42
  10    1.79
  12    2.25
  14    2.84
  16    3.60
  18    4.56
  20    5.75
  22    7.37

This allows me to elliminate the saturation of MON1 of the first RF AM detector. I can go ahead to the next step for the first channel.

Now the bias feedback of the second detector is also behaving better. Now TP2 is railing.

Still MON2 is saturated. So, the behavior of the peak detectors needs to be reviewed.

We wanted to know the  transimpedance of the OMC DCPD at high frequency (1M~10M).
For this purpose, the OMC DCPD chain was built at the 40m. The measurement setup is shown in Attachment 1.

- As the preamp box has the differential output (pin1 and pin6 of the last DB9), pomona clips were used to measure the transfer functions for the pos and neg outputs individually.

- In order to calibrate the measurements into transimpedances, New Focus 1611 is used. The output of this PD is AC coupled below 30kHz.
This cutoff was calibrated using another broadband PD (Thorlabs PDA255 ~50MHz).

Result: Attachment 2
- Up to 1MHz, the transimpedance matched well with the expected AF transfer function. At 1MHz the transimpedance is 400.

- Above 1MHz, sharp cut off at 3MHz was found. This is consistent with the openloop TF of LT1128.

The noise levels of the output pins (pin1/pin6) are measured. Note that the measurement is done with SE. i.e. There was no common mode noise rejection.

Worked on the calibration of the RF AM Measurement Unit.

The calibration concept is as follows:

• Generate AM modulated RF output
• Measure sideband amplitude using a network anayzer (HP4395A). This gives us the SSB carrier-sideband ratio in dBc.
• Measure the output of the RF AM measurement unit with the same RF signal
• Determine the relationship between dBc(SSB) and the output Vrms.

The AM modulated signal is produced using DS345 function generator. This FG allows us to modulate
the output by giving an external modulation signal from the rear panel. In the calibration, a 1kHz signal with
the DC offset of 3V was given as the external modulation source. The output frequency and output power of
DS345 was set to be 30.2MHz (maximum of the unit) and 14.6dBm. This actually imposed the output
power of 10.346dBm. Here is the result with the modulation amplitude varied

                 RF Power measured             Monitor output
Modulation       with HP4395A (dBm)           Measure with SR785 (mVrms)
1kHz (mVpk)   Carrier    USB      LSB          MON1       MON2
   0.5        9.841    -72.621  -73.325          8.832      8.800
   1          9.99     -65.89   -65.975         17.59      17.52
   2          9.948    -60.056  -59.747         35.26      35.07
   3          9.90     -56.278  -56.33          53.04      52.9
   5          9.906    -51.798  -51.797         88.83      88.57
  10          9.892    -45.823  -45.831        177.6      177.1
  20          9.870    -39.814  -39.823        355.3      354.4
  30          9.8574   -36.294  -36.307        532.1      531.1
  50          9.8698   -31.86   -31.867        886.8      885.2
 100          9.8735   -25.843  -25.847       1772       1769
 150          9.8734   -22.316  -22.32        2656       2652
 200          9.8665   -19.819  -19.826       3542       3539
 300          9.8744   -16.295  -16.301       5313       5308

The SSB carrier sideband ratio is derived by SSB[dBc] = (USB[dBm]+LSB[dBm])/2 - Carrier[dBm]

This measurement suggests that 10^(dBc/20) and Vrms has a linear relationship. (Attachment 1)
The data points were fitted by the function y= a x.

=> 10^dBc(SSB)/20 = 108*Vrms (@10.346dBm input)

Now we want to confirm this calibration.

DS345 @30.2MHz was modulated with the DC offset + random noise. The resulting AM modulated RF was checked with the network analyzer and the RFAM detector
in order to compare the calibrated dBc/Hz curves.

A) SR785 was set to produce random noise
B) Brought 2nd DS345 just to produce the DC offset of -2.52V (Offset display -1.26V)
Those two are added (A-B) by an SR560 (DC coupling, G=+1, 50 Ohm out).
The output was fed to Ext AM in DS345(#1)

DS345(#1) was set to 30.2MHz 16dBm => The measured output power was 10.3dBm.

On the network analyzer the carrier power at 30.2MHz was 9.89dBm

Measurement 1) SR785     1.6kHz span 30mV random noise (observed flat AM noise)
Measurement 2) SR785   12.8kHz span 100mV random noise (observed flat AM noise)
Measurement 3) SR785 102.4kHz span 300mV random noise (observed cut off of the AM modulation due to the BW of DS345)

The comparison plot is attached as Attachment 2. Note that those three measurements are independent and are not supposed to match each other.
The network analyzer result and RFAM measurement unit output should agree if the calibration is correct. In fact they do agree well.

In order to check the noise level of the RFAM detector, the power and cross spectra for the same signal source
were simultaneously measured with the two RFAM detectors.

As a signal source, 37MHz OCXO using a wenzel oscillator was used. The output from the signal source
was equaly splitted by a power splitter and fed to the RFAM detector CHB(Mon1) and CHA(Mon2).

The error signal for CHB (Mon1) were monitored by an oscilloscope to find an appropriate bias value.
The bias for CHA are adjusted automatically by the slow bias servo.

The spectra were measured with two different power settings:

Low Power setting: The signal source with 6+5dB attenuation was used. This yielded 10.3dBm at the each unit input.
The calibration of the low power setting is dBc = 20*log10(Vrms/108). (See previous elog entry)

High Power setting: The signal source was used without any attenuation. This yielded 22.4dBm at the each unit input.
The calibration for the high power setting was measured upon the measurement.
SR785 was set to have 1kHz sinusoidal output with the amplitude of 10mVpk and the offset of 4.1V.
This modulation signal was fed to DS345 at 30.2MHz with 24.00dBm
The network analyzer measured the carrier and sideband power levels
30.2MHz 21.865dBm
USB    -37.047dBm
LSB    -37.080dBm  ==> -58.9285 dBc (= 0.0011313)

The RF signal was fed to the input and the signal amplitude at Mon1 and Mon2 were measured
MON1 => 505   mVrms => 446.392 Vrms/ratio
MON2 => 505.7 mVrms => 447.011 Vrms/ratio
dBc = 20*log10(Vrms/446.5).

Using the cross specrum (or coherence)of the two signals, we can infer the noise level of the detector.

Suppose there are two time-series x(t) and y(t) that contain the same signal s(t) and independent but same size of noise n(t) and m(t)

x(t) = n(t) + s(t)
y(t) = m(t) + s(t)

Since n, m, s are not correlated, PSDs of x and y are

Pxx = Pnn + Pss
Pyy = Pmm+Pss = Pnn+Pss

The coherence between x(t) and y(t) is defined by

Cxy = |Pxy|^2/Pxx/Pyy = |Pxy|^2/Pxx^2

In fact |Pxy| = Pss. Therefore

sqrt(Cxy) = Pss/Pxx

What we want to know is Pnn

Pnn = Pxx - Pss = Pxx[1 - sqrt(Cxy)]
=> Snn = sqrt(Pnn) = Sxx * sqrt[1 - sqrt(Cxy)]

This is slightly different from the case where you don't have the noise in one of the time series (e.g. feedforward cancellation or bruco)

Measurement results

Power spectra:
Mon1 and Mon2 for both input power levels exhibited the same PSD between 10Hz to 1kHz. This basically supports that the calibration for the 22dBm input (at least relative to the calibration for 10dBm input) was corrected. Abobe 1kHz and below 10Hz, some reduction of the noise by the increase of the input power was observed. From the coherence analysis, the floor level for the 10dBm input was -178, -175, -155dBc/Hz at 1kHz, 100Hz, and 10Hz, respectively. For the 22dBm input, they are improved down to -188, -182, and -167dBc/Hz at 1kHz, 100Hz, and 10Hz, respectively.

TEST Result: S1500118

Additional notes

- Checked the power supply. All voltages look quiet and stationary.

- Checked the internal RF cables too see if there is any missing shield soldering => Looked fine

- Noticed that the RFAM detector board has +/-21V for the +/-24V lines => It seems that this is nominal according to the schematic

- Noticed that the RFAM detector sensitivity were doubled fomr the other unit.
  => This is reated to the modification (E1500353) of  "Controller Board D0900761-B Change 1" (doubling the monitor output gain)

- Noticed that the transfer function of the CTRL signal on the BNC and the DAQ output.
  => This is reated to the modification (E1500353) of  "Servo Board D0900847-B Change 1"  (servo transfer function chage)
  => The measured transfer function did not agree with the prediction from the circuit constants in this document
  => From the observation of the servo board it was found that R69 was not 200Ohm but 66.5 Ohm (See attachment 1).
       This explained the measured transfer function. The actuator TF has: P 2.36, Z 120., K -1@DC (0.020@HF)

- Similarly, the TF between the CTRL port on the unit and the CTRL port on the test rig was also modified.

Noise level

Attachment 2

- The amplitude noise in dBc (SSB) was measured at the output of 27dBm. From the test sheet, the noise level with 13dBm output was also referred. From the coherence of the MON1 and MON2, the noise level was inferred. It suggests that the floor level is better than 180dBc/Hz. However, there is a 1/f like noise below 1k and is dominating the actual noise level of the RF output. Daniel suggested that we should check nonlinear downconversion from the high frequency noise due to the noise attenuator. This will be check with the coming units.

Test sheet: https://dcc.ligo.org/LIGO-E1400445

Test Result (S1500114): https://dcc.ligo.org/S1500114

Linearity of the EOM driver was tested. This test has been done on Nov 10, 2015.

- Attachment 1: Output power vs requested power. The output start to deviate from the request above 22dBm request.

- Attachment 2: Ctrl and Bias voltages vs requested power. This bias was measured with the out-of-loop channel.
The variable attenuator has the voltage range of 0~15V for 50dB~2dB attenuation.

Therefore this means that:

- The power setting gives a voltage logarithmically increased as the requested power increases. And the two power detectors are watching similar voltages.

- And the servo is properly working. The control is with in the range.

- Even when the given RF power is low, the power detectors are reporting high value. Is there any mechanism to realize such a condition???

This measurement has been done on Dec 1st, 2015.

The phase noise added by the EOM driver was tested.

The test setup is depicted in the attached PDF. The phase of the RF detector was set so that the output is close to zero crossing as much as possible with the precision of 0.5ns using a switchable delay line box. The phase to voltage conversion was checked by changing the delayline by 1ns. This gave me somewhat larger conversion factor compared to the sine wave test using an independent signal generator. This was due to the saturation of the phase detector as the LO and RF both have similar high RF level for the frequency mixer used.

The measurement has done with 1) no EOM driver involved, 2) one EOM driver inserted in the RF path, and 3) EOM drivers inserted in both the LO and RF paths.

I could not understand why the measurement limit is so high. Also the case 2 seems too low comsidering the noise level for 1) and 3).

At least we could see clear increase of the noise between the case 1) and 3). Therefore, we can infer the phase noise added by the EOM driver from the measurements.

Note: The additional phase noise could be associated with the original amplitude noise of the oscillator and the amplitude-to-phase conversion by the variable attenuator. This means that the noise could be corellated between two EOM drivers. The true test could be done using a PLL with a quiet VCO. Unfortuantely I don't have a good oscillator sufficient for this measurement.

Dark current of the HQE PD and other PDs were measured.

- The HQE PDs were loaded on the new PD transportation cages (Attachment 1)
The PDs are always shorted by a clean PD plugs. The PD element is still capped with Kapton seals.

- The assignment of the container/slot and the PDs are as follows

- The measurement has been done with KEITHLEY sourcemeter SMU2450.

- The result is shown in Attachment 2. Most of the PDs show the dark current of ~3nA at 15V bias. C1-05 and C1-07 showed higher dark current at high V region. We should avoid using them for the aLIGO purpose. I hope they are still OK at low bias V if there is no noise issue (TBC). You can not read the PD names on the plot for the nominal ones, but that's OK as they are almost equivalent.

- As a comparison, the dark current of a C30655 (serial #10) was measured. Considering a DC current due to an anbient light (although the PD was covered), the dark current of the HQE PD seems double of C30655.

- Taking an advantage of having the setup, I took the same measurement for the Laser Comp. PDs in ATF. I gave the identification as #1 and #2. #1 has full-length legs while #2 has trancated legs. As Zach reported before, they showed significantly high dark current. (Attachment 3)

Transfered for RGA scan

B4 (C1-05) -> F1
C1 (C1-07) -> F2

Dark noise measurement for 6 HQEPDs and 1 C30665. All of these showed sufficiently low dark current noise levels compared with the noise level of the DCPD preamp. The measurement was limited by the input noise (ADC) noise of the FFT analyzer as the line noises were too big.

The measurement has been done with the transimpedance of 1e7. The bandwidth of the measurement was 50kHz.

For the production of the aLIGO PDs, the following transfer of the PDs were carried out
A1-23 Cage A1 -> G1
A1-25 Cage A2 -> G2

The cage A will be baked at 75degC to see if this improves AMU=64 emission.

At the same time, we will put C1-05 (F1) and C1-07 (F2) into the contamination test cavity.

Possible reduction of the QE was observed after air-bake at 75degC.

Yesterday I received Cage G from Bob for intermediate test of the PD performance after air bake but before vacuum bake.
This cage was prepared to be the production pair.

According to the ICS, https://ics-redux.ligo-la.caltech.edu/JIRA/browse/Bake-8047
the PDs were air baked at 75degC for 48 hours.

I took the PDs to my lab to check if there is any issue in terms of the performance.
- Dark current: No change observed
- Dark noise: No noise increase observed
- QE: Probably reduced by ~0.5%.

Here I attached the result of the QE measurement. I have measured the QEs of the baked ones (A1-23 and A1-25) and the reference. Since the reference PD has not been baked, this gives us the measure of the systematic effect. The reference showed the reduction of ~0.1%. Assuming this reduction came from the systematic effect of the measurement system, I observed at least 0.5% QE reduction (A1-23). Note that the previous measurement of 99.8% for A1-25 was too high and dubious. But both A1-23 and A1-25 showed ~0.4% lower QEs.

So I believe the air-baking process reduced the QE.

Another evidence was that now I could clearly see the beam spots on these air-baked-PDs with an IR viewer when the PDs were illuminated with a 1064nm beam. Usually it is difficult to see the spot on the PD. The spot on the reference PD was still dark. So this difference was very obvious. I was afraid that something has been deposited on the surface of the photosensitive element. The surface of the diodes looked still very clean when they were checked with a green LED flash light.

All survived PDs have been measured.

Attachment 1: PDH amp RF part (before the preamp was installed)

Attachment 2: RF-AF transmission

Attachment 3: Attachment 3: LO dependence

Attachment 4: RF amp gain (saturation)

Attachment 5: Input/output noise level

Attachment 6: Attachment 6: Preamp/DCPD out buffer AF circuit

attachment 6: DCPD preamp looks like the opamp is wired for positive feedback?

The polarities indicated in the right circuits were opposite, obviously.

[Rich Koji]

3IFO EOM (before any modification) was tested to measure the impedance of each port.

The impedance plot and the impedance data (triplets of freq, reZ, imZ) were attached to this entry.

[Rich Koji]

The impedances of the individual components from the 3IFO EOM (before modification) were tested.
Each component was modeled by LISO. The LISO model (in PDF and txt) are attached at the end of the entry.

Coils
There are three inductors taken from the EOM unit. They showed the Q ranging from 150~300.
Their impedances are compared with the coil taken from the 9MHz port of the spare EOM (=current LHO EOM).
The inductance of the 8.7MHz inductor indicated higher L but still higher Q.

Todd made a replica of the 45.3MHz coil. He used a silver plated wire and it actually showed highest Q of ~400.

Crystal capacitance
The crystal capacitances were measured by attaching a test rig on the DB15 connector of the crystal housing. The rig was calibrated such that the impedances of the attched components on the rig were measured. They showed somewhat similar feature with parasitic resonances at ~50MHz. Above this frequnecy the capacitance went down (i.e. Abs(Z) went up). This indicates there are stray series LCR in pararrel to the crystal. Not sure what is the cause of this.

The central (24.1MHz) port showed smaller capacitance. This probably means the plates for the central port is smaller. Not sure the actual dimensions of the plates for this unit.
 

Koji, Rich

We took apart the unit removed from the 3rd IFO (Unit serial number aLIGO #3, XTAL 10252004) to see what makes it tick.  Koji has done a fine job of adding the plots of the impedance data to this log book.  Attached are some details of the physical construction showing the capacitor values used in shunt before the coils.

Attached please see my notes summarizing the models for the electrodes and inductors within the 3rd IFO EOM

The circuit models for the 3IFO EOM (before mods) were made using LISO.
Then the modification plan was made to make it a new LLO EOM.

Impedance data, LISO model, Mathematica files are zipped and attached at the end.