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ID Date Author Type Category Subjectup
  12835   Thu Feb 16 21:55:47 2017 ranaSummaryGeneralAlternative Calibration Scheme

Question for Craig: What does the SNR of our lines have to be? IF we're only trying to calibrate the actuator in the audio band over long time scales, it seems we could get by with more frequency noise. Assuming we want a 1% calibration at 50-500 Hz, what is the requirement on the frequency noise PSD curve?

  12842   Tue Feb 21 13:51:35 2017 CraigSummaryGeneralAlternative Calibration Scheme

We get SNR in two ways: the amplitude of applied force and the integration time.  So we are limited in two ways: stability of the lock to applied forces and time of locklosses / calibration fluctuations.

At the sites, you probably know that we blow our spectrum out of the water with the calibration lines, with SNRs of about 100 on the scale of about 10 seconds.  For us this might be impossible, since we aren't as quiet.

If we want 1% calibration on our sweeps, we'll need  0.01 = Uncertainty = sqrt( (1 - COH^2)/(2 * Navg * COH^2) ), where COH is the coherence of the transfer function measurement and Navg is the number of measurements at a specific frequency.  This equation comes from Bendat and Piersol, and is subject to a bunch of assumptions which may not be true for us (particularly, that the plant is stationary in time).

If we let Navg = 10, then COH ~ 0.999.

Coherence = Gxy^2/(Gxx * Gyy), where x(t) and y(t) are the input signal and output signal of the transfer function measurement, Gxx and Gyy are the spectral densities of x and y, and Gxy is the cross-spectral density.  

Usually SNR = P_signal / P_noise, but for us SNR = A_signal / A_noise.

Eric Q and Evan H helped me find the relationship between Coherence and SNR:

P = Pn + Pc, Pn = P * (1 - Coh), Pc = P * Coh

==> SNR = sqrt( Pc / Pn ) = sqrt( Coh / 1 - Coh )

From Coh ~ 0.999, SNR ~ 30.


Question for Craig: What does the SNR of our lines have to be? IF we're only trying to calibrate the actuator in the audio band over long time scales, it seems we could get by with more frequency noise. Assuming we want a 1% calibration at 50-500 Hz, what is the requirement on the frequency noise PSD curve?


  12845   Wed Feb 22 10:16:54 2017 ranaSummaryGeneralAlternative Calibration Scheme

OK, but the questions still stands: "Assuming we want a 1% calibration at 50-500 Hz, what is the requirement on the frequency noise PSD curve?"


We get SNR in two ways: the amplitude of applied force and the integration time.  So we are limited in two ways: stability of the lock to applied forces and time of locklosses / calibration fluctuations.

  11981   Mon Feb 8 15:36:37 2016 gautamUpdateGreen LockingAlternative mode-matching scheme

I looked in the optics cabinet to see what lenses we have available, and re-ran the mode-matching calculation to see if we could find a better solution - I'm attaching a plot for what looks like a good candidate (optimized mode-matching efficiency for the X mode is 100%, and for the Y mode, it is 97.98%), though it does involve switching "L1", which is currently a 175mm efl lens, for a 125mm efl lens. I've also indicated on the plot where the various other components are relative to the optimized positions of the lens, and it doesn't look like anything is stacked on top of each other. Also, the beam width throughout is well below 4.7mm, which is the maximum cited width the Faraday can handle, as per its datasheet. "L1" doesn't quite get the waist of the beam to coincide with the geometrical center of the Faraday, but I don't think this is requried? Also, I've optimized the mode matching using the measured X width of the beam (red curve in Attachment #1), and have overlaid the calculated Y width of the beam for the optimized position of the lenses (red curve in Attachment #1). The target waist was 35um at the center of the doubling oven, which the X profile achieves, but the Y profile has a width of 32 um at the same point.

In all the calculations, I've not accounted for possible effects of the HWPs and the Faraday on the beam profile....

  3981   Tue Nov 23 22:45:59 2010 kiwamuUpdateComputersAltium

I installed and activated Altium, a PCB design software, on the Windows machine M2.

With Altium I am going to design the triple resonant circuit for the broadband EOM.

  1476   Sun Apr 12 19:31:43 2009 ranaSummaryElectronicsAmphony 2500 Headphones
We bought the Amphony 2500 Digital Wireless headphones recently. The other cheapo headphones we have are OK for control room use, but have a lot of noise
and are, therefore, not useful for noise hunting.

The new digital ones are pretty much noise-free. With the transmitter next to rosalba, you can walk halfway down the east arm and all around the MC area
before the reception goes bad. For real noise hunting, we will want to put the transmitter next to the BS chamber and take an analog pickoff from the DC PDs.

In the OMC diagram, we should put an AUDIO filterbank and wire it to the DAC so that we can do arbitrary IIR filtering on the audio signal.
  14482   Sun Mar 17 21:06:17 2019 AnjaliUpdateALSAmplifier characterisation

The goal was to characterise the new amplifier (AP1053). For a practice, I did the characterisation of the old amplifier.This test is similar to that reported in Elog ID 13602.

  • Attachment #1 shows the schematic of the setup for gain characterisation and Attachment #2 shows the results of gain characterisation. 
  • The gain measurement is comparable with the previous results. From the data sheet, 10 dB gain is guaranteed in the frequency range 10-450 MHz. From our observation, the gain is not flat pver this region. We have measured a maximum gain of 10.7 dB at 6 MHz and it has then decreased upto 8.5 dB at 500 MHz
  • Attachement #3 shows the schematic of the setup for the noise characterisation and Attachment # 4 shows the results of noise measurment. 
  • The noise measurement doesn't look fine. We probably have to repeat this measurement.
  9991   Sat May 24 22:56:57 2014 JenneUpdateElectronicsAmplifier removed from BeatX path

I just realized that I forgot to elog this, but yesterday afternoon I bypassed the amplifier in the BeatX path, and now the X beatnote is about -27dBm.  Arms lock nicely with ALS.

  9992   Mon May 26 07:59:23 2014 KojiUpdateElectronicsAmplifier removed from BeatX path

And the out-of-loop level of the ALSX compared with the previous measurement is ...?


I just realized that I forgot to elog this, but yesterday afternoon I bypassed the amplifier in the BeatX path, and now the X beatnote is about -27dBm.  Arms lock nicely with ALS.


  9995   Tue May 27 11:58:45 2014 JenneUpdateElectronicsAmplifier removed from BeatX path

Sorry, I had been in a hurry when I worked on this last week, and again when I wrote the elog, but I wanted to at least put in a note for any weekend workers.

The ALS beatnote setups need alignment on the PSL table.  However, even at very low RF beat frequency, the X beatnote now at low frequencies matches our best measurement from last week.  The "HEPA off" (teal and purple) measurements are from last week, and the red and blue are from this week.  The X beatnote was 10MHz and the Y beatnote today was 31MHz.


  5156   Tue Aug 9 16:00:58 2011 JennyUpdatePSLAmplitude response of PZT


The top plot shows a sweep from 10 kHz to 5 MHz of the ratio of the voltage output of the PD detecting power from the NPRO laser beam and the RF source voltage (the magnitude of the complex transfer function). The black trace was taken with the laser beam blocked. For runs 2 and 3 I changed the laser temperature set point by 10 mK and 100 mK respectively to see if there was a significant change in the AM response. The bottom plots shows runs 2 and 3 compared to run 1 plotted in dB (to be explicit, i'm plotting 10 times the base 10 log of the magnitude of the ratio of two complex transfer functions). Changing the temperature seems to have only a minor effect on the output except at around 450kHz, where the response has a large peak in run 1 and much smaller peaks in runs 2 and 3. 

The traces in the top plot consist of 16 averages taken with a 300Hz IF bandwidth, 15 dBm source power (attenuated with a 6 dB attenuator) and with 20dB attenuation of the input power from the PD.

Next I'm going to probe a narrow band region where the response is low (2.0MHz or 2.4MHz perhaps) and choose a bandwidth for the dither frequency for the PDH locking.

  15457   Mon Jul 6 17:41:19 2020 gautamUpdateLSCAn LSC puzzler

Last Tuesday evening, while attempting the PRFPMI locking, I noticed a strange feature in the LSC signals, which is shown in Attachment #1 (the PDF exported by dataviewer is 14MB so I upload the jpeg instead). As best as I can tell, the REFL33 and POP22 channels show an abrupt jump in the signal levels, while the other channels do not. POP110 shows a slight jump at around the same time, and the large excursion in AS110_Q actually occurs a few seconds later, and is probably some angular excursion of the PRC/BS. I'm struggling to interpret how this can be explained by some interferometric mechanism, but haven't come up with anything yet. The LO for the 3f error signals is the 2f field, but then why doesn't the POP110 channel show a similar jump if there is some abrupt change in the resonant condition? Is such a change even feasible from a cavity length change point of view? Or did the sideband frequency somehow abruptly jump? But if so, why is the jump much more clearly visible in one sideband than the other?

Does anyone have any ideas as to what could be going on here? This may give some clue as to what's up with the weird sensing matrices, but may also be something boring like broken electronics... 

  10093   Tue Jun 24 16:52:43 2014 NichinUpdateElectronicsAn RF cable re-installed


 [Nichin, Eric G]

As mentioned in Elog 10062, we found RF cables running between demodulators in rack 1Y2 and RF switch in 1Y1 to have bad SMA connectors (No shield / bad soldering / no caps).

we pulled out all the cables belonging to PD frequency response measurement system , 8 in total, and all of them about 5.5m in length.

Their labels read :

REFL33, REFL11, REFL55, AS55, POX11, REFL165, POP22 and POP110. 

All of them are now sitting inside a plastic box in the contorl room.

On another note, instead of fixing all the cables ourselves, Steve and Eric G decided to order custom made RF cables from Pasternack as professionally soldered cables are worth it. We have placed an order for 2 cables (RG405-550CM) to check out  and test them before we order all of the cables.

 The new RF cables arrived. But unfortunately we did not realize that RG405 was a Semi-rigid coax cable, with a copper shielding. These are meant to be installed in setups that will not be changed / disturbed. We need to order a different set of cables. The new cables have joined the other cables in the plastic box mentioned above.

For now to check if the old setup is still working, I have installed an RF cable (that we earlier pulled out and looks like in good shape, labelled REFL33) between the AS55 Demodulator output PD RF MON in rack 1Y2 and the network analyzer input. Since Manasa and the others were busy working with the interferometer, I did not switch on the laser and did not take any readings. The power supply to REF DET remains off.

I will continue with the measurements tomorrow morning and also try to get the data wirelessly using Alex's code. 

  10667   Tue Nov 4 19:17:53 2014 ericqUpdateComputer Scripts / ProgramsAnaconda + CDSutils

I've fallen down the rabbit hole of trying to reconcile our desire for newer versions of the Numpy and Scipy python packages with the use of our handy cdsutils tools. 

I've set up an installation of Anaconda python in /ligo/apps/anaconda. Installing pyepics, nds2, and cdsutils was straightforward, but there were a myriad of odd python packages that cdsutils depends on, that are typically installed at the OS level (python-gst, gobject, glib) which I just manually copied over to the anaconda directories. Also, the version of readline that anaconda ships with is somewhat borked (dark voodoo fix was found here: github link. The issue mentioned there wasn't why I needed the fix. Somehow libreadline was causing pyepics initialization to fail). 

I was initially hoping this kind of exercise would be useful, as having a separate python environment that we control buffers us from the system installation and allows us to use whatever version of packages we want, but the amount of hackery I did to get to get cdsutils to work probably didn't result in the most robust solution. (Maybe there was a better way!)

In any case, I have not changed any of our machines' default paths or environment variables. Instead, I have simply created an alias that points to Anaconda python: "apython"


controls@pianosa|scriptTesting > cat foo.py
import scipy as sp
import sys
from ezca import Ezca
print 'Python Version: '+ sys.version
print 'ez.read test:' + str(ez.read('LSC-TRY_OUT16'))
print 'Scipy Version: '+sp.__version__
controls@pianosa|scriptTesting > python foo.py
Python Version: 2.7.3 (default, Feb 27 2014, 19:58:35)
[GCC 4.6.3]
ez.read test:0.0154613731429
Scipy Version: 0.9.0
controls@pianosa|scriptTesting > apython foo.py
Python Version: 2.7.8 |Continuum Analytics, Inc.| (default, Aug 21 2014, 18:22:21)
[GCC 4.4.7 20120313 (Red Hat 4.4.7-1)]
ez.read test:0.00307549210265
Scipy Version: 0.14.0

Thus, Diego should now be able to complete his script that needs the newer Scipy, as well as CDSutils. 

Final note: I've tested z (read|write|avg) with $PATH modified to have /ligo/apps/anaconda/bin at the start, and they seem to work. If things seem to hold up, maybe we can replace the default command-line python, but its not strictly necessary. 

  10688   Sat Nov 8 11:31:51 2014 ranaUpdateComputer Scripts / ProgramsAnaconda + CDSutils


I've fallen down the rabbit hole of trying to reconcile our desire for newer versions of the Numpy and Scipy python packages with the use of our handy cdsutils tools. 

 Avoid rabbit holes! What I did at LLO which works is to install an Anaconda in some shared directory and then just make an alias which puts that directory at the head of the path when running the more advanced SciPy installs. It works fine and cannot interfere with our usual operation since its only sourced when running peak find.

  14923   Wed Oct 2 10:50:20 2019 gautamUpdateCDSAnaconda updated

The anaconda distribution used by the control room workstations is actually installed on the shared drive (/cvs/cds/ligo/apps/anaconda/) for consistency reasons. The version was 4.5.11. I ran the following commands to update it today. Now it is version 4.7.12.

conda update conda
conda update anaconda

The second command takes a while to resolve conflicts, so I've left it running inside a tmux session for now.

Recall that the bash alias for using the anaconda managed python is "apython". I recommend everyone set up a virtual environment when trying out new package installs, to avoid destroying the locking scripts.

  4254   Sat Feb 5 23:03:04 2011 rana, kojiSummaryElectronicsAnalog Frequency Discriminator: splitter + mixer + long cable

This diagram shows the setup of the analog Mixer-Frequency Discriminator (MFD).

The idea is similar to the one of the Schnupp Asymmetry for our Michelson interferometers. The signal from the PD (or any signal source for which you want to know the frequency) is split into two legs; one leg is much longer than the other. The two legs are recombined at a mixer/demodulator. The demodulator output varies sinusoidally with the phase difference of two legs, the same as when we try to measure the phase noise of an oscillator, for example. This is the same concept as the digital frequency discriminator that Aidan and Joe put into the GFD FE system recently.

With a ~1m cable length asymmetry, we get 180 deg of phase shift for a ~100 MHz signal (recall that the speed of light in most of our cables is ~2 x 10^8 m/s). The mixer gives a linear output at 50 MHz (and 150 MHz, 250 MHz, etc.).

This single mixer based setup is fine for most everything we do. In order to get even more resolution, one can just use 2 mixers by splitting the signal with a 4-way instead of 2-way mixer. One setup can have a 0.5-1 m asymmetry to have a large range. The other can have a ~10-30m asymmetry to get a comb of linear readouts.

Typically, we will have some kind of weak signal at the photodiode and will use a 20 or 40 dB gain RF amp to get the signal into the mixer. In this case, the mixer output noise will be at the level of tens of nV/rHz. Any usual low noise audio amplifier (SR560 variety) will be enough to read out the signal.

Why the 50 Ohm terminator? If you look at the specs of the BLP-1.9 filter from Mini-Circuits (its the same for almost all of their LP filters) you see that there's ~90 dB of attenuation above ~30 MHz (where our signals 2*f product will show up). If we use an RF input signal of ~0 dBm, this means that we get a high frequency product of -95 dBm (~10 uVrms) which is OK. But the return loss is 0 dB above 5 MHz - this means that all of the high frequency content is reflected back into the mixer! The 50 Ohm terminator is there to absorb the RF signals coming out of the mixer so as to prevent them from going back into the mixer and mixing with the RF/LO signals. The 50 Ohm terminator does attenuate the DC/audio frequency signals we get out of the mixer by a factor of two, but that's OK since we are not limited by the mixer's thermal noise.

Noise Measurement:

To checkout the noise, we used a 6m RG-58 cable in one leg. We used the DS345 signal generator for the source. We adjusted the frequency to (~21 MHz) give a ~zero mean signal at the demod output. The 6m cable makes the demod output's peak-peak swing correspond to ~16 MHz. We then used an SR560, DC coupled, G=1000, low-noise, 2pole low pass at 1 kHz, to get the signal into the ADC.


The attached plot shows the noise. We have caibrated the digital gain in the channel to make the output into units of Hz. The high frequency noise floor is ~0.3 Hz/rHz and the 1/f knee is at 10 Hz. This setup is already good enough for all of the green locking work at the 40m. In order to make this useful for the reference cavity work or the gyro, we will have to use a longer cable and a lower noise audio amplifier.

As can be seen from the plot, the ADC noise is below the measured noise. The noise of the SR560 with the input terminated is shown in grey - the measured noise of the MFD is very close to this. In order to improve the performance, the next step should be to use a longer cable. There's clearly going to be some trade-off between the temperature dependent effects which come with long cables (dphi/dT gets bigger) and trying to use a high gain ~1 nV/rHz amplfier at the mixer output.

Temperature Drift of the long cable:


This 24-hour minute-trend shows the frequency wander as well as the room temperature. This is not proof of a temperature dependence, but if it is then we get ~3 kHz/deg for the sensitivity. If this is actually the cable and not the amplifier, then we'll have to hunt for a lower tempco cable and put it in a box to isolate it.

  4255   Sun Feb 6 02:29:28 2011 ranaUpdateElectronicsAnalog MFD: longer cable

I swapped over to a 3x longer cable (old 65 ft. Pasternak cable from ancient 40m days). The old one was 6m, the new one is 18.2 m. It was already coiled up so I put it into a tupperware box to shield it somewhat from the HVAC wind.

The noise went down nearly proportional to the length (after I recalibrated the DAQ channel for the ~3x higher phase->voltage gain). With this length, the peak-peak mixer range is 5.5 MHz, so still enough to go an FSR here.


I give credit to the low frequency improvement entirely to Tupperware for their excellent containers. The current noise limit is most likely the SR560.

  9810   Tue Apr 15 02:19:54 2014 JenneUpdateLSCAnalog phasing of REFL11 and REFL55

[Jenne, EricQ]

I told Koji that I wanted to play with the common mode servo this evening, and he pointed out that we only get the signals after the digital demod phase angle in the digital system (obviously).  So, if I want to use either REFL11 or REFL55 for my CARM signal, I want to do something in analog-land so that my digital demod phase is close to 0 or 90. 

While we had the PRFPMI locked (with CARM offset of 2 or 3 nm), we set the demod phases of REFL11 and REFL55 to minimize a CARM line in the Q-phase.  This gave us -34 degrees for REFL11, and -75 degrees for REFL55. 

We calculated that about 1 degree of phase shift is about 1/(2 * pi * freq), or about 1.4e-8 seconds of delay for 11MHz.  We took the speed of light in the cables to be about 2/3*c, so 1.4e-8 * 2e8 = 2.9 meters per degree for 11MHz.  Since REFL11 was 34 degrees from 0, we estimate that we need to add about 98 meters of cable to the REFL11 signal path.  The same calculation for 55 MHz, but with a 15 degree shift required, gives 8.8 meters of cable to be added to the REFL55 signal path. 

I connected up some long BNC cables, and inserted them between the heliax breakout board on the LSC rack, and the respective PD inputs of the REFL11 and REFL55 demod boards.  I used (45 meters + 45 meters + a little bit) for REFL11, and used about 9 meters for REFL55. 

When we relocked the PRFPMI, and redid the phasing, we were very close to zero for both REFL11 and REFL55!  REFL11's digital demod phase is now +1 degree, and REFL55's digital demod phase is -5 degrees.

We changed the input of the CM servo board from POY (which Den and Koji had been using in December - see elog 9500) to REFL11 I MON. 

Q locked the FPMI (separate reply elog), and then we tried engaging the CM analog servo.  We were not successful. 


These settings were mostly copied from elog 9500, so they are almost surely not correct. 

CM servo screen:  In1 gain = 31dB, switch on, offset = -2.7V, boost off, super boosts off, option=disable, 79:1.6k switch disabled, polarity minus, option disable, AO gain=8dB, limiter enable.

For the slow path, CM_SLOW -> MC LSC servo had a +1 in the input matrix. 

CM filters in the AUX_ERR screen:  FM1 (unwhite) on, all others off, gain = 2.6. 

MC servo filters:  FM7, FM10 on, all others off (no triggered filter modules).  Gain = 0 initially.

MC servo board AO path disabled initially, G=-32dB initially.


Once Q had the FPMI locked, I tried increasing just the CM analog gain (by enabling the AO path on the MC board, and increasing the gain).  Doing this, I lost lock at -3 dB. 

I then tried again, this time alternating increasing the analog gain, and increasing the MC LSC servo gain.  I got up to 3e-3 for the MC digital gain, and -7 dB for the analog gain before we lost lock again.


We have determined that we should probably try just locking one of the arms with POX or POY, as Den and Koji did, to get a feel for how the system works.



  9811   Tue Apr 15 02:26:45 2014 ericqUpdateLSCAnalog phasing of REFL11 and REFL55

For future reference:

As we were poking around with the common mode servo in an FPMI configuration, we locked CARM/DARM with ALS as in recent ELOGs.

MICH was locked on ASDC: ASDC -> MICH = 10.0 in the DCPD DoF Matrix (I couldn't easily get AS55Q working, ASDC worked quickly and good enough)

MICH gain +25, FM4 FM5 On, FM2 switched on once locked. Offset was manually adjusted to get closer to dark fringe.

Actuated on BS: MICH->BS = 0.5 in Output Matrix.

  9812   Tue Apr 15 08:55:57 2014 KojiUpdateLSCAnalog phasing of REFL11 and REFL55

I have never used such a long cable for RF phase adjustment. The speed of the signal is 2e8 m/s and the frequency is ~10e6 Hz.
This means that the wavelength is only about 20m. How could you end up with ~100meters?
The convenient way to remember the cable delay is "1m, 1MHz, 2deg". This gives us ~1.5m for 11MHz and 34deg.

In fact, 1 degree of phase shift is not 1/(2 pi freq) second of delay, but f/360.

For such a precise phase adjustment, it is better to calibrate the delay with the network analyzer.


We calculated that about 1 degree of phase shift is about 1/(2 * pi * freq), or about 1.4e-8 seconds of delay for 11MHz.  We took the speed of light in the cables to be about 2/3*c, so 1.4e-8 * 2e8 = 2.9 meters per degree for 11MHz.  Since REFL11 was 34 degrees from 0, we estimate that we need to add about 98 meters of cable to the REFL11 signal path.  The same calculation for 55 MHz, but with a 15 degree shift required, gives 8.8 meters of cable to be added to the REFL55 signal path.   


  14348   Wed Dec 12 18:27:07 2018 JonOmnistructureUpgradeAnalog signals, A/D Acromag added to vacuum system

There turned out to be a few analog signals for the vacuum system after all. The TP2/3 foreline pressure gauges were never part of the digital system, but we wanted to add them, as some of the interlock conditions should be predicated on their readings. Each gauge connects to an old Granville-Phillips 375 controller which only has an analog output. Interfacing these signals with the new system required installing an Acromag XT1221 8-channel A/D unit. Taking advantage of the extra channels, I also moved the N2 delivery line pressure transducer to the XT1221, eliminating the need for its separate Omega DPiS32 controller. When the new high-pressure transducers are added to the two N2 tanks, their signals can also be connected.

The XT1221 is mounted on the DIN rail inside the chassis and I have wired a DB-9 feedthrough for each of its three input signals. It is assigned the IP on the vacuum subnet. Testing the channels in situ revealed a subtley in calibrating them to physical units. It was first encountered by Johannes in a series of older posts, but I repeat it here in one place.

An analog-input EPICS channel can be calibrated from raw ADC counts to physical units (e.g., sensor voltage) in two ways:

  1. Via LINR="LINEAR" by setting the engineering-units fields EGUF="[V_max_adc]", EGUL="[V_min_adc]"
  2. Via LINR="NO CONVERSION" by manually setting the gain ASLO="[V/count]" and offset AOFF="[V_offset]"

From the documentation, under the engineering-units method EPICS internally computes:

where EGUF="eng units full scale", EGUL="eng units low", and "full scale A/D counts" is the full range of ADC counts. EPICS automatically infers the range of ADC counts based on the data type returned by the ADC. For a 16-bit ADC like the XT1221, this number is 2^16 = 65,536.

The problem is that, for unknown reasons, the XT1221 rescales its values post-digitization to lie within the range +/-30,000 counts. This corresponds to an actual "full scale A/D counts" = 60,001. If a multiplicative correction factor of 65,536/60,000 is absorbed into the values of EGUF and EGUL, then the first term in the above summation can be corrected. However, the second term (the offset) has no dependence on "full scale A/D counts" and should NOT absorb a correction factor. Thus adjusting the EGUF and EGUL values from, e.g., 10V to 10.92V is only correct when EGUL=0V. Otherwise there is a bias introduced from the offset term also being rescaled.

The generally correct way to handle this correction is to use the manual "NO CONVERSION" method. It constructs calibrated values by simply applying a specified gain and offset to the raw ADC counts:

calibrated val = (measured A/D counts)  x ASLO + AOFF

The gain ASLO="[(V_max_adc - V_min_adc) / 60,001]" and the offset AOFF="0". I have tested this on the three vacuum channels and confirmed it works. Note that if the XT1221 input voltage range is restricted from its widest +/-10V setting, the number of counts is not necessarily 60,001. Page 42 of the manual gives the correct counts for each voltage setting.

  8375   Fri Mar 29 19:23:49 2013 Gabriele, JenneFrogsLSCAnalog whitening filter of REFL55 not switching

We discovered that the analog whitening filter of the REFL55_I board is not switching when we operate the button on the user interface. We checked with the Stanford analyzer that the transfer function always correspond to the whitening on.

The digital one is actually switching. We decided to keep the digital de-whitening on to compensate for the analog one. Otherwise we get a very bad shape of the PDH signal. Sorry Rana...

  8377   Fri Mar 29 19:58:24 2013 Gabriele, JenneFrogsLSCAnalog whitening filter of REFL55 not switching


We discovered that the analog whitening filter of the REFL55_I board is not switching when we operate the button on the user interface. We checked with the Stanford analyzer that the transfer function always correspond to the whitening on.

The digital one is actually switching. We decided to keep the digital de-whitening on to compensate for the analog one. Otherwise we get a very bad shape of the PDH signal. Sorry Rana...

 I forgot to say that the analog gain of the REFL55 channels has been reduced to 9db

  10104   Wed Jun 25 19:29:19 2014 AkhilUpdateElectronicsAnalog-to-Digital Converter

 I have been trying to use an ADC with the Raspberry Pi to be able to measure the phase difference between FC input and output signals.I had a hard time interfacing the ADC  with the Pi (setup attached) even after trying to debug the issue for last two days. So I and Eriq Q performed a system reboot on the Pi and tried all the possible ways for the Pi to detect the ADC but we were not able to. At the end we decided to order another IC(Microchip MCP 3008) which we hope can be interfaced with the Pi. Till then I will finish to write data from the FC into pipes so that the control computers can access the real time data. I will also look the correctness of the sampling time that is provided by the spec of the MCL-Mini circuits that is if we could really achieve 0.1 s sampling time with the FC.

  2559   Tue Feb 2 13:14:09 2010 KojiHowToIOOAnatomy of New Focus Resonant EOM

Joe let me use the resonant EOM for GigE phase camera for a while.
Then, I immediately started to open it :)

it uses the MiniCIrcuits T5-1T transformer and a TOKO RCL variable inductor.

The photos are on the Picasa 40m album.


  161   Mon Dec 3 19:44:58 2007 Accelerometers on new mountsConfigurationPEMAndrey

I (Andrey) continued today working with new accelerometer mounting. (see entry #151 about my Friday work).

I bought screws/washers and attached those mounts with accelerometers to metallic frames which are firmly cemented to the floor.

One such mount with three accelerometers (in X-, Y-, Z-directions) is installed near the ITMX (in the previous location, but NOT on top of the unused stack as before Friday), the other mount with three accelerometers in three orthogonal directions is installed near ETMX in the east end of the room (this set of accelerometers was installed between MC and BS before Friday). I uncoiled the cables, put them into the cable tray towards the ETMX, and hooked-up the three accelerometers near ETMX in the east end of the room.

Now all six accelerometers are hooked-up (that is, connected to power supply board with cables).

We decided with Steve Vass to put red cones (similar to those that are on highways in the road construction zones) in order to prevent people from bumping into accelerometers. Please use caution when walking along the X-arm.

I took several pictures of the new accelerometer setup. Picture "DSC_0194.JPG" shows the mount with accelerometers near the the ITMX and the beamsplitter chamber,
picture "DSC_0195.JPG" is the "zoomed-in" view of the same accelerometers, while picture "DSC_0196.JPG" shows the mount with accelerometers near ETMX in the east end of the room.

Many thanks to Mr. Steve Vass for his thorough explanation/showing me how to drill the metal and put threads in the holes.
  53   Thu Nov 1 19:55:03 2007 Andrey RodionovBureaucracyPhotosAndrey's photo
  515   Tue Jun 3 12:33:36 2008 AndreyUpdateCamerasAndrey, Josephb

Continuing our work with cameras,

1) we removed both cameras from their places on Monday afternoon, and were taking the beam-scans with a special equipment (see elog-entry 511) from Bridge bld.,

2) and on Tuesday morning we putted back the GC-750 camera into the transmitted beam path, camera GC-650 into the reflected beam path. We plan to compare the images from the "reflection camera" for several different angles of tilt of the camera.
  54   Thu Nov 1 19:55:59 2007 Andrey RodionovBureaucracyPhotosAndrey, Tobin, Robert - photo
  8887   Mon Jul 22 03:10:41 2013 ranaSummaryloreAngel of the Y End Table?

 Trying to take an image or movie of the ETMY Transmon cam, we got instead this attached image.

I think it is just some scattered green light, but others in the control room think that it is a message from somewhere or someone...

  8888   Mon Jul 22 06:58:17 2013 LisaSummaryloreAngel of the Y End Table?


 Trying to take an image or movie of the ETMY Transmon cam, we got instead this attached image.

I think it is just some scattered green light, but others in the control room think that it is a message from somewhere or someone...

 It is not an angel, it is clearly a four leaf clover (also known as "quadrifoglio"). It is very rare, it brings good luck!

  11206   Tue Apr 7 04:21:45 2015 ericqUpdateASCAngular Control during Locking

[J, Q]

Alignment is making it tough for locks to last more than 10 minutes. Many (but not all) locklosses correlate with some optic drifting away, and taking all of the light with it. The other locklosses are the quick ones that seem to pop up out of nowhere; we haven't made any headway on these. We wanted to get to a state where we could just let the interferometer sit for some minutes, to explore the data, but got caught up with alignment and PRMI things.

We're finding that both ITMs experience some DC force when entering full PRFPMI lock. I will calculate the torque expected from radiation pressure + offset beam spot, especially for ITMX, where we choose the spot position to be uncontrolled by ASS. 

I set up the QPD ASC servos to act in a common/differential way on the ETMs. The C1:ASC-XARM_[PIT/YAW] filter modules act on the common alignment, whereas the C1:ASC-YARM_[PIT/YAW] filter modules act on the differential alignment. This can soon be cleaned up with some model renaming to reduce confusion. 

Using DC oplev values as a guide, we are hand tuning ITM alignment once the AO path is engaged and we see the DC drift occurring. Then, we set the QPD servo offsets and engage them. 

In this manner, we were able to lock the interferometer at:

  • Arm transmission 150 x single arm power
  • POPDC indicated a recycling gain of ~5.5
  • ASDC/POPDC indicated a contrast of 99.8%
  • REFLDC indicated a visibility of 80%

We made the PRMI transition to 1f numerous times, but found that the sideband power fluctuations would get significantly worse after the transition. 

We found that the gains that were previously used were too small by a factor of a few. There is a DC change visible in REFL165 before and after the transition (Also POP55, aka REFL55, is not DQ'd angry). Really, it isn't certain that we've zero'd the offset in the CARM board either, so REFL55's zero crossing isn't necessarily more trustworthy that REFL165's. We can go back in the data and do some 2D histograming to see where in the error signal space the sideband power is maximized. 

Jenne reports:

  • The all RF transition succeeded 13/29 times. 
  • PRMI 1f transision succeeded 10/10 times. 
  16125   Thu May 6 16:13:39 2021 AnchalSummaryIMCAngular actuation calibration for IMC mirrors

Here's my first attempt at doing angular actuation calibration for IMC mirrors using the method descibed in /users/OLD/kakeru/oplev_calibration/oplev.pdf by Kakeru Takahashi. The key is to see how much is the cavity mode misaligned from the input mode of beam as the mirrors are moved along PIT or YAW.

There two possible kinds of mismatch:

  • Parallel displacement of cavity mode axis:
    • In this kind of mismatch, the cavity mode is simply away from input mode by some distance \large \beta.
    • This results in transmitted power reduction by the gaussian factor of \large e^{-\frac{\beta^2}{w_0^2}} where \large w_0 is the beam waist of input mode (or nominal waist of cavity).
    • For some mismatch, we can approximate this to
                                                                               \large 1 - \frac{\beta^2}{w_0^2}
  • Angular mismatch of cavity mode axis:
    • The cavity mode axis could be tilted with respect to input mode by some angle \large \alpha.
    • This results in transmitted power reduction by the gaussian factor of \large e^{- \frac{\alpha^2}{\alpha_0^2}}  where \large \alpha_0 is the beam divergence angle of input mode (or nominal waist of cavity) given by \large \frac{\lambda}{\pi w_0}.
    • or some mismatch, we can approximate this to
                                                                                \large 1 - \frac{\alpha^2}{\alpha_0^2}

Kakeru's document goes through cases for linear cavities. For IMC, the mode mismatches are bit different. Here's my take on them:


  • MC2 is the easiest case in IMC as it is similar to the end mirror for linear cavity with plane input mirror (the case of which is already studies in sec 0.3.2 in Kaker's document).
  • PIT:
    • When MC2 PIT is changed, the cavity mode simple shifts upwards (or downwards) to the point where the normal from MC2 is horizontal.
    • Since, MC1 and MC3 are plane mirrors, they support this mode just with a different beam spot position, shifted up by \large (R-L)\theta.
    • So the mismatch is simple of the first kind. In my calculations however, I counted the two beams on MC1 and MC3 separately, so the factor is twice as much.
    • Calling the coefficient to square of angular change \large \eta, we get:
                                     \large \eta_{._{2P}} = \frac{2 (R-L)^2}{w_0^2}
    • Here, R is radius of curvature of MC1/3 taken as 21.21m and L is the cavity half-length of IMC taken as 13.545417m.
  • YAW:
    • For YAW, the case is bit more complicated. Similar to PIT, there will be a horizontal shift of the cavity mode by \large (R-L)\theta.
    • But since the MC1 and MC3 mirrors will be fixed, the angle of the two beams from MC1 and MC3 to MC2 will have to shift by \large \theta/2.
    • So the overall coefficient would be:
                                     \large \eta_{._{2Y}} = \frac{2 (R-L)^2}{w_0^2} + \frac{2}{4\alpha_0^2}
    • The factor of 4 in denominator of seconf term on RHS above comes because only half og angular actuation is felt per arm. The factor of 2 in numerator for for the 2 arms.


  • First, let's establish that the case of MC1 and MC3 is same as the cavity mode must change identically when the two mirrors are moved similarly.
  • YAW:
    • By tilting MC1 by \large \theta, we increase the YAW angle between MC1 and MC3 by \large \theta.
    • Beam spot on both MC1 and MC3 moves by \large (R-L)\theta.
    • The beam angles on both arms get shifted by \large \theta/2.
    • So the overall coefficient would be:
                                     \large \eta_{._{13Y}} = \frac{2 (R-L)^2}{w_0^2} + \frac{2}{4\alpha_0^2}
    • Note, this coefficient is same as MC2, so it si equivalent to moving teh MC2 by same angle in YAW.
  • PIT:
    • I'm not very sure of my caluculation here (hence presented last).
    • Changing PIT on MC1, should change the beam spot on MC2 but not on MC3. Only the angle of MC3-MC2 arm should deflect by \large \theta/2.
    • While on MC1, the beam spot must change by \large (R-L)\theta/2 and the MC1-MC2 arm should deflect by \large \theta/2.
    • So the overall coefficient would be:
                                     \large \eta_{._{13P}} = \frac{(R-L)^2}{4 w_0^2} + \frac{2}{4\alpha_0^2}

Test procedure:

  • We first clicked on MC WFS Relief (on C1:IOO-WFS_MASTER) to reduce the large offsets accumulated on WFS outputs. This script took 10 minutes and reduced the offsets to single digits and IMC remained locked throughout the process.
  • Then we switched off the WFS to freeze the outputs.
  • We moved the MC#_PIT/YAW_OFFSET up and down and measured the C1:IOO-MC_TRANS_SUMFILT_OUT channel as an indicater of IMC mode matching.
  • Attachement 1 are the 6 measurements and there fits to a parabola. Fitting code and plots are thanks to Paco.
  • We got the curvature of parabolas \large \gammafrom these fits in units of 1/cts^2.
  • The \large \eta coefficients calculated above are in units of 1/rad^2.
  • We got the angular actuation calibration from these offsets to physical angular dispalcement in units of rad/cts by \large \sqrt{\gamma / \eta}.
  • AC calibration:
    • I parked the offset to some value to get to the side of parabola. I was trying to reduce transmission from about 14000 cts to 10000-12000 cts in each case.
    • Sent excitation using MC#_ASCPIT/YAW_EXC using awg at 77 Hz and 10000 cts.
    • Measured the cts on transmission channel at 77 Hz. Divided it by 2 and by the dc offset provided. And divided by the amplitude of cts set in excitation. This gives \large \eta_{ac} analogous to above DC case.
    • Then angular actuation calibration at 77 Hz from these offsets to physical angular dispalcement in units of rad/cts by \large \sqrt{\gamma/\eta_{ac}}.
  • Following are the results:
    Optic Act
    Calibration factor at DC [µrad/cts]
    Calibration factor at 77 Hz [prad/cts]
    MC1 PIT 7.931+/-0.029 906.99
    MC1 YAW 5.22+/-0.04 382.42
    MC2 PIT 13.53+/-0.08 869.01
    MC2 YAW 14.41+/-0.21 206.67
    MC3 PIT 10.088+/-0.026 331.83
    MC3 YAW 9.75+/-0.05 838.44


  • Note these values are measured with the new settings in effect from 16120. If these are changed, this measurement will not be valid anymore.
  • I believe the small values for MC1 actuation have to do with the fact that coil output gains for MC1 are very weird and small, which limit the actuation strength.
  • TAbove the resonance frequencies, they will fall off by 1/f^2 from the DC value. I've confirmed that the above numbers are of correct order of magnitude atleast.
  • Please let me know if you can point out any mistakes in the calculations above.
  8489   Thu Apr 25 03:35:28 2013 JenneUpdateLockingAngular motion does not explain RIN

Den made a nice elog about the PRMI RIN that we see a few weeks ago:  8464.  The RIN that we're seeing is typically about ~30%.  The question at hand is: what is causing this power fluctuation, and more specifically, is it the angular motion of the mirrors?

I find that no, the angular motion that we see does not explain the RIN that we see.

In the attached Mathematica notebook, I calculate the power lost due to angular misalignments of one or more mirrors.  (Math comes from Appendix A of Keita's thesis.)

From calibrated oplev spectra, our mirrors are moving about 1 microradian (RMS, which is dominated by low frequencies).  From a super sophisticated "draw on the TV, then measure" method (details below), I have estimated that the maximum static misalignment that we're seeing is about 2 microradians.

With all of this, I find that for a g-parameter of 0.94, the power lost due to misalignments should, at maximum, be 0.6%.  I need a g-parameter of 0.995 to get a power loss of 23%.  Alternatively, if I take the derivative of the power coupling function, to find the static misalignment at the steepest slope of the curve (and thus, the place where any AC misalignment would have the most effect), for 1urad of AC misalignment, I get 40% power loss. 

So, in order for the AC angular motion that we see to explain the RIN that we see, either our mirrors are very, very misaligned (so much so that we couldn't really be locking), or our cavity is much closer to unstable than expected from Jamie's calculations.  Since both of these cases (static misalignment or incorrect g-parameter calculation) have to be taken to extremes before they approximate the RIN that we see, I do not think that this power loss is due to angular fluctuations.

This means that we have to think of another potential cause for this RIN that we're seeing.

Details on the "draw on TV and measure" technique for determining static cavity misalignments:  Looking at the POP camera view, with the PRM significantly misaligned, I traced the straight-through beam spot.  I then restored the PRM, and during several momentary locks, I traced the beam spot, which I took to be the saturated area of the camera.  The idea here is that the straight-through beam represents the incident beam axis, while the locked beam represents the cavity axis.  I'm assuming that the camera image plane is at the face of PR2. I approximately found the center of each of my tracings, and found them to be ~1/4 inch apart.  I also measured the "spot size" of the sideband-locked PRMI, and found it to be ~3.5 inches.  So, very roughly, the ratio of (distance between spots)/(size of beam) is ~0.07. This corresponds to a static misalignment of either the ITM or the PRM of ~2urad, rounding up. (I use the Jamie's calculated g-parameters from elog 8316, the case of flipped PR2, tangential = 0.94 to calculate the effective RoC of the PRM). 

  890   Wed Aug 27 10:55:35 2008 YoichiHowToComputersAnnoying behavior of the touch pads of the lab. laptops is fixed
I was sick of the stupid touch pad behavior of the lab. laptops, i.e. firefox goes back and forth in the history when the cursor is moved.
It was caused by firefox mis-interpreting the horizontal scroll signal as back/forward command.
I stopped it by going to about:config in firefox and set mousewheel.horizscroll.withnokey.action to 0 and
mousewheel.horizscroll.withnokey.sysnumlines to true.
  13919   Wed Jun 6 10:44:52 2018 gautamUpdateVACAnnulus pressure channels added to frames

[steve, gautam]

We added the following channels to C0EDCU.ini and restarted the daqd processes. Channels seem to have been added successfully, we will check trend writing later today. Motivation is to have a long term record of annulus pressure (even though we are not currently pumping on the annulus).






plot next day

  11477   Mon Aug 3 18:19:09 2015 JessicaUpdateGeneralAnodization of front panels accounted for

Previously, I had gotten the same results for the conductive and the isolated front panels. Today, I sanded off the anodized part on the back of the conductive front panel. I checked afterwards with a mulitmeter to ensure that it was indeed conductive through all the SMA connectors. 

I drove a frequency of 29.359 Hz through the X Arm cable and 29.3592 Hz through the Y Arm cable, giving a difference of 200 Hz. Previously, there would only be a spike in the Y Arm at the difference, while the X Arm did not change if the Y arm was on or off. Now that the panel is fully conductive, a spike can also be seen in the X arm, indicating that crosstalk may possibly be happening with this panel, now that the spike corresponds to both the X arm and Y arm. These results are only after one set of data. Tomorrow I'll take two more sets of data with this panel and do a more in depth comparison of these results to what had been previously seen.  

  14768   Wed Jul 17 20:12:26 2019 KruthiUpdateCamerasAnother GigE in place of analog camera

I've taken the MC2 analog camera down and put another GigE (unit 151) in its place. This is just temporary and I'll put the analog camera back once I finish the MC2 loss map calibration. I'm using a 25mm focal length camera lens with it and it gives a view of MC2 similar to the analog camera one. But I don't think it is completely focused yet (pictures attached).

...more to follow

gautam - Attachment #3 is my (sad) attempt at finding some point scatterers - Kruthi is going to play around with photUtils to figure out the average size of some point scatterers.

  14727   Fri Jul 5 20:57:04 2019 KojiUpdateSUSAnother M7.1 EQ

[Kruthi, Koji]

Koji came to the lab to align the IMC/IFO, but found the mirrors are dancing around. Kruthi told me that there was M7.1 EQ at Ridgecrest. Looks like there are aftershocks of this EQ going on. So we need to wait for an hour to start the alignment work.

ITMX and ETMX are stuck.

  14728   Fri Jul 5 21:53:10 2019 KojiUpdateSUSAnother M7.1 EQ

- ITM unstuck now
- IMC briefly locked at TEM00

A series of aftershocks came. I could unstick ITMX by turning on the damping during one of the aftershocks.
Between the aftershocks, MC1~3 were aligned to the previous dof values. This allowed the IMC flashing. Once I got the lock of a low order TEM mode, it was easy to recover the alignment to have a weak TEM00.
Now at least temporarily the full alignment of the IMC was recovered.

  14729   Fri Jul 5 22:21:13 2019 KojiUpdateSUSAnother M7.1 EQ

In fact, ETMX was not stuck until the M7.1 EQ today. After that it got stuck, but during the after shocks, all the OSEMs occasionally showed full swing of the light levels. So I believe the magnets are OK.

  584   Fri Jun 27 18:03:46 2008 JenneUpdateElectronicsAnother bad cable in the MC servo
Eric was helping me to measure the response of the LO input on the MC's Demod board, and when we disconnected the end of the cable between the demod board and the delay line phase shifter for the 29.5MHz oscillator, we noticed that the phase shifter's end of the cable was loose, like the connector wasn't fully connected. When we checked it by wiggling the connector, the SMA end fell off. I made a new SMA end for the cable, and reinstalled the cable. The MC locked as soon as I plugged the cable in, so everything seems good again. I tried to not change the cable length when I remade the connector, but the cable is shorter than it was by a small amount, due to the way the end fell off.
  7114   Wed Aug 8 10:15:13 2012 jamieUpdateEnvironmentAnother earthquake, optics damped

There were another couple of earthquakes at about 9:30am and 9:50am local.


All but MC2 were off the watchdogs.  I damped and realigned everything and everything looks ok now.


  14135   Sun Aug 5 15:43:50 2018 gautamUpdateSUSAnother low noise bias path idea

OK, how about this:

  • Attachment #1 shows the proposed schematic.
    • It consists of a second order section with Gain x10 to map the +/-10V DC range of the DAC to +/- 100V DC such that we preserve roughly the same amount of DC actuation range.
    • Corner frequency of the SOS is set to ~0.7 Hz. In hindsight, maybe this is more aggressive than necessary, we can tune this.
    • DC gain is 20 dB (typo in the text where I say the DC gain is x15, though we could go with this option as well I think if we want a larger series resistance).
    • A first order passive low-pass stage is added to filter out the voltage noise of the PA91, which dominates the output voltage noise (next bullet).
  • Attachment #2 shows the transfer function from input to output
    • The two traces compare having just a single SOS filtering stage vs the current topology of having two SOS stages.
    • The passive output RC network is necessary in either case to filter the voltage noise of the PA91 OpAmp.
    • For the DAC noise, I just assumed a flat noise level of 5 \mu V / \sqrt{\mathrm{Hz}}, I don't actually know what this is for the Acromag DACs.
  • Attachments #3 shows a breakdown of the top 5 noise contributions.
    • The PA91 datasheet doesn't give current noise information so I just assumed 1 fA / \sqrt{\mathrm{Hz}}, which was what was used for the PA85 in the existing opamp.lib file.
    • The voltage noise is modelled as 4.5 \sqrt{1+\frac{80}{f}} nV / \sqrt{\mathrm{Hz}}, which seems to line up okay with the plot on Pg4 of the datasheet.
    • So the model suggests we will be dominated by the voltage noise of the PA91.
  • Attachment #4 translates the noise into current noise seen by the actuator.
    • I add the Johnson noise contribution of the series resistance for this path, which is assumed to be 10 k \Omega.
    • For comparison, I add the filtered DAC noise contribution, and Johnson noise of the proposed series resistance in the fast path.
    • For the bias path, we are dominated by the Johnson noise of the series resistor from ~60 Hz upwards.
    • It's not quite fair to say that the Johnson noise of the resistance in the fast path dominates, the quadrature sum of fast and bais paths will be ~1.2 times of the former alone. 
    • Bottom line: we will be in the regime of total current noise of ~2.2 pA/rtHz, where I think Kevin's modeling suggests we can see some squeezing.

The question still remains of how to combine the fast and bias paths in this proposed scheme. I think the following approach works for prototyping at least:

  • Remove the series resistance on the existing coil driver boards' bias path, hence isolating this from the coil.
  • Route the DB15 output connector from the coil driver board (which is now just the fast actuation signals) into a sub-sattelite box housing the bias path electronics.
  • Sum the two signals as it is done now, by simply having a conductor (PCB trace) merge the two paths after their respective series resistances.

In the longer term, perhaps the Satellite Box revamp can accommodate a bias voltage summation connector.


Bah! Too complex.

I have neglected many practical concerns. Some things that come to mind:

  1. Is it necessary to protect the upstream DAC from some potential failure of the PA91 in which the high voltage appears at the input?
  2. What is the correct OpAmp for this purpose? This chart on Apex's page suggests that PA15, PA85, PA91 and PA98 are all comparable in terms of drive capability, and the spec sheets don't suggest any dramatic differences. Some LIGO circuits use PA85, some use PA90, but I can't find any that use PA91. Perhaps Rana/Koji can comment about this.
  3. What kind of protection is necessary for the PA91 power?
  4. What is the correct way to do heat management? Presumably we need heatsinks, and in fact, there is a variant of the packaging style that has "formed" legs, which from what I can figure out, allow the heat sink plane on the PA91 to be parallel to the PCB surface. But I think the heat-sink wisdom suggests vertical fins are the most efficient (not sure if this holds if the PCB is inside a box though). What about the PCB itself? Are some kind of special traces needed?
  5. Can we use the current-limiting resistor feature on the PA91? The datasheet seems to advice against it for G>10 configurations, which is what we need, although our requirement is only at DC so I don't know if that table is applicable to this circuit.
  6. Are 3W resistors sufficient? I think we require only 10mA maximum current to preserve the current actuation range, so 100 V * 10mA = 1W, so 3W leaves some safety margin.
  7. All capacitors should be rated for 500 V per the datasheet.  
  14147   Wed Aug 8 23:06:59 2018 gautamUpdateSUSAnother low noise bias path idea

Today while Rich Abbott was here, Koji and I had a brief discussion with him about the HV amplifier idea for the coil driver bias path. He gave us some useful tips, perhaps most useful being a topology that he used and tested for an aLIGO ITM ESD driver which we can adapt to our application. It uses a PA95 high voltage amplifier which differs from the PA91 mainly in the output voltage range (up to 900V for the former, "only" 400V for the former. He agrees with the overall design idea of 

  • Having a LN opamp with the HV amp inside the feedback loop for better voltage noise at low frequencies.
  • Having a passive RC network at the output of the HV amp to filter out noise at high frequencies.

He also gave some useful suggestions like 

  • Using the front panel of the box that as a heatsink for the HV amps.
  • Testing the stability of the nested opamp loop by "pinging" the output of the opamp with some pulses from a function generator and monitoring the response to this perturbation on a scope.

I am going to work on making a prototype version of this box for 5 channels that we can test with ETMX. I have been told that the coupling from side coil to longitudinal motion is of the order of 1/30, in which case maybe we only need 4 channels.

  14165   Wed Aug 15 19:18:07 2018 gautamUpdateSUSAnother low noise bias path idea

I took another pass at this. Here is what I have now:

Attachment #1: Composite amplifier design to suppress voltage noise of PA91 at low frequencies.

Attachment #2: Transfer function from input to output.

Attachment #3: Top 5 voltage noise contributions for this topology.

Attachment #4: Current noises for this topology, comparison to current noise from fast path and slow DAC noise.

Attachment #5: LISO file for this topology.

Looks like this will do the job. I'm going to run this by Rich and get his input on whether this will work (this design has a few differences from Rich's design), and also on how to best protect from HV incidents.

  14169   Thu Aug 16 23:06:50 2018 gautamUpdateSUSAnother low noise bias path idea

I had a very fruitful discussion with Rich about this circuit today. He agreed with the overall architecture, but made the following suggestions (Attachment #1 shows the circuit with these suggestions incorporated):

  1. Use an Op27 instead of LT1128, as it is a more friendly part especially in these composite amplifier topologies. I confirmed that this doesn't affect the output voltage noise at 100 Hz, we will still limited by Johnson noise of the 15kohm series resistor.
  2. Take care of voltage distribution in the HV feedback path
    • I overlooked the fact that the passive filtering stage means that the DC current we can drive in the configuration I posted earlier is 150V / 25kohm = 6mA, whereas we'd like to be able to drive at least 10 mA, and probably want the ability to do 12 mA to leave some headroom.
    • At the same time, the feedback resistance shouldn't be too small such that the PA91 has to drive a significant current in the feedback path (we'd like to save that for the coil).
    • Changing the supply voltage of the PA91 from 150 V to 320 V, and changing the gain to x30 instead of x15 (by changing the feedback resistor from 14kohm to 29kohm), we can still drive 12 mA through the 25 kohms of series resistance. This will require getting new HV power supplies, as the KEPCO ones we have cannot handle these numbers.
    • The current limiting resistor is chosen to be 25ohms such that the PA91 is limited to ~26 mA. Of this, 300V / 30kohm ~ 10 mA will flow in the feedback path, which means under normal operation, 12 mA can safely flow through the coils.
    • Rich recommended using metal film resistors in the high voltage feedback path. However, these have a power rating, and also a voltage rating. By using 6x 5kohm resistors, the max power dissipated in each resistor is 50^2 / 5000 ~ 0.5 W, so we can get 0.6 W (or 1W?)  rated resistors which should do the job. I think the S102K or S104K series will do the job.
  3. Add a voltage monitoring capability.
    • This is implemented via a resistive voltage divider at the output of the PA91.
    • We can use an amplifier stage with whitening if necessary, but I think simply reading off the voltage across the terminating resistor in the ladder will be sufficient since this circuit will only have DC authority.
  4. Make a Spice model instead of LISO, to simulate transient effects.
    • I've made the model, investigating transients now.
  5. High voltage precautions:
    • When doing PCB layout, ensure the HV points have more than the default clearance. Rich recommends 100 mils.
    • Use a dual-diode (Schottky) as input protection for the Op27 (not yet implemented in Spice model).
    • Use a TVS diode for the moniotring circuit (not yet implemented in Spice model).
    • Make sure resistors and capacitors that see high voltage are rated with some safety margin.
  6. Consider using the PA95 (which Rich has tested and approves of) instead of the PA91. Does anyone have any opinions on this?

If all this sounds okay, I'd like to start making the PCB layout (with 5 such channels) so we can get a couple of trial boards and try this out in a couple of weeks. Per the current threat matrix and noises calculated, coil driver noise is still projected to be the main technical noise contribution in the 40m PonderSqueeze NB (more on this in a separate elog).


Looks like this will do the job. I'm going to run this by Rich and get his input on whether this will work (this design has a few differences from Rich's design), and also on how to best protect from HV incidents.

  6454   Tue Mar 27 17:38:03 2012 JenneUpdateIOOAnother possibility / thought

I'm meditating over the mode matching from the mode cleaner to the ITMs, and I had another thought:

Have we changed the pointing of the MC significantly enough that we are no longer on the center of the MMT mirrors?  To be this significant, we would probably also have had to scoot the Faraday a bit too, since it's skinny like a straw.  It looks like our measurements of the input beam have been the following:

MC waist, 21 May 2010

After MMT2, 18 June 2010 (a few days before this, we flipped the MMT2 mount to 'perfect' the mode matching up to 99.3%, so I don't think the MMT has moved since then.)

After MMT2, 26 March 2012

There's a big o' ~2 year gap between our measurements, and we've been in and out of the vacuum a few times since then.  I'll flip through the elog, but does anyone have any memory of us moving the Faraday after June 2010?  When was the last time we made sure that we were at least close to the center of the MMT mirrors? 

  6455   Tue Mar 27 17:52:08 2012 KojiUpdateIOOAnother possibility / thought

It is quite likely that we touched the Faraday in Nov 2010.

In this entry http://nodus.ligo.caltech.edu:8080/40m/3874 I wrote that I removed the MCT optics in the chamber.
This is the pickoff between the IMC and the Faraday. This causes the beam shift. Therefore, the Faraday had
to be moved.

There were intensive in-chamber activities from Nov to Dec 2010. I am sure that almost everytime we went into
the chamber, we checked the spot position on the MMT mirrors as well as the TT and PZT mirrors.

Does the miscentering of the spots on the MMT mirrors cause the mode matching significantly changed?

ELOG V3.1.3-