ELOG 40m

Broken PMC Servo Board

I broke the PMC servo board (on accident).

I was trying to measure the resistance of the extra resistor that someone put between the board and the HV OUT connector, since this is part of an RC filter (where C is the capacitance of the PZT on the PMC) that I need to know the values of as part of my mission to make a 14.6kHz notch for the PMC body mode. The resistance is 63.6k. I had to pull the board to get in to measure this resistance.

This resistor between the board and the center pin of the panel-mount HV OUT connector made a rigid connection between the board and the panel. When I was putting the board back in, I must have strained this connection enough that it broke. We don't have any of the same kind of resistor here at the 40m, so I'm waiting until after lunch to go to Wilson house and see if they've got any. The IFO is down until I get this sorted out.

879

Mon Aug 25 14:18:36 2008

PMC servo board is fixed

The PMC servo board is back in place, all fixed up with a shiny new resistor. The PMC locks, and the MC locks (I'm not saying anything either way about how long the MC will stay locked, but it is locked for now). The resistor is connected to the connector using a short piece of wire, so this problem won't happen again, at least with this connector on this board.

884

Tue Aug 26 09:04:59 2008

PMC Servo Board: Out for Repairs

I've started modifying our PMC board to bring it up to the 21st century - leave the screen alone or else you might zap something.

892

Wed Aug 27 13:55:43 2008

Board is back in. PMC is locked.

Nominal gain is now 15 dB with brick. We need to do more studies:

Find out why there is still 35 MHz signal at the error point. Order some low pass filters to cut off above 35 MHz.
Explore brick + no-brick loop shapes and error spectra.
Measure and set the OLG.

We've left the copper-wrapped lead brick installed to let it slowly conform to the glass better.

893

Thu Aug 28 18:56:14 2008

There was a beam block after the Mach Zender. Who or what put this there?

The going to the MC now looks distorted as if someone has left something funny in the beam or maybe the new PMC has started to degrade??

Use the ELOG people...its good for you.

895

Fri Aug 29 02:40:43 2008

Quote:

Board is back in. PMC is locked.

This entry has details about the low pass filter after the PMC mixer. This filter has a few purposes:

1] Remove the beat signal (at 2*f_mod) between the PD RF signal at f_mod and the LO signal at f_mod.
2] Remove the beat signal (at f_mod) between the PD RF signal at 2*f_mod (which comes from the
beating of the upper and lower RF sidebands) and the LO signal at f_mod.
3] Remove other RF signals from non-ideal behavior of the LO drive signal and distortion in the RF PD pre-amp.

So its important to have a very good rejection at 35 MHz and higher. I used the Hartmut LC network design which is
installed on H1, H2, & L1. Since there is a high gain in the audio amps right after the mixer we have to get rid of
the RF or else we'll get slew rate limited or otherwise rectified downconversion of the RF signal into our audio band.

Of course, what everyone immediately realizes from the above 3 points, is that this filter can't protect the PMC
noise performance from homodyne mixing (e.g. 2*f_mod in the LO and 2*f_mod in the RF PD). To get around that, we're
ordering some filters from Mini-Circuits to remove the 2f from those signals by ~30 dB. As long as we install
the same filters on the RF and LO legs, there should be no significant phase shift in the demodulated signal.

The attached 2 page PDF shows the calculated before and after TFs of this filter. The 2 attached .m files
calculate the TF's and have ascii art which shows how the filter works.

Here's a comparison of the attenuation (in dB) of 2 candidate Mini-circuits filters:

f(MHz)	SLP-30	SLP-50
31	0.5	0.4
35	1.3	0.4
38	6.1	0.4
40	10.8	0.42
61	46.3	14.8
71	60	29
91	76.9	48
107	80	60

We don't have tabulated data at the same frequencies for both filters so I just made up some of the points by eye-balling the
plots from the catalog - but you get the idea: we can get away with using the SLP-30 at 35 MHz since it only attenuates the
signals by ~1.5 dB. So if someone can find 4 of these then Steve doesn't have to order any from Mini-Circuits.

Attachment 1: pmclp-07.pdf

Attachment 2: pmclp_40m_080824.m

% PMCLP is a TF of the IF filter after the PMC mixer
%       

% Mixer_Voltage -- Rs -- L1 --- L2 ---------Vout
%                            |      |   |
%                           C1     C2   Rl                   
%                            |      |   |
%                           GND    GND  GND
%

... 58 more lines ...

Attachment 3: pmclp.m

% PMCLP is a TF of the IF filter after the PMC mixer
%       

% Mixer_Voltage -- Rs -- L1 --- L2 ---------Vout
%                            |      |   |
%                           C1     C2   Rl                   
%                            |      |   |
%                           GND    GND  GND
%

... 57 more lines ...

896

Fri Aug 29 10:20:32 2008

Quote:

I put the block. I was frequently reaching to the FSS box to change the test point probes. I put the block to protect my hands/clothes from being burnt accidentally.

901

Fri Aug 29 15:01:45 2008

steve

MOPA_HTEMP in increasing

The laser chiller temp is 21.9C ( it should be 20.0C )
Control room temp 73F ok, no obvious block

Ops, there is a piece of paper blocking the intake of the chiller

This is a four day plot. The paper was blocking the air flow all day.

Attachment 1: htcl.jpg

902

Fri Aug 29 16:35:18 2008

Quote:

The apparent distortion of the MC refl. was caused by mis-alignment of the MC mirrors.
Because the MC1 was mis-aligned, the reflected light was clipped by a steering mirror.
I restored the MC angle bias values from the conlog history and now the MC locks.
According to conlog, the MC alignment was changed at around 18:30 on Thursday PDT.
It could have been caused by the computer reboots.

903

Fri Aug 29 17:39:25 2008

The attached PNG shows the PMC error and controls signals with no calibration.

There are 3 states:

DARK - RF input disabled & output blanked. This should be a measure of the ADC noise

(-10 dB) - This is with the gain slider down at 5 dB instead of the nominal 15 dB.

Looks like the Generic DAQ board whitening is good enough for these signal levels above ~1 Hz.

From the low and high gain spectra it also looks like the UGF is ~500 Hz with the gain at 15 dB.

Attachment 1: mcf.png

904

Fri Aug 29 18:24:48 2008

I calibrated the PMC PZT at DC by using 'trianglewave' to drive the DC offset slider
and reading back PMC_PZT and PMC_TRANSPD_F (both are DC coupled DAQ channels).

The attached PDF illustrates the method: look at the voltage required to span 1 FSR and then divide.

PMC_cal (m/V) = (1064 nm)/2 / V_FSR

The calibration for our PZT is therefore 10.4 nm/V.
The full scale (0-300 V) range is 3.1 microns.

From Jenne's elog entry we know that the series resistor to the PZT is 63.6 kOhms. The PZT is labeled as
having a capacitance of 279 nF. So the PMC drive's pole frequency is 1/2/pi/63.6e3/279e-9 = 9 Hz +/- 0.5 Hz.
The cable capacitance is ~20 pF/foot so its not significant for this.

The template file is Templates/PMC-PZTcal.xml.

Using the above calibrations, also plot the calibrated PMC ERR and PZT spectra.

Attachment 1: pmc-pzt-cal.pdf

Attachment 2: mcf.png

905

Fri Aug 29 22:57:48 2008

FSS loop transfer functions

I've been measuring a bunch of transfer functions of the FSS related stuffs.
There are a lot to be analyzed yet, but here I put one mystery I'm having now.
Maybe I'm missing something stupid, so your suggestions are welcome.

Here is a conceptual diagram of the FSS control board

TP3 TP4
^ ^
| |
RF PD -->--[Mixer]-----[Sum Amp]------>--[Common Gain]--->----[Fast Gain]----[Filter]--> NPRO PZT
^ | ^ | |
| V | V |
LO ---->------- TP1 IN TP2 -->---[Filter]--[High Volt. Amp.] --> Phase Corrector

What I did was first to measure a "normal" openloop transfer function of the FSS servo.
The FSS was operated in the normal gain settings, and a signal was injected from "IN" port.
The open loop gain was measured by TP1/TP2.
Now, I disconnected the BNC cable going to the phase corrector to disable the PC path and locked the ref. cav.
only using the PZT. This was done by reducing the "Common Gain" and "Fast Gain" by some 80dB.
Then I measured the open loop gain of this configuration. The UGF in this case was about 10kHz.
I also measured the gain difference between the "normal" and "PZT only" configurations by injecting
a signal from "IN" and measuring TP3/TP2 and TP4/TP3 with both configurations (The signal from the Mixer was
disconnected in this measurement).

The first attachment shows the normal open loop gain (purple) and the PZT only open loop gain scaled by the
gain difference (about 80dB). The scaled PZT open loop gain should represent the open loop gain of the PZT
path in the normal configuration. So I expected that, at low frequencies, the scaled PZT loop TF overlaps the normal
open loop TF.
However, it is actually much larger than the normal open loop gain.
When I scale the PZT only TF by -30dB, it looks like the attachment #2.
The PZT loop gain and the total open loop gain match nicely between 20kHz and 70kHz.
Closer look will show you that small structures (e.g. around 30kHz and 200kHz) of the two
TFs also overlap very well. I repeated measurements many times and those small structures are always there (the phase is
also consistently the same). So these are not random noise.

I don't know where this 30dB discrepancy comes from. Is it the PC path eating the PZT gain ?

I have measured many other TFs. I'm analyzing these.
Here is the TO DO list:

* Cavity response plot from AOM excitation measurements.
* Cavity optical gain plot.
* Reconstruct the open loop gain from the electric gain measurements and the optical gain above.
* Using a mixer and SR560(s), make a separate feedback circuit for the PZT lock. Then use the PC path
to measure the PC path response.
* See the response of the FSS board to large impulse/step inputs to find the cause of the PC path craziness.
etc ...

Attachment 1: OPLTFs.pdf

Attachment 2: OPLTFsScaled.pdf

906

Sat Aug 30 13:28:01 2008

This is a list of changes made to the PMC board while we had it out for modifying the notch:

LC-LC 4th order low pass filter
Replace the AD797 (U2) with an OP27. AD797's are bad - do not use them anywhere for any reason. The OP27 is slower and has a 3x worse input noise but doesn't compromise the bandwidth or noise performance of the PMC by any significant amount. The rule is: use OP27 everywhere unless you have a very good reason why not.
There is no 'H1' jumper on board. R9 is 90.9 Ohms and R2=900 Ohms so that the U2 stage has a gain of 10.
Cut a trace and inserted a 500 Ohm resistor between U2-pin6 and U5A-pin2 (the AD602). The AD602 has a 100 Ohm input impedance which cannot be driven without limiting by the AD797 or the OP27. The 500 Ohm resistor makes it a driveable load for low level signals which is all that should be there since its the error point of the servo. it also becomes a 6:1 voltage divider. Since the AD602 has a fixed output voltage noise of 100 nV/rHz, this will limit the noise performance if the VGA gain is less than 20 dB, but whatever.
R11 7.87k -> 1.74k, R12 = 78.7k -> 700k. This increases the high frequency gain of that stage by 7.87/1.74 = 4.5 and lowers the low frequency pole from 2 to 0.2 Hz to give the PMC some more staying power at DC. The loop shape is now 1/f^2 in the 9-480 Hz band and so the phase dips enough to make it almost conditionally stable, but not quite.
C26 changed from ??? + a 30 pF trim cap into a fixed NNN pF cap to set the notch frequency for the 14.5 kHz body mode that we measured. Once our brick configuration is more settled we can increase the Q of this notch from small to big.
Grounded pin 5 of U14 & U15 (AD620). These have sometimes been used as "differential" drivers in LIGO by connecting this reference voltage pin to the remote ground of the next board. This has always lead to insidious oscillation and noise. This beauty also has an output noise of 100 nV/rHz. Just never use this chip if you can help it; we can make true differential drivers - we have the technology.

Of course, we didn't have a current version of a schematic sitting around so I printed out a Rev E schematic and marked it up with red pen. I'll post pictures later and put the schematic into the PSL schematics notebook. Would be useful to take the old schematic and update it in Acrobat so that we have something electronic.

907

Mon Sep 1 04:34:00 2008

FSS loop transfer functions

I started from 6th item in Yoichi's todo list.

1) Increased the setpoint of the thermostat next to the framebuilder from 73F to 79F. Its freezing over there
in the room with the drill press. Steve's illegal mercury thermometer is reading 19 C.

2) Looked the RFPD's output spectrum using the 20 dB coupled output from the coupler that's in-line.
The first attached PDF file (n.pdf) has several plots:
page 1: 0-500 MHz anomolous peaks at 138 & 181 MHz but nothing too crazy
page 2: 0-100 MHz 80 MHz peak is RF pickup from the VCO Driver - not on the light
page 3: 10-30 MHz totally nuts
page 4: 18-25 MHz that's just wrong

The RF spectrum should only have some action around 21.5 MHz and a little peak at 2x 21.5 MHz. All that extra
junk means that something is broken!

3) To see if I could rid of any of the 80 MHz signal or any of that other trash from 18-25 MHz, I wound the RF cable
around a large toroidal ferrite core. This should have given us many uH of inductance for any signals common to
both the center and shield of the cable with no effect on the differential RF signals. There was no effect.

4) Next went to look at the 21.5 MHz Crystal Oscillator Reference card (D980353...I bet you can't figure out how
this one works). These have the Mini-Circuits SMA 30 MHz low pass (SLP-30) filters on both the LO and EOM outputs.

FSSLO.PNG shows the waveform after 20 dB attenuation going into a scope terminated with 50 Ohms.
FSSLO-Spec.png shows the spectrum of this signal - its pretty distorted. Here's the levels

   f (MHz) |  before filter (dBm) | after filter (dBm)
   ---------------------------------------------------
     21.5  |       -12.8               -13.1       
     43            -24                 -46
     64.5          -50               < -80
     86            -64               < -80

This would be OK after the filter, but the level is very low. Only 7dBm (accounting for my 20 dB att) !!
The FSS uses a JMS-1H mixer which needs, as everyone knows, a +17 dBm LO signal. Que lastima.

There seems to be something wrong already, but wait...

5) PC25.PNG shows the output signal going to the EOM from 0 - 25 MHz. The step that's visible there at
around 10 MHz is just something inherent to the analyzer (??). But see all that crap there down below
5 MHz ? That is NOT supposed to be there.

pc.pdf shows on the first page the comparison in EOM drive with 2 different slider values on the
RF AM adjust screen for the FSS. But page 2 is the punchline of this long entry: There is a bunch of
excess junk on the drive signal going to the FSS's phase modulator. The FSS is then trying to handle
this extra frequency noise and getting into trouble.

We have to fix this board. I have also ordered a few SBP-21.4 from mini-circuits (SMA bandpass around 21.4 MHz)
just in case. Another option is to just replace this thing with a Marconi and an RF amp.

Attachment 1: n.pdf

Attachment 2: FSSLO.PNG

Attachment 3: FSSLO-Spec.png

Attachment 4: PC25.png

Attachment 5: pc.pdf

908

Mon Sep 1 19:23:17 2008

FSS on an auxiliary loop

Summary: The FSS is now temporarily disabled. Naturally, the MC won't lock. I will fix it tomorrow morning.

Today, I did the 4th item of my TO DO list.
Using a mini-circuit mixer and two SR560s, I constructed an auxiliary servo loop for the reference cavity.
With this loop, I was able to lock the reference cavity without using the FSS box.
By locking the reference cavity with this auxiliary servo, I was able to measure the PC path transfer function.
I will post the analyzed results later.

I borrowed the PD RF and the LO signals from the main FSS loop by power splitters. Therefore, the gain of the main FSS loop
is now about 3dB low. I tried to compensate it by increasing the EOM modulation depth, but the PC path is still a bit noisy.
Probably the already too low LO power is now seriously low (the LO power cannot be changed from EPICS).
Because I did not want to leave the PC path with large output overnight (it will heat up the PA85, and might cause damage, though unlikely),
I disabled the FSS for now.

909

Tue Sep 2 07:58:34 2008

Summary

FSS & PMC LO trends for 2 years

The attached plot is a 2 year minute trend of the EPICS readback of the PMC & FSS LO Monitors (FSS_LODET & PMC_LODET).
Clearly the FSS LO has been dying for at least 2 years. The step up from 10 months
ago is probably when Rob removed a 3dB attenuator from in front of the box.

Attachment 1: psl-lo-trend.png

910

Tue Sep 2 09:58:42 2008

FSS on an auxiliary loop

Quote:

Summary: The FSS is now temporarily disabled. Naturally, the MC won't lock. I will fix it tomorrow morning.

Now I removed the power splitters for the aux. reference cavity servo. The FSS is back and the MC locks.
I'm now returning one of the active high-impedance probes to the Wilson house. They need it today.
We are left with only one active probe. If anyone finds another active probe in the 40m lab.,
please let me know (according to Rana we should have one more).

911

Tue Sep 2 10:09:03 2008

steve

head temp is cooling down

The chiller was over flowing this morning.
800 cc of water was removed.
PSL-126MOPA_HTEMP peaked at 20.7 C (normal is 18.7 C)

912

Tue Sep 2 14:28:41 2008

FSS EOM driving signal spectra

Rich advised me to change the +10V input of the FSS crystal frequency reference board from whatever voltage supply we use now to a nice one.
This voltage is directory connected to the signal lines of both LO and RF output amps. Therefore, fluctuations in the voltage directly appear
in the outputs, though DC components are cut off by the AC coupling capacitors.

I changed the source of this voltage from the existing Sorensen one to a power supply sitting next to the rack.
The attached plots shows the difference of the RF output spectra between the two 10V sources.
The low frequency crap is almost gone in the new 10V spectrum.

I tried to increase the FSS gain with the new 10V, but still it goes crazy. I suspect it is because the LO power is too low.

Attachment 1: RFDrive1.png

Attachment 2: RFDrive2.png

913

Tue Sep 2 22:43:16 2008

Updated FSS open loop TF

Since the LO level of the FSS servo was too low, I replaced the RF oscillator board with a combination of
a Stanford signal generator and an RF amplifier.
Right now, the POY RF amplifier is used for this purpose temporarily.
Now the LO level is about 16dBm. The RF power going into the EOM is attenuated by 20dB from the LO level.
I played with the cable length to get the phase right.
Then I was able to lock the FSS with the new RF signal source.

Attached is the open loop transfer function of the current FSS. Now the UGF is a bit above 200kHz, a factor of 2 improvement.
This gain was achieved with the common gain slider at 13.5dB and the fast gain = 30dB.
With the old RF oscillator board, UGF=100kHz was achieved with the common gain =30dB. Therefore, the increase of the LO gave
us a large signal gain.

Increasing the gain further, again ,makes the PC path crazy.
Rich suggested that this craziness was caused either by the slew rate limit of the PA85 or the output voltage limit of the bypass Op-amp(A829)
is hit.

TO DO:
* Look at the error signal spectrum to see if there is any signal causing the slew rate saturation at high frequencies.
* Find out what the RF signal level for the EOM should be. 20dB attenuation is an arbitrary choice.
* Find out the cross over frequency. Determine where the fast gain slider should be.
etc ...

Attachment 1: OPLTF.png

918

Thu Sep 4 00:38:14 2008

Entry 663 has a plot of this using the PSL/FSS/SLOWscan script. It shows that the SB's were ~8x smaller than the carrier.

P_carrier   J_0(Gamma)^2 
--------- = ------------
P_SB        J_1(Gamma)^2

Which I guess we have to solve numerically for large Gamma?

919

Thu Sep 4 07:29:52 2008

Quote:

Entry 663 has a plot of this using the PSL/FSS/SLOWscan script. It shows that the SB's were ~8x smaller than the carrier.

P_carrier   J_0(Gamma)^2 
--------- = ------------
P_SB        J_1(Gamma)^2

Which I guess we have to solve numerically for large Gamma?

P_carrier/P_SB = 8 yields gamma=0.67.

923

Thu Sep 4 13:48:50 2008

I scanned the reference cavity with the NPRO temperature (see the attached plot).
The power ratio between the carrier and the sideband resonances is about 26.8.
It corresponds to gamma=0.38.
The RF power fed into the EOM is now 14.75dBm (i.e. 1.7V amplitude). The NewFocus catalog says 0.1-0.3rad/V. So
gamma=0.38 is a reasonable number.

Attachment 1: RCScan.png

924

Thu Sep 4 14:43:58 2008

I have measured the PMC's open loop gain. UGF is 629.7Hz, with a phase margin of 53 degrees.

I injected into FP2 on the front panel, and measured MixOut/Source from 100Hz to 100kHz using the SR785. I did this both when the loop was open, and when the loop was closed (open the loop by enabling FP1, which breaks the loop).

We have 2 transfer functions involved: The actual open loop gain of the PMC servo loop (G1), and the gain between FP2 and the MixerOut monitor point (G2). This gives us:

TF(closed loop) = G2*(1+G1)
TF(broken loop) = G2

G1 = TF(closed)/TF(broken) - 1

This G1 is the final open loop gain, and it is plotted below.

Attachment 1: OpenLoopTF04Sept2008.png

926

Thu Sep 4 17:03:25 2008

RF oscillator noise comparison

I measured current spectra of the RF signal going to the FSS EOM.
The attachment compares the spectra between a Stanford signal generator and a Marconi.
I borrowed the Marconi from the abs. length measurement experiment temporarily.
The measurement was done using the signal going to the EOM. That means the spectra include
noise contributions from the RF amp., splitter and cables.

21.5MHz peak was not included because that would overload the ADC and I would have to use a large attenuation.
This means the measurement would be totally limited by ADC noise everywhere except for 21.5MHz.

I noticed that with the Marconi, the FSS is a little bit happier, i.e. the PC path is less loaded
(0.9Vrms with Stanford vs. 0.7Vrms with Marconi). But the difference is small.
Probably the contribution from the 77kHz harmonics in the laser light is more significant (see entry #929).
Also the peaks in the Stanford spectrum are not harmonics of 77kHz, which we see in the FSS error signal.

I returned the Marconi after the measurement to let Alberto work on the abs. length measurement.

Attachment 1: RFSpectra.png

927

Thu Sep 4 17:12:57 2008

I changed the gain settings of the FSS servo.
Now the Common Gain is 5dB (the last night it was 2dB) and the Fast Gain is 12dB (formerly 16dB).
I measured the open loop TF with this setting (the attachment).
I also plotted the OPLTF when CG=2dB, FG=20.5dB. With this setting, the MC looses lock every 30min.

You can see that the OPLTF is smoother with FG=12dB.
When the FG is high, you can see some structure around 250kHz. This structure is reproducible.
This may be some interruption from the fast path to the PC path through a spurious coupling.

Attachment 1: FSS-OPLTFs.png

929

Thu Sep 4 17:44:27 2008

FSS error signal spectrum

Attached is a spectrum of the FSS error signal.
There are a lot of sharp peaks above 100kHz (the UGF of the servo is about 200kHz).
These are mostly harmonics of 77kHz. They are the current suspects of the FSS slew rate saturation.
I remember when I blocked the light to the PD, these peak went away. So these noises must be
in the light. But I checked it a few weeks ago. So I will re-check it later.

One possible source of the lines is a DC-DC converter in the NPRO near the crystal.
We will try to move the converter outside of the box.

Attachment 1: FSS-Error-Spe.png

931

Fri Sep 5 08:34:03 2008

The MC is happy.
The MZ can be locked if you move the slider by hand.

Attachment 1: mzhv.jpg

937

Mon Sep 8 15:38:57 2008

POY RF amp is back to its original task

I temporarily fixed the busted ZHL-32A RF amplifier's power connector by simply soldering a cable to the internal circuit and pulling the cable out of the box through a hole for the power connector.
So I released the POY RF amplifier from the temporary duty of serving the FSS RF distribution and put it back to the original task,
so that Rob can finally re-start working on the lock acquisition.
Now the temporarily fixed ZHL-32A is sitting next to the IOO rack along with the power supply and a Stanford signal generator.
Please be careful not to topple over the setup when you work around there. They will be there until Peter's Wentzel RF box arrives.

951

Tue Sep 16 16:47:01 2008

pete

Prototype FSS reference installed

After verifying output, I installed the new prototype 21.5 MHz FSS reference (Wenzel crystal oscillator and ZHL-2 amp). Yoichi and I successfully locked the MC, and have left the new reference in place. It's temporarily sitting on the corner of the big black optics table (AP table?).

954

Wed Sep 17 13:43:54 2008

I'm doing a cavity sweep of the RC. Please leave the IFO untouched until the meeting is over.

957

Wed Sep 17 15:22:31 2008

Quote:

I'm doing a cavity sweep of the RC. Please leave the IFO untouched until the meeting is over.

The measurement is still going on.
I will post an entry when it is done.
Thank you for the patience.

958

Wed Sep 17 17:31:24 2008

I calibrated the reference cavity error signal with the following procedure.

(1) I disconnected the PC path BNC cable and locked the RC only using the PZT. To do so, I had to insert a 20dB attenuator
in the RF signal path going to the EOM to reduce the gain of the loop sufficiently.
The normal RF level going to the EOM is 17dBm. With the attenuator it is of course -3dBm.

(2) Using the SR785, I injected signal into the Test-IN2 (a sum-amp after the mixer) of the FSS box and measured the TF from the Ramp-IN to the IN1.
When the Ramp-In switch is off, the Ramp-IN port can be used as a test point connected to the PZT drive signal path just before the output.
There is a RC low-pass filter after the Ramp-IN. IN1 is the direct output from the mixer (before the sum-amp).
The attm1 is the measured transfer function along with the fitting by a first order LPF.
From this measurement, the DC transfer function from the applied voltage on the PZT to the error signal is determined to be 163.6 (V/V).
Since the RF level is lowered by 20dB, the cavity gain in the normal operation mode is 10 times larger (assuming that the modulation depth is
linearly proportional to the applied voltage to the EOM).

(3) According to elog:791, the conversion factor from the voltage on the PZT to the frequency change of the NPRO is 11.172MHz/V. Combining this with the
number obtained above, we get 6.83kHz/V as the calibration factor for converting the error signal (mixer output) to the frequency at DC.
Using 38kHz cavity pole frequency, the calibration factor is plotted as a function of frequency in the attm2.

(4) I took a spectrum of the error signal of the FSS and calibrated it with the obtained calibration factor. See attm3.
The spectrum was measured by SR785. I will measure wide band spectra with an RF analyzer later.

TO DO:
1: Use the actual modulation depth difference to extrapolate the calibration factor obtained by with a low RF signal for the EOM.
The cavity sweep was already done.

2: I assumed 38kHz cavity pole. I will measure the actual cavity pole frequency by cavity ringdown.

3: Measure out-of-the-loop spectrum of the frequency noise using PMC and MC.

Attachment 1: PZTresp.png

Attachment 2: Calibration.png

Attachment 3: FreqNoiseSpectrum.png

959

Wed Sep 17 17:58:35 2008

The cavity sweep is done. The IFO is free now.

967

Thu Sep 18 23:31:26 2008

ISS: Saturating too often at nominal gain

The ISS has been saturating whenever the MC relocks and puts the gain up to +8dB. I have
lowered the gain to +1 dB for now to stop this, but we need to revisit the ISS loop and
performance. Stefan can fix it up for us as penance when he returns from the hedonism of Amsterdam.

Attachment 1: FIRE_BLOWER.jpg

971

Fri Sep 19 08:09:55 2008

I have just turned on the PSL HEPA filters at 60% operational speed.

978

Mon Sep 22 18:54:54 2008

PMC transfer functions with various brick-on-top configurations

Attached below is a graphical summary of different things that I have tried putting on the PMC to reduce the noise in the loop. The motivation behind these measurements is the current inability here at the 40m to increase the UGF of the PMC. This is part of a broader ISS loop/gain/noise problem that we are having, which is causing Rob's locking efforts to have trouble. (The ISS is next on the to-do list, after we find the best configuration for the PMC, if we are still having problems). Right now, it looks like we are being limited by the gain of the PMC (as mentioned by Rana in elog #968).

Anyhow, Rana and I had noticed that piling heavy things on top of the PMC seemed to reduce the noise. What follows are the transfer functions that I took with the different items on top of the PMC, so that we can compare their effects:

Nothing on the PMC (like it used to be)
New ~14kg lead brick wrapped in copper foil on top of the PMC
A stack of a piece of aluminum, a chunk of steel, and then the lead brick on top of the PMC
The lead brick + Rob pushing on top of the PMC

Unfortunately, I need to retake the power spectra in these configurations, but from eye-balling it, as one might expect, pushing on the PMC with a hand added more noise than the nominal nothing-on-PMC configuration.

Also unfortunately, none of these configurations seems to have significantly helped our noise reduction situation. We need a new plan. Rana is currently trying out some other configurations, including just aluminum+brick.

Attached is an open loop gain TF from 100Hz - 100kHz. Below that is a zoomed-in version from 5kHz - 30kHz. As you can see more clearly in the zoomed in version, the notch that Rana put onto the board at ~14.5kHz is working, but we need to make the notch deeper, to catch more of that 14.5kHz peak. We're going to try removing the resistor or reducing it's value in the RLC filter on the board (see elog #906). Also, we see that there is a giant peak at 18.3kHz. This is probably much more limiting to our stability at this point than the 14.5kHz peak. We need to add another filter to take care of this, or find another way to reduce this peak. Note that it is present even when there is no brick on the PMC, so it is not an artifact of the new brick.

Attachment 1: PMC_OLG_100Hz_to_100kHz.png

Attachment 2: PMC_OLG_5kHz_to_30kHz.png

980

Mon Sep 22 21:30:06 2008

The MC refl power was going up and the FSS PC drive was so large that I had to turn up the FSS
common gain from 1.5 dB to 10.5 dB to get it to be better. Attached are the before (REF) and
after plots of frequency noise. Is the FSS gain really supposed to be 1.5 dB?? How did we gain
so many dB's of optical gain? Is there a loop measurement from after Peter's oscillator change?

Attachment 1: DAQ.png

982

Mon Sep 22 22:24:19 2008

Quote:

The MC refl power was going up and the FSS PC drive was so large that I had to turn up the FSS...

Looks like I bumped the PS for the 21.5 MHz test setup and changed the supply voltage of the amplifier
from +24 to +38 V. This made the amplifier go hot after a few hours and the output eventually dropped.

Yoichi and I walked out there now and it was too hot to touch. We turned it off and put it on a heat
sink to make it chill out and it came back after a few minutes. We have set the input to the amp to
be -7 dBm instead of -8 dBm after deciding that we should take into account the 1 dB loss in the cable
run and deliver a real +17 dBm to the mixer.

The right way is to calibrated the LO mon of the FSS.

984

Tue Sep 23 11:17:59 2008

The PMC output side has a new madly scattering spot at chamfer 2 o'clock position

Attachment 1: rainbow.png

Attachment 2: pmcclip.png

986

Tue Sep 23 15:28:06 2008

pete

new 21.5 MHz FSS reference installed

The new 21.5 MHz FSS reference is now installed in the rack with the 7 Sorensen PS. Both outputs give 18.7 dBm. The MC seems happy.

Bob did the +24 V and +15 V hookups for the amp and the Wenzel oscillator respectively, off of the din strips on the right of the rack.

I have attached two photographs. One shows the front of the box as mounted in the rack, and the other shows the inside of the box. From the second photo the circuit is apparent. Black wire coming in has ground, green has +15, and white has +24. After the switches, ground and +15 go to the Wenzel crystal oscillator, and ground and +24 go to the mini-circuits amp. There is 5 dB attenuation between the Wenzel 21.5 MHz output and the amp input. There is 3 dB attenuation between the amp output and the splitter.

The Wenzel crystal oscillator is their "streamline" model, and puts out 13.2 dBm. The amp is mini-circuits ZHL-2.

Attachment 1: fss_ref_001.jpg

Attachment 2: fss_ref_013.jpg

989

Thu Sep 25 02:35:21 2008

It looks like the FAST signal has started moving a lot - this is partly what inspired us to tune the SLOW loop.

Some of the spiking events happen when people go on the table or the MC loses lock. But at other times it just
spikes for no apparent reason. You can also see from the first plot (9 day 10-minute trend) that there is no
great change in DTEC so we shouldn't be worried about clogging in the NPRO head.

The second plot is a 1 day minute-trend.

Attachment 1: Untitled.png

Attachment 2: Untitled.png

991

Thu Sep 25 10:48:29 2008

FSS calibration again

I did a calibration of the FSS error signal again with a different method.
This time, I swept the laser frequency with the NPRO PZT around a resonance.
The attached figures show the transmitted light power and the PDH error signal vs the applied voltage to the PZT.
From the width of the transmitted light power peak, we can obtain the PZT voltage to the laser frequency coefficient,
i.e. the HWHM (Half Width Half Maximum) equals to the FSR (38kHz).
Once the PZT is calibrated, the PDH error signal can be calibrated by the fitting the central slope with a line.

I repeated the measurement 8 times and fitted the obtained data to get the HWHM and the slope.
The results are the following:
PZT calibration = 6.3 +/-0.1 MHz/V
PDH calibration = 6.5 +/-0.5 kHz/V

Note:
(1) The calibration coefficient (6.5kHz/V) is almost consistent with the previous value (6.83kHz/V elog:958). However, that calibration
used a considerably different value for the PZT calibration (11.172MHz/V elog:791). The discrepancy in the PZT calibration is understandable
because my previous PZT calibration was very rough. The fact that the two calibrations still agree is a mystery.

(2) In the transmitted power curve, there seems to be a slight distortion, probably due to the thermal effect.
We should reduce the power to the reference cavity to remove this effect.

(3) This measurement was done after Peter installed his RF source. The demodulation phase had not yet been optimized. We should
repeat the calibration after he optimizes the phase.

(4) I used the Tektronix oscilloscope for the measurement.
Using the ethernet-wireless converter, you can connect the scope to the network from anywhere in the lab.
No hard wire required anymore.
Then you can download the data from a web browser. It is cool !

Attachment 1: PDTrans.png

Attachment 2: PDHsignal.png

995

Fri Sep 26 00:19:54 2008

Filter-action with the PMC

Written, but not posted on 24Sept2008:

PMC adventures for this evening

Today's mission was to make more progress on increasing the bandwidth of the PMC servo.

First order of business was to improve the performance of the 14.6kHz notch that Rana put in the PMC servo board a few weeks ago to remove the 14.6kHz body mode resonance of the PMC. Looking at the zoomed in TF that I posted Monday (elog #978), we see that there is still a remnant of a peak near 14.5kHz. A first gut-reaction is that the notch is not tuned properly, that we have just missed the peak. As previously noted in the elog, the peak that we are trying to notch out is at 14.68kHz (elog #874). By unlocking the PMC and measuring the transfer function between FP2 and OutMon (OutMon is the monitor for the high voltage going to the PMC's PZT), I measure the transfer function of the notch, and find that it is notching at 14.63kHz. So we're a teensy bit off, but the Q of the notch is such that we're still getting improvement at the peak frequency. After checking that we are hitting the correct frequency, I put a short (just some wire) around R21, which is the R in the RLC notch filter, to increase the depth of the notch. At the peak frequency of 14.68kHz, we see a 2.5dB improvement of the notch. At the actual notch frequency of 14.63kHz, we see a 3.2dB increase in the depth of the notch. So, shorting R21 helped a little, but not a lot. Also, it's clear that we don't get that much more improvement by being on the resonant frequency, so there's no need to go in and tune the notch on the board.

Second order of business was to investigate the 18.34kHz peak in the transfer function. (Rana spent some time Monday night measuring this peak, and determined that it was at 18.34kHz) We decided that the best plan was to re-implement the Pomona Box notch filter that had previously existed to remove a higher frequency body mode, but tuned for the 18.34kHz mode. I am still not entirely sure what this mode is, but clearly it's a problem by about 20dB (on the TF, the next highest peak is 20dB below the 18.34kHz peak). Unfortunately, while the components should, by Matlab calculations, give me an 18.3kHz notch, I ended up with something like a 21.7kHz notch. This notch is approximately -30dB at 21.7kHz, and -20dB at 18.3kHz. I still need to take transfer functions and power spectra of the PMC servo with this new filter in place to (a) confirm that it did some good, and (b) to determine how important it is that the notch be right-on. More likely than not, I'll take the filter out and fiddle with the capacitors until I get the correct notch frequency.

Third on the list was to lock everything back up (FSS, PMC) after my tinkering, and see what kind of gain we get. Rob and I fiddled with the PMC gain, and it looks like the servo oscillates just before we get up to the max slider gain of 30dB. Looking at the power spectra in DTT, we do not see any significant peaks that suggest oscillation, so it is likely that there is some investigation to be done at frequencies above the 7kHz that we were able to look at with DTT (which isn't surprising, since all of this work has been at 14kHz and higher).

A final note is that we see a feature around 9kHz in the transfer function, and it is not at all clear where it comes from. At this time, it does not seem to be the dominant feature preventing us from increasing the gain, but at some point if we want the bandwidth of the PMC servo to be 10kHz, we'll have to figure this one out.

Still on the PMC todo list:

Measure the new transfer function, see if 18.34kHz peak is reduced
Tune Pomona Box notch filter to 18.3kHz instead of the current 21.7kHz
Retake power spectra of different items on top of PMC, compare to see if there is any one configuration that it obviously better than the others.
Find out why the PMC still oscillates when we try to take it up to the max slider gain, and fix it.

PS, is anyone else having trouble getting to the elog from laptops on other parts of the Caltech network (but not LIGO network)? My laptop won't go to the elog, but I can get to the rest of the internet using the Caltech wireless. My computer stopped seeing the elog on Tuesday or so. Joe, do you have any inspiration? Thanks.

1000

Fri Sep 26 18:35:17 2008

PMC filter is out for tuning

The PMC's new Pomona Box filter is out for tuning. I'd like to get the notch right on the 18.3kHz, rather than being off in 21.7kHz land.

1001

Fri Sep 26 19:08:43 2008

Refcav Trans: PD + Camera + Dumps

I went out to improve the Refcav trans path.

I removed all ND filters to get rid of the fringing.

I removed the anodized Al dump that was there. Black anodized Aluminum dumps are forbidden for use as
dumps in any low phase noise setup (such as our frequency stabilization cavity). The scatter was going
directly back into the cavity and making noise. For now its undumped, but Steve will find the
reflections and dump them on unblemished razor blade dumps mounted stiffly.

I will post a photo of the new setup later - the new setup is sketched on the control room markerboard.

The transPD level is now 8 V, up from its previous 3-4 V. We will probably have to also put a lens
in front of it to get the beam size down.

1003

Mon Sep 29 01:19:40 2008

Summary

Laser chiller running a little hot

I looked at it some last night and my suspicion was the ISS. Whenever the ISS switch came on the FAST got a kick.

We should try to disable the MC locking and ISS and see if the FSS/PMC/MZ are stable this way. If so this may be
a problem with the ISS / Current Shunt.

1005

Mon Sep 29 13:23:40 2008

rob

Summary

Laser chiller running a little hot

Quote:

My entry about the laser chiller got deleted. The PSL appears to be running with the ISS gain at -5dB, so that's good, but the
chiller is still showing 21+ degrees. It should be at twenty, so there's something causing it to run out of
headroom. We'll know more once Yoichi has inspected the ISS.

In the deleted entry I noted that the VCO (AOM driver), which is quite warm, has been moved much closer to the MOPA.
This may be putting some additional load on the chiller (doubtful given the amount of airflow with the HEPAs on,
but it's something to consider).

1007

Mon Sep 29 15:09:36 2008

steve