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  15343   Fri May 22 01:43:18 2020 gautamUpdateElectronicsRF electronics trouble

To test a hypothesis, I have left the PSL shutter closed. I notice significant glitches in the dark electronics offsets on all the 11 MHz photodiode I/Q demodulated input channels, which appear coherent. These are non-negligible in magnitude - for now they are uncalibrated in cts, but for an estimate, the POX11 channel shows a shift of ~20 cts (~200uV at the input to the whitening board), while the PDH fringe is ~200 cts pk2pk. A first look is in Attachment #1. The fact that it's in all the 11 MHz channels makes me suspect something in the RF chain, maybe some amplifier? I'll open the shutter tomorrow.

Attachment 1: RFPDglitches.png
RFPDglitches.png
  15347   Tue May 26 01:58:57 2020 gautamUpdateElectronicsSome electronics thoughts

A big factor in how much IFO locking activities can take place is how cooperative the IMC is.

Since the c1psl upgrade, the IMC duty cycle has definitely deteriorated. I took a measurement of the dark noise at the IMC error point with 1 Hz FFT binwidth, with all electrical connections to the IMC servo board except the Acromag and Eurocrate power disconnected. I was horrified at the prominence of 60 Hz harmonics - see Attachment #1. In the past, this kind of feature has been indicative of some error in the measurement technique - but I confirmed that the lines remain even if I unplug the GPIB box, and all combinations of floating/grounded inputs that I tried. We know for sure that there is some excess noise imprinted on the laser light post upgrade. While these lines almost certainly are not responsible for the PCdrive RMS going bonkers, surely this kind of electrical situation isn't good?

Attachment #2 shows the same information translated to frequency noise units, taking into account the complementary sensitivity function, L/(1+L) - the sum contribution of the 60 Hz peaks to the RMS is ~11.5% of the total over the entire band (c.f. 1.7 % that is expected if the noise at multiples of 60 Hz was approximately equal to the surrounding noise levels). Moreover, the measured RMS is 55 times higher than a LISO model. 

How can this be fixed?

Attachment 1: IMCsensingNoise.pdf
IMCsensingNoise.pdf
Attachment 2: IMCsensingNoise.pdf
IMCsensingNoise.pdf
  15365   Wed Jun 3 01:40:13 2020 gautamUpdateElectronicsMore electronics woes

There were many locklosses from the point where the arm powers were somewhat stabilized. Attachments #1 and #2 show two individual locklosses. I think what is happening here is that the BS seismometer X channel is glitching, and creating a transient in the angular feedforward filter that blows the lock. The POP QPD based feedback loop cannot suppress this transient, apparently. For now, I get around this problem by boosting the POP QPD feedback loop a bit, and then turning the feedforward filters off. The fact that the other seismometer channels don't report any transient makes me think the problem is either with the seismometer itself, or the readout electronics. The seismometer masses were recently recentered, so I'm leaning towards the latter.

I didn't explicitly check the data, but I am reasonably certain the same effect is responsible for many PRMI locklosses even with the arms held off resonance (though the tolerance to excursions there is higher). Pity really, the feedforward filters were a big help in the lock acquisition...

Attachment 1: glitchySeis2.png
glitchySeis2.png
Attachment 2: glitchySeis3.png
glitchySeis3.png
  15433   Fri Jun 26 16:53:38 2020 gautamUpdateElectronicsRFPD characterization

Summary:

While the vacuum system was knocked out, I measured the RF transimpedance (using the AM laser setup, didn't do the shot noise intercept current measurement for now) of all the RFPDs (except PMC REFL). At the very least, the following photodiodes are suspect:

  1. WFS heads - expected transimpedance is 50 kohm unattenuated, and 5 kohm attenuated. I measure values that are x10 lower than this, and the segments are significantly imbalanced. Morover, the attenuators for some quadrants appear to do nothing. This could be a problem with the Acromag system I guess, but the measured transimpedance is nowhere close to the "expected" value. See Attachments #1 and #2. You can also see that the response at 55 MHz is significantly attenuated, so I'm guessing trying to measure the AS port ASC sensing response is going to be difficult.

    Note that I assumed a 1kohm DC transimpedance, which is what I expect from the schematic and also is consistent with the DC voltage I measured, knowing the approximate optical power incident on the photodiode.
  2. POP 22/ POP 110 - this is a Thorlabs PDA10CF diode. It should have a flat gain profile out to ~100 MHz, but I measure some weird features. The other PDA10CF we use, at AS110, shows a more reasonable response. See Attachment #3. I don't know what kind of failure mode this is? Anyway I'll try testing another PDA10CF and if it looks more reasonable, I'll switch out this diode. FWIW, the measured AS110 gain is ~3kohms, whereas the datasheet tells us to expect 5 kohms.

For the remaining photodiodes, I measure a transimpedance that is within ~20% of what is on the wiki page. The notches may benefit from some retuning. While I have the data, I will fit this and post a more complete report on the wiki.

Update July 6 1145am: WFS response plots now have legends mapping quadrants, and I've also added the response of a spare PDA10CF (which is now the new POP22/POP110 photodiode).

Attachment 1: WFS1.pdf
WFS1.pdf
Attachment 2: WFS2.pdf
WFS2.pdf
Attachment 3: buildupMons.pdf
buildupMons.pdf
  15439   Mon Jun 29 15:56:02 2020 gautamUpdateElectronicsRFPD characterization

A more comprehensive report has been uploaded here. I'll zip the data files and add them there too. In summary:

  1. There are several problems with the WFS heads
    • Some attenuators don't seem to work. This could be a problem with the Acromag BIO, or with the relay on the head itself.
    • The measured transimpedance at 29.5 MHz is much lower than expected. We expect ~50 kohms with no attenuation, and 5 kohms without. I measure 100 ohm - 2 kohm with the attenuation disabled, and ~200 ohms with it enabled.
    • Quadrant #3 on both WFS heads behaves differently from the others. There is also evidence of a 200 MHz oscillation for quadrant 3.
    • For some reason, there is a relative minus sign between the TFs measured for the WFS and for the RFPDs. I don't understand where this is coming from - all the OpAmps in the LSC PDs and WFS heads are configured as non-inverting, so why should there be a minus sign? Is this indicative of the polarity of the LEMO output being somehow flipped?
  2. POX 11 photodiode does not have a notch at 22 MHz.
  3. AS55 resonance appears to have shifted closer to 60 MHz, would benefit from a retuning. But the notches appear fine.
  4. PDA10CF photodiode used as the POP22/POP110 readback appears broken in some strange way. As shown in the linked document, a spare PDA10CF in the lab has a much more reasonable response, so I am going to switch out the POP22/POP110 diode with this spare.

I'll upload the data and analysis notebook + liso fit files to the wiki as well shortly. The data, a Jupyter notebook making the plots, and the LISO fit files have been uploaded here.

I didn't do it this time but it'd be nice to also do the noise measurement and get an estimate for the shot-noise intercept current.

Quote:

While I have the data, I will fit this and post a more complete report on the wiki.

  15443   Tue Jun 30 22:00:04 2020 gautamUpdateElectronicsGlitchy POX resurfaces

This problem reared its ugly head again. I am inclined to believe the problem is electronic and not on the light, since the POY channels seem immune to this issue (see Attachment #1). I will investigate in the daytime tomorrow. Note that while the POX photodiode head has ~twice the transimpedance than POY (per measurement), the POY signal gets amplified by a ZHL-500-HLN amplifier before heading to the demod electronics (nominal gain is 19dB = x9). There is also some imbalance in the light level at the photodiodes I guess, because overall, the PDH fringe is ~twice as large for the Y arm as the X arm. Basically, the y-axes of the attached plot cannot be directly compared between POX and POY.

Mostly this is an annoyance - right now, the POX signal is only used for locking and dither aligning the X arm cavity, and so once that is done, the locking can proceed (as long as the other channels, e.g. REFL11, aren't glitching as well...)

Attachment 1: glitchyPOX.jpg
glitchyPOX.jpg
  15450   Sun Jul 5 18:25:42 2020 ranaUpdateElectronicsWFS characterization

in the lab, checkin on the WFS

Sun Jul 5 18:25:50 2020

I redid Gautam's measurements to get a baseline before changing the head, and my results are very different: To me it looks like the WFS2 quadrants are all OK.

 

Measurement Details:

  1. The whole AG4395 + breadboard Jenne laser is wheeled over next to the SW side of the AP table.
  2. The output of the 1611 goes into channel R of the 4395
  3. I disconnected all the LEMO cables from the head and then plugged a LEMO-BNC cable into the plugs one at a time. The existing LEMO connectors, which take the signals back to the demode board, were all a little loose, so I adjusted them with some pliers (see video).
  4. The Atten = 0 dB for all AG4395 channels
  5. Source drive = 0 dBm. Checked with a -10 dBm drive that there was no change in the observed TFs, so I guess a 0 dBm drive doesn't make things nonlinear.
  6. When I first turned the setup on, the Yellow 'limit' light was ON on the ILX laser current driver, so maybe the modulation wasn't getting to the laser diode as we wish.
  7. did not change any WFS MEDM settings for these measurements. Not sure if any of those buttons work anyway.

I've left the setup as is in case either me or Gautam want to double check. If we're agreed on this response, I'll remove the notches and disable the RF attenuators.

Sun Jul 5 21:42:45 2020

Attachment 1: WFS_attenOff.pdf
WFS_attenOff.pdf
  15472   Sun Jul 12 22:40:35 2020 gautamUpdateElectronicsWFS characterization - old SURF report

After some hunting, I found this old SURF report with the WFS head measurements. The y-axes don't make much sense to me, and I can't find the actual data anywhere (her wiki page doesn't actually exist). So I think it's still unknown if these heads ever had the advertised transimpedance gain, or if the measured transimpedance of ~1kohm was what it always was.

  15480   Tue Jul 14 16:52:47 2020 gautamUpdateElectronicsCoil drivers for the test masses

Summary:

Koji and I had a discussion last Friday about the suspension electronics. I think there are still a few open questions - see Attachment #1. We should probably make a decision on these soon.

Other useful links:

  1. High-voltage coil driver circuit - D1900163
    • This board is ready to be fabricated and tested on the bench.
    • The way the connectors J2 and J3 are designed currently is meant to interface with the existing coil driver electronics.
    • Depending on the eventual coil driver we choose for the fast path, it may be benificial to change the signals on the connectors J2 and J3, to avoid the need for a custom interface board.
  2. HAM-A coil driver noise analysis.
    • The linked attachment evaluates the noise for the design value of the fast path series resistor, which is 1.2 kohms.
    • Iff we still have ambitions of measuring ponderomotive squeezing, we will need the resistance to be much higher, ~10 kohms (in the linked noise budget, only the Johnson noise of the series resistor is considered, but in reality, the OpAmp voltage and current noises also matter). 
    • This corresponds to a maximum current of 10V/10kohms = 100uA
    • Looking at signals to the ETMs from the current lock acquisition sequence, the RMS current to a single coil is approximately _____ (to be filled in later).
    • So we may need a version of the fast coil driver that supports a low noise mode (with large series resistance) and a high-range mode (with lower series resistance for lock acquisition).
  3. You can follow the links to DCC entries for other parts from Attachment #1.
Attachment 1: coilDriverSchem.pdf
coilDriverSchem.pdf
  15488   Wed Jul 15 21:08:43 2020 gautamUpdateElectronicsETM coil outputs DQed

To facilitate this investigation, I've DQed the 4 face coil outputs for the two ETMs. EX is currently running with 5 times the series resistance of EY, so it'll be a nice consistency check. Compilation, installation etc went smooth. But when restarting the c1scx model, there was a weird issue - the foton file, C1SCX.txt, got completely wiped (all filter coefficients were empty, even though the filter module names themselves existed). I just copied the chiara backup version, restarted the model, and all was well again.

This corresponds to 8 additional channels, recorded at 16k as float 32 numbers, so in the worst case (neglecting any clever compression algorithms), we are using disk space at a rate of ~4 MB/s more. Seems okay, but anyway, I will remove these DQ channels in a few days, once we're happy we have enough info to inform the coil driver design.

spoke too soon - there was an RFM error for the TRX channel, and restarting that model on c1sus took down all the vertex FEs. Anyways, now, things are back to normal I think. The remaining red light in c1lsc is from the DNN model not running - I forgot to remove those channels, this would've been a good chance! Anyways, given that there is an MLTI in construction, I'm removing these channels from the c1lsc model, so the next time we restart, the changes will be propagated.

For whatever reason, my usual locking scripts aren't able to get me to the PRFPMI locked state - some EPICS channel value must not have been set correctly after the model reboot 😞. I'll debug in the coming days.

Fun times lie ahead for getting the new BHD FEs installed I guess 🤡 ....

Quote:
 

Looking at signals to the ETMs from the current lock acquisition sequence, the RMS current to a single coil is approximately _____ (to be filled in later).

So we may need a version of the fast coil driver that supports a low noise mode (with large series resistance) and a high-range mode (with lower series resistance for lock acquisition).

Attachment 1: CDS.png
CDS.png
Attachment 2: coilOutDQed.png
coilOutDQed.png
  15494   Mon Jul 20 17:23:46 2020 gautamUpdateElectronicsCoil drivers for the test masses

Summary:

Looking at the signals to the test mass coils, it seems borderline to me that we will be able to acquire lock and run in a low noise configuration with the same series resistor in the coil driver circuit. The way I see it, options are:

  1. Use a moderately high series resistance (e.g. 5 kohms) for the time being, and go ahead with the HAM-A coil driver.
    • This will mean a current noise of ~3pA/rtHz, which translates to ~3e-18 m/rtHz @ 100 Hz in DARM displacement noise (assuming the ITMs have much higher series resistance than the ETMs).
    • If the lock acquisiton looks smooth, double the resistance to 10 kohms.
    • With 5 kohm series resistance, there is negligible possibility of measuring ponderomotive squeezing for any of the input powers we consider feasible, but this is under the assumption that we will expose coil driver noise, which is very optimistic imho.
  2. Re-design a new coil driver that allows switchable impedance, so we can have a higher noise acquisition mode for acquiring and holding the ALS lock, then transition to a lower noise, lower range config once the RF / BHD lock has been acquired.
    • On paper, this solves all the problems, but the design of such a circuit is probably pretty non-trivial and time consuming.

Details:

I only looked at the ETMs for this study. The assumption is that we will have no length actuation on the ITMs, only local damping and Oplev loops (and maybe some ASC actuation?), which can be sufficiently low-pass filtered such that even with coil de-whitening, we won't have any range issues.

Attachment #1 shows the time-domain traces of the coil driver signals as we transition from POX/POY lock to the ALS lock. There are some transients, but I think we will be able to hold the lock even with a 5 kohm resistor (~twice what is on ETMX right now). From just these numbers, it would seem we can even go up to 10 kohms right away and still be able to acquire lock, especially if we re-design the digital feedback loop to have better low-pass filtering of the high-frequency ALS noise, see the next attachment.

Attachment #2 shows the f-domain picture, once the arm lengths are fully under ALS control (~25 seconds onwards in Attachment #1). The RMS is dominated by high frequency ALS length loop noise, which we can possibly improve with better design of the digital control loop.

Finally, Attachment #3 shows the situation once DARM control has been transitioned over to AS55_Q. Note that the vertex DoFs are still under 3f control, so there is the possibility that we can make this even lower noise. However, one thing that is not factored in here is that we will have to de-whiten these signals to low-pass filter the DAC noise (unless there is some demonstrated clever technique with noise-mons or something to subtract the DAC noise digitally). Nevertheless, it seems like we can run safely with 5 kohms on each ETM coil and still only use ~2000 cts RMS, which is ~1/10th the DAC range (to allow for dealing with spurious transients etc). 

Quote:

Looking at signals to the ETMs from the current lock acquisition sequence, the RMS current to a single coil is approximately _____ (to be filled in later).

Attachment 1: ALSlock_timeDomain.pdf
ALSlock_timeDomain.pdf
Attachment 2: ALSlock.pdf
ALSlock.pdf
Attachment 3: RFlock.pdf
RFlock.pdf
  15516   Wed Aug 12 17:42:58 2020 gautamUpdateElectronicsPhotodiode inventory

See Attachments #1 and #2. We don't have any Q3000 QPDs in hand, at least not in the photodiode box stored in the clean optics cabinet at the south end. I also checked a cabinet along the east arm where we store some photodiodes - but didn't find any there either. The only QPDs we have in hand are the YAG-444-4AH, which I believe is what is used in the iLIGO WFS heads.

So how many do we want to get?

Attachment 1: IMG_8709.JPG
IMG_8709.JPG
Attachment 2: IMG_8708.JPG
IMG_8708.JPG
  15517   Wed Aug 12 18:08:54 2020 gautamUpdateElectronicsNumber of the beast

The "source" output of the SR785 has a DC offset of -6.66 V. I couldn't make this up.

Upshot is, this SR785 is basically not usable for TF measurements. I was using the unit to characterize the newly stuffed ISC whitening board. The initial set of measurements were sensible, and at some point, I started getting garbage data. Unclear what the cause of this is. AFAIK, we don't have any knob to tune the offset - adjusting the "offset" in the source menu, I can change the level of the offset, but only by ~1 V even if I apply an offset of 10 V. I also tried connecting the ground connection on the rear of the SR785 to the bench power supply ground, no change.

Do we have to send this in for repair?

Attachment 1: IMG_8710.JPG
IMG_8710.JPG
  15519   Wed Aug 12 20:15:42 2020 KojiUpdateElectronicsNumber of the beast

Grrr. Let's repair the unit. Let's get a help from Chub & Jordan.

Do you have a second unit in the lab to survive for a while?

  15520   Wed Aug 12 20:16:52 2020 KojiUpdateElectronicsPhotodiode inventory

When I tested Q3000 for aLIGO, the failure rate was pretty high. Let's get 10pcs.

  15534   Thu Aug 20 00:21:51 2020 gautamUpdateElectronicsFirst look at HV coil driver

Summary:

A single channel of this board was stuffed (and other channels partially populated). The basic tests passed, and nothing exploded! Even though this is a laughably simple circuit, it's nice that it works.

HV power supplies:

A pair of unused KEPCO BHK300-130 switching power supplies that I found in the lab were used for this test. I pulled the programmable cards out at the rear, and shorted the positive output of one unit to the negative of the other (with both shorted to the supply grounds as well), thereby creating a bipolar supply from these unipolar models. For the purposes of this test, I set the voltage and current limits to 100V DC, 10mA respectively. I didn't ramp up the supply voltage to the rated 300 V maximum. The setup is shown in Attachment #1.

Tests:

  1. With the input to the channel shorted to ground, I confirmed with a DMM that the output was (nearly) zero (there was an offset of ~40mV but I think this is okay).
  2. Used the calibrated voltage source, and applied +/- 3 V in steps of ~0.5 V, while monitoring the output with a DMM. Confirmed the output swing of ~ +/-90 V, which is what is expected, since the design voltage gain of this circuit is 31.
  3. Drove a 0.1 Hz, 500mVpp sine wave at the input while monitoring the output and the Vmon testpoints, see Attachment #2. Note the phasing between input and output, and also the fact that the gain is slightly lower than the expected gain of 31, because there are three poles at ~0.7 Hz, which already start showing some influence on the transfer function at 0.1 Hz.
  4. Noise measurement 
    • The whole point of this circuit is to realize sub 1pA/rtHz current noise to the coil, when it is connected.
    • For this test, no load was connected (i.e. voltage noise was measured at the output of the 25 kohm resistor), and the input was shorted to ground so that the DC value of the output was close to 0 (the idea was to not overload the SR560/SR785 with high voltage).
    • An SR560 preamp with gain x50 (DC coupled) was used to preamplify the signal. This was the maximum gain that could be used with the unit DC coupled, due to the small DC offset. I opted to keep the DC coupling to get a look at the low frequency noise as well, but in hindsight, maybe I should have used AC coupling as we only care about the current noise at ~100 Hz.
    • See Attachment #3 for results. The measurement is close to the model above ~100 Hz

Need to think more about how to better characterize this noise. An estimate of the required actuation can be found here.

Attachment 1: IMG_8724.JPG
IMG_8724.JPG
Attachment 2: timeDomain.pdf
timeDomain.pdf
Attachment 3: HVampNoise.pdf
HVampNoise.pdf
  15536   Sun Aug 23 23:36:58 2020 gautamUpdateElectronicsFirst look at HV coil driver

Summary:

A more careful analysis has revealed some stability problems. I see oscillations at frequencies ranging from ~600kHz to ~1.5 MHz, depending on the voltage output requested, of ~2 V pp at the high-voltage output in a variety of different conditions (see details). My best guess for why this is happening is insufficient phase margin in the open-loop gain of the PA95 high voltage amplification stage, which causes oscillations to show up in the closed loop. I think we can fix the problem by using a larger compensation capacitor, but if anyone has a better suggestion, I'm happy to consider it

Details:

The changes I wanted to make to the measurement posted earlier in this thread were: (i) to measure the noise with a load resistor of 20 ohms (~OSEM coil resistance) connected, instead of the unloaded config previously used, and (ii) measure the voltage noise on the circuit side (= TP5 on the schematic) with some high voltage output being requested. The point was to simulate conditions closer to what this board will eventually be used in, when it has to meet the requirement of <1pA/rtHz current noise at 100 Hz. The voltage divider formed by the 25 kohm series resistor and the 20 ohm OSEM coil simulated resistance makes it hopeless to measure this level of voltage noise using the SR785. On the other hand, the high voltage would destroy the SR785 (rated for 30 V max input). So I made a little Pomona box to alllow me to do this measurement, see Attachment #1. Its transfer function was measured, and I confirmed that the DC high voltage was indeed blocked (using a Fluke DMM) and that the output of this box never exceeded ~1V, as dictated by the pair of diodes - all seemed okay .

Next, I wanted to measure the voltage noise with ~10mA current flowing through the output path - I don't expect to require more than this amount of current for our test masses. However, I noticed some strange features in the spectrum (viewed continuously on the SR785 using exponential averaging setting). Closer investigation using an oscilloscope revealed:

  1. 600kHz to 1 MHz oscillations visible, depending on output voltage.
  2. The oscillations vanish if I drive output above +30 V DC (so input voltage > 1 V).
  3. The oscillations seem to be always present when the output voltage is negative.
  4. No evidence of this offset if circuit is unloaded and voltage across 25k resistor is monitored. But they do show up on scope if connected to circuit side even in this unloaded config.

Some literature review suggested that the capacitor in the feedback path, C4 on the schematic, could be causing problems. Specifically, I think that having that capacitor in the feeddback path necessitates the use of a larger compensation capacitor than the nominal 33pF value (which itself is higher than the 4.7pF recommended on the datasheet, based on experience of the ESD driver circuit which this is based on, oscillations were seen there too but the topology is a bit different). As a first test of this idea, I removed the feedback capacitor, C4 - this seemed to do the trick, the oscillations vanished and I was able to drive the output between the high voltage supply rails. However, we cannot operate in this configuration because we need to roll off the noise gain for the input voltage noise of the PA95 (~6 nV/rtHz at 100 Hz will become ~200 nV/rtHz, which I confirmed using the SR785). Using a passive RC filter at the output of the PA95 (a.k.a. a "snubber" network) is not an option because we need to sum in the fast actuation path voltage at the output of the 25 kohm resistor.

Some modeling confirms this hypothesis, see Attachment #2.  The quantity plotted is the open-loop gain of the PA95 portion of the circuit. If the phase is 0 degrees, then the system goes unstable.

So my plan is to get some 470pF capacitors and test this idea out, unless anyone has better suggestions? I guess usually the OpAmps are compensated to be unconditionally stable, but in this case maybe the power op-amp is more volatile?

Quote:

Need to think more about how to better characterize this noise. An estimate of the required actuation can be found here.

Attachment 1: IMG_5379.JPG
IMG_5379.JPG
Attachment 2: stabilityCriterion.pdf
stabilityCriterion.pdf
  15542   Wed Aug 26 16:12:25 2020 gautamUpdateElectronicsTest mass coil current requirements

Attachment #1 is a summary of the current to each coil on the suspensions. The situation is actually a little worse than I remembered - several coils are currently drawing in excess of 10mA. However, most of this is due to a YAW correction, which can be fixed somewhat more easily than a PIT correction. So I think the circuit with a gain of 31 for an input range of +/-10 V, which gives us the ability to drive ~12mA per coil through a 25kohm series resistor, will still provide sufficient actuation range. As far as the HV supplies go, we will want something that can do +/- 350 V. Then the current to the coils will at most be ~50 mA per optic. The feedback path will require roughly the same current. The quiescent draw of each PA95 is ~10mA. So per SOS suspension, we will need ~150mA.

If it turns out that we need to get more current through the 25kohm series resistance, we may have to raise the voltage gain of the circuit. Reducing the series resistance isn't a good option as the whole point of the circuit is to be limited by the Johnson noise of the series resistance. Looking at these numbers, the only suspension on which we would be able to plug in a HV coil driver as is (without a vent to correct for YAW misalignment) is ITMY.


Update 2 Sep 2020 2100: I confirmed today that the number reported in the EPICS channel, and the voltage across the series resistor, do indeed match up. The test was done on the MC3 coil driver as it was exposed and I didn't need to disable any suspensions. I used a Fluke DMM to measure the voltage across the resistor. So there is no sneaky factor of 2 as far as the Acromag DACs are concerned (unlike the General Standards DAC).

Attachment 1: coilCurrents.png
coilCurrents.png
  15543   Wed Aug 26 22:49:47 2020 gautamUpdateElectronicsCheckout of Trek Model 603

I unboxed the Trek amplifier today, and performed some basic tests of the functionality. It seems to work as advertised. However, we may have not specified the correct specifications - the model seems to be configured to drive a bipolar output of +/- 125 V DC, whereas for PZT driving applications, we would typically want a unipolar drive signal. From reading the manual, it appears to me that we cannot configure the unit to output 0-250V DC, which is what we'd want for general PZT driving applications. I will contact them to find out more. 

The tests were done using the handheld precision voltage source for now. I drove the input between 0 to +5 V and saw an output voltage (at DC) of 0-250 V. This is consistent with the voltage gain being 50V/V as is stated in the manual, but how am I able to get 250 V DC output even though the bipolar configuration is supposed to be +/- 125 V? On the negative side, I am able to see 50V/V gain from 0 to -1 V DC. At which point making the input voltage more negative does nothing to the output. The unit is supposed to accept a bipolar input of +/- 10 V DC or AC, so I'm pretty sure I'm not doing anything crazy here...

Update:

Okay based on the markings on the rear panel, the unit is in fact configured for unipolar output. What this means is we will have to map the +/- 10 V DC output from the DAC to 0-5 V DC. Probably, I will stick to 0-2.5 V DC for a start, to not exceed 125 V DC to the PI PZT. I'm not sure what the damage spec is for that. The Noliac PZT I think can do 250 V DC no problem. Good thing I have the inverting summing amplifier coming in tomorrow...

Attachment 1: IMG_8951.JPG
IMG_8951.JPG
  15547   Sat Aug 29 20:07:48 2020 ranaUpdateElectronicsWFS characterization

I set up to do the WFS head modifications today, but I was shot down in flames due to a missing AC/DC adapter.

The Prologix GPIB-ethernet dongle needs +8-13 V to run. Some riff raff has removed the adapter and I was thunderstruck to see that it had not been returned.

I did the usual hunt around the lab looking for something with the right specs and connector. I found one that could do +9V and had the right connector, but it didn't light up the adapter so I put it back in black SP table.

I'll order a couple of these (5 ordered for delivery on Wednesday) in case there's a hot demand for the jack / plug combo that this one has. The setup is in the walkway, but I returned the AS table to the usual state and made sure the IMC is locking well.

  15548   Sat Aug 29 22:10:09 2020 gautamUpdateElectronicsWFS characterization

Clearly this "riff raff" is referring to me. It won't help today I guess but there is one each on the carts holding the SR785 (currently both in the office/electronics bench area), and the only other unit available in the lab is connected to a Prologix box on the Marconi inside the PSL enclosure. 

Quote:

The Prologix GPIB-ethernet dongle needs +8-13 V to run. Some riff raff has removed the adapter and I was thunderstruck to see that it had not been returned.

  15551   Tue Sep 1 01:49:49 2020 KojiUpdateElectronicsTeledyne AP1053 etc were transported

Teledyne AP1053 etc were transported from Rich's office to the 40m. The box is placed on the shelf at the entrance.

My record tells that there are 7 AP1053 in the box. I did not check the number this time.

Attachment 1: 20200831203756_IMG_9931.jpg
20200831203756_IMG_9931.jpg
Attachment 2: 20200831203826_IMG_9932.jpg
20200831203826_IMG_9932.jpg
Attachment 3: 20200831205126_IMG_9934.jpg
20200831205126_IMG_9934.jpg
  15552   Tue Sep 1 15:39:04 2020 gautamUpdateElectronicsHV coil driver oscillations fixed

Summary:

Increasing the compensation capacitance (470 pF now instead of 33 pF) seems to have fixed the oscillation issues associated with this circuit. However, the measured noise is in excess of the model at almost any frequency of relevance. I believe the problem is due to the way the measurement is done, and that we should re-do the measurement once the unit is packaged in a shielded environment.

Details:

Attachment #1 shows (schematically) the measurement setup. Main differences from the way I did the last round of testing are:

  1. A 20 ohm series resistor was connected between the high voltage output and ground to simulate the OSEM coil.
  2. The test was done under driven conditions (i.e. some non-zero input voltage) to better simulate conditions under which the circuit will be used.
  3. An Acromag XT1541 DAC was used to provide the input signal, to simulate more realistic operating conditions.
  4. A pomona box filter was used to block out the high voltage DC signal which would otherwise destroy the SR785.

Attachment #2 shows the measurement results:

  • Tests were done at a few different drive levels to check if there was any difference.
  • The difference between "Idrive=0mA" and "Input Grounded" traces is that in the former, the Acromag DAC was connected but putting out 0V, wheras in the latter, I shorted the input to the circuit ground.
  • Because the measurement was done at the output of the PA95, the Johnson noise of 25 kohms (~20 nV/rtHz) was manually summed in quadrature to all the measured traces.
  • The plotted spectra were collected in 3 spans, 0-200 Hz, 200Hz-1.8kHz, and 1.8kHz-14.6kHz. The input range was kept constant, so I'm not sure what to make of the discontinuity around 1.8 kHz. Maybe the comb of lines that were being picked up were distorting the spectra for lower frequencies?
  • The "Model" is only for the electronics noise of the circuit. The low-pass filtered noise of the Acromag should be totally negligible above 10 Hz, see here. The fact that there is little difference between the "Idrive=0mA" and "Input Grounded" traces further supports this claim.
  • The diodes in the Pomona box are also unlikely to be the culprit, because with this Pomona box connected to the SR785 and its input terminated with 50ohms, I don't see the comb of spectral lines.

I didn't capture the data, but viewing the high voltage output on an Oscilloscope threw up no red flags - the oscillations which were previously so evident were nowhere to be seen, so I think the capacitor switch did the trick as far as stability is concerned.

There is a large excess between measurement and model out to a few kHz, if this is really what ends up going to the suspension then this circuit is useless. However, I suspect at least part of the problem is due to close proximity to switching power supplies, judging by the comb of ~10 Hz spaced peaks. This is a frequent problem in coil driver noise measurements - previously, the culprit was a switching power supply to the Prologix GPIB box, but now a Linear AC-DC converter is used (besides, disconnecting it had no visible effect). The bench supplies providing power to the board, however, is a switching supply, maybe that is to blame? I think the KEPCO supplies providing +/-250 V are linear. I tried the usual voodoo of twisting the wires used to receive the signal, moving the SR785 away from the circuit board etc, but these measures had no visible effect either. 

Conclusions:

The real requirement of this circuit is that the current noise above 100 Hz be <1pA/rtHz. This measurement suggests a level that is 5x too high. But the problem is likely measurement related. I think we can only make a more informed conclusion after shielding the circuit better and conducting the test in a more electromagnetically quiet environment.

Attachment 1: testSetup.pdf
testSetup.pdf
Attachment 2: HVampNoise_driven.pdf
HVampNoise_driven.pdf
  15571   Tue Sep 15 12:20:36 2020 gautamUpdateElectronicsSR785 repaired

The unit was repaired and returned to the 40m. Now, with a DMM, I measure a DC offset value that is ~1% of the AC signal amplitude. I measured the TF of a simple 1/20 voltage divider and it looks fine. In FFT mode, the high frequency noise floor levels out around 5-7nV/rtHz when the input is terminated in 50 ohms.

I will upload the repair documents to the wiki.

Quote:

The "source" output of the SR785 has a DC offset of -6.66 V. I couldn't make this up.

Attachment 1: dividerTF.pdf
dividerTF.pdf
  15572   Tue Sep 15 17:04:43 2020 gautamUpdateElectronicsDC adaptors delivered

These were delivered to the 40m today and are on Rana's desk

Quote:

I'll order a couple of these (5 ordered for delivery on Wednesday) in case there's a hot demand for the jack / plug combo that this one has. 

  15613   Mon Oct 5 14:01:41 2020 gautamUpdateElectronicsaLIGO demod boards stuffed and delivered

We received 20pcs of stuffed demodulator boards from Screaming Circuits today. Some caveats:

  1. The AP1053 amplifiers weren't stuffed. Note that this part is no longer in standard production, and lead time for a custom run is ~half a year. I recommend stuffing R2 and using a minicircuits amplifier upstream of this board. We have 6 pcs of AP1053 in hand so we can use those for the first AS WFS, but a second WFS will require some workaround.
  2. AD8306ARZ weren't sent to Screaming Circuits. This part is used for the LO and RF signal level detection/monitoring stage, and so aren't crucial to the demodulation operation. @Chub, did we order the correct part now? They are rather pricey so maybe we can just adapt the footprint using some adaptor board?
  3. DQS-10-100 hybrid 90 degree splitters were delivered to us after the lot was sent to Screaming Circuits. We have the pieces in hand, so we can just stuff them as necessary.

I removed 1 from the group to stuff some components that weren't sent to Screaming Circuits and test the functionality on the benchtop, the remaining have been stored in a plastic box for now as shown in Attachment #1. The box has been delivered to Chub who will stuff the remaining 19 boards once I've tested the one piece.

Attachment 1: IMG_8888.JPG
IMG_8888.JPG
  15633   Mon Oct 19 15:38:42 2020 KojiUpdateElectronicsLoan: A file binder "40m wiring diagram"

I'll bring a file binder "40m wiring diagram" to home at the next chance.
There is another one on the shelf in the control room.

(I thought I put it in my bag, but it looks like that I left it somewhere around the fax area)

  15636   Thu Oct 22 11:14:47 2020 gautamUpdateElectronicsHV coil driver packaged into 2U chassis

I packaged the HV coil driver into a 2U chassis, hoping for better shielding from pickup. There is still considerable excess noise in measurement vs model around 100 Hz, see Attachment #1. The projected displacement noise from this noise contribution is shown in Attachment #2 - I've also plotted the contribution from the 4.5kohm (planned value for fast path series resistance) for comparison. Attachment #3 has some photos of the measurement setup so if someone sees some red flags, please let me know.

  • The noise was measured with the output load connected to a 20ohm load resistor, to simulate an OSEM.
  • The input signal was driven with an Acromag, to try and mimic the actual operating conditions as closely as possible (although the fast path input was left unconnected).
  • The KEPCO switching HV power supplies were used to power the unit.

I've run out of ideas to try and make the measurement cleaner - the presence of the rather prominent power line harmonics suggests that this is still not perfect, but what more shielding can we implement? I have to make the measurement on the circuit side of the 25 kohm series resistor, so I am using some Pomona minigrabbers to clip onto the leg of the wirewound resistor (see photos in Attachment #3), so that's not great maybe, but what's the alternative?

So if this is truly the noise of the circuit, then while it's an improvement on the current situaiton, it's unsatisfying that such a simple circuit can't match the design expectations. But how do we want to proceed?

Attachment 1: HVampNoise_driven_chassis.pdf
HVampNoise_driven_chassis.pdf
Attachment 2: HVampNoise_dispUnits.pdf
HVampNoise_dispUnits.pdf
Attachment 3: D1900163_measurementSetup.zip
  15638   Thu Oct 22 13:04:42 2020 ranaUpdateElectronicsHV coil driver packaged into 2U chassis

what is the noise level before the HV stage? i.e. how well is the acromag noise being filtered?

  15639   Thu Oct 22 22:01:53 2020 gautamUpdateElectronicsHV coil driver packaged into 2U chassis

It's not so easy to directly measure this I think, because the filtering is rather aggressive. Attachment #1 shows the measured transfer function (dots) vs the model and Attachment #2 shows the noise. I think this checks out - but I can't definitively rule out some excess noise at 100 Hz from this stage. Because the gain of the HV stage is x31, we'd need a preamp with better than 1nV/rtHz to directly measure the noise I guess. The Acromag noise model in Attachment #2 is based on a measurement I describe here.

Quote:

what is the noise level before the HV stage? i.e. how well is the acromag noise being filtered?

Attachment 1: DACnoiseFilterGain.pdf
DACnoiseFilterGain.pdf
Attachment 2: DACnoiseFilterNoises.pdf
DACnoiseFilterNoises.pdf
  15640   Fri Oct 23 09:03:43 2020 anchalUpdateElectronicsHV coil driver packaged into 2U chassis

Andrew made a battery-powered 0.7 nVrtHz input-referred noise pre-amplifier for gain of 200. That might help you.

Quote:

we'd need a preamp with better than 1nV/rtHz to directly measure the noise I guess.

RXA: 0.7 nV is OK if you're not interested in low noise measurements. Otherwise, we have the transformer coupled pre-amp from SRS which does 0.15 nV/rHz and the Rai Weiss FET amp which has 0.35 nV for high impedance sources.

  15674   Thu Nov 12 14:31:27 2020 gautamUpdateElectronicsSR560s in need of repair/battery replacement

I had to go through five SR560s in the lab yesterday evening to find one that had the expected 4 nV/rtHz input noise and worked on battery power. To confirm that the batteries were charged, I left 4 of them plugged in overnight. Today, I confirmed that the little indicator light on the back is in "Maintain" and not "Charge". However, when I unplug the power cord, they immediately turn off.

One of the units has a large DC output offset voltage even when the input is terminated (though it is not present with the input itself set to "GND" rather than DC/AC). Do we want to send this in for repair? Can we replace the batteries ourselves?

Attachment 1: IMG_8947.jpg
IMG_8947.jpg
  15675   Thu Nov 12 14:55:35 2020 gautamUpdateElectronicsMore systematic noise characterization

Summary:

I now think the excess noise in this circuit could be coming from the KEPCO switching power supply (in fact, the supplies are linear, and specd for a voltage ripple at the level of <0.002% of the output - this is pretty good I think, hard to find much better).

Details:

All component references are w.r.t. the schematic. For this test, I decided to stuff a fresh channel on the board, with new components, just to rule out some funky behavior of the channel I had already stuffed. I decoupled the HV amplifier stage and the Acromag DAC noise filtering stages by leaving R3 open. Then, I shorted the non-inverting input of the PA95 (i.e. TP3) to GND, with a jumper cable. Then I measured the noise at TP5, using the AC coupling pomona box (although in principle, there is no need for this as the DC voltage should be zero, but I opted to use it just in case). The characteristic bump in the spectra at ~100Hz-1kHz was still evident, see the bottom row of Attachment #1. The expected voltage noise in this configuration, according to my SPICE model, is ~10 nV/rtHz, see the analysis note.

As a second test, I decided to measure the voltage noise of the power supply - there isn't a convenient monitor point on the circuit to directly probe the +/- HV supply rails (I didn't want any exposed HV conductors on the PCB) - so I measured the voltage noise at the 3-pin connector supplying power to the 2U chassis (i.e. the circuit itself was disconnected for this measurement, I'm measuring the noise of the supply itself). The output is supposedly differential - so I used the SR785 input "Float" mode, and used the Pomona AC coupling box once again to block the large DC voltage and avoid damage to the SR785. The results are summarized in the top row of Attachment #1.

The shape of the spectra suggests to me that the power supply noise is polluting the output noise - Koji suggested measuring the coherence between the channels, I'll try and do this in a safe way but I'm hesitant to use hacky clips for the High Voltage. The PA95 datasheet says nothing about its PSRR, and seems like the Spice model doesn't include it either. It would seem that a PSRR of <60dB at 100 Hz would explain the excess noise seen in the output. Typically, for other Op-Amps, the PSRR falls off as 1/f. The CMRR (which is distinct from the PSRR) is spec'd at 98 dB at DC, and for other OpAmps, I've seen that the CMRR is typically higher than the PSRR. I'm trying to make a case here that it's not unreasonable if the PA95 has a PSRR <= 60dB @100 Hz.

So what are the possible coupling mechanisms and how can we mitigate it?

  1. Use better power supply - I'm not sure how this spec of 10-50 uV/rtHz from the power supply lines up in the general scheme of things, is this already very good? Or can a linear power supply deliver better performance? Assuming the PSRR at 100 Hz is 60 dB and falls off as 1/f, we'd need a supply that is ~10x quieter at all frequencies if this is indeed the mechanism.
  2. Better grounding? To deliver the bipolar voltage rails, I used two unipolar supplies. The outputs are supposedly floating, so I connected the "-" input of the +300 V supply to the "+" input of the -300 V supply. I think this is the right thing to do, but maybe this is somehow polluting the measurement?
  3. Additional bypass capacitors? I use 0.1 uF, 700V DC ceramic capacitors as bypass capacitors close to the leads of the PA95, as is recommended in the datasheet. Can adding a 10uF capacitor in parallel provide better filtering? I'm not sure if one with compatible footprint and voltage rating is readily available, I'll look around.

What do the analog electronics experts think? I may be completely off the rails and imagining things here.


Update 2130: I measured the coherence between the positive supply rail and the output, under the same conditions (i.e. HV stage isolated, input shorted to ground). See Attachment #2 - the coherence does mirror the "bump" seen in the output voltage noise - but the coherence is. only 0.1,  even with 100 averages, suggesting the coupling is not directly linear - anyways, I think it's worth it to try adding some extra decoupling, I'm sourcing the HV 10uF capacitors now.

Attachment 1: powerSupplyNoise.pdf
powerSupplyNoise.pdf
Attachment 2: coherence.pdf
coherence.pdf
  15676   Thu Nov 12 15:40:42 2020 KojiUpdateElectronicsMore systematic noise characterization

Yes. The datasheet has a recommendation circuit with 10uF caps. Companies are careful to show reproducible, reliably functional circuit examples on datasheets. So, if the caps are there you should try to replicate the design.

Quote:

Additional bypass capacitors? I use 0.1 uF, 700V DC ceramic capacitors as bypass capacitors close to the leads of the PA95, as is recommended in the datasheet. Can adding a 10uF capacitor in parallel provide better filtering? I'm not sure if one with compatible footprint and voltage rating is readily available, I'll look around.

  15677   Mon Nov 16 00:02:34 2020 ranaUpdateElectronicsMore systematic noise characterization

true. also try to choose a cap with a goow high frequency response. In the Electronics Noise book by Ott there's some graph about this. I bet you good do a Bing search and also find something more modern. Basically we want to make sure that the self resonance is not happening at low frequencies. Might be tought to find one with a good HF response, a high voltage rating, and > 1uF.

Quote:

Yes. The datasheet has a recommendation circuit with 10uF caps. Companies are careful to show reproducible, reliably functional circuit examples on datasheets. So, if the caps are there you should try to replicate the design.

Quote:

Additional bypass capacitors? I use 0.1 uF, 700V DC ceramic capacitors as bypass capacitors close to the leads of the PA95, as is recommended in the datasheet. Can adding a 10uF capacitor in parallel provide better filtering? I'm not sure if one with compatible footprint and voltage rating is readily available, I'll look around.

  15679   Tue Nov 17 00:26:32 2020 ranaUpdateElectronicsSR560s in need of repair/battery replacement

yes, both problems can be fixed. Usually we just order some spare lead-acid batteries from SRS (Steve may have some spare ones somewhere). The DC offset often comes from a busted FET input. I bought 50 of those at one point - they're obsolete. Its also possible to replace the input stage with any old FET pair.

I'll handle the one with the offset if you leave it on my desk.

  15699   Thu Dec 3 10:46:39 2020 gautamUpdateElectronicsDC power strip requirements

Since we will have several new 1U / 2U aLIGO style electronics chassis installed in the racks, it is desirable to have a more compact power distribution solution than the fusable terminal blocks we use currently. 

  • The power strips come in 2 varieties, 18 V and 24 V. The difference is in the Dsub connector that is used - the 18 V variant has 3 pins / 3 sockets, while the 24V version uses a hybrid of 2 pins / 1 socket (and the mirror on the mating connector).
  • Each strip can accommodate 24 individual chassis. It is unlikely that we will have >24 chassis in any collection of racks, so per area (e.g. EX/EY/IOO/SUS), one each of the 18V and 24V strips should be sufficient. We can even migrate our Acromag chassis to be powered via these strips.
  • Details about the power strip may be found here.

I did a quick walkaround of the lab and the electronics rack today. I estimate that we will need 5 units of the 24 V and 5 units of the 18 V power strips. Each end will need 1 each of 18 V and 24 V strips. The 1Y1/1Y2/1Y3 (LSC/OMC/BHD sus) area will be served by 1 each 18 V and 24 V. The 1X1/1X2 (IOO) area will be served by 1 each 18 V and 24 V. The 1X5/1X6 (SUS Shadow sensor / Coil driver) area will be served by 1 each of 18 V and 24 V.  So I think we should get 7 pcs of each to have 2 spares.

Most of the chassis which will be installed in large numbers (AA, AI, whitening) supports 24V DC input. A few units, like the WFS interface head, OMC driver, OMC QPD interface, require 18V. It is not so clear what the input voltage for the Satellite box and Coil Drivers should be. For the former, an unregulated tap-off of the supply voltage is used to power the LT1021 reference and a transistor that is used to generate the LED drive current for the OSEMs. For the latter, the OPA544 high current opamp used to drive the coil current has its supply rails powered by again, an unregulated tap-off of the supply voltage. Doesn't seem like a great idea to drive any ICs with the unregulated switching supply voltage from a noise point of view, particularly given the recent experience with the HV coil driver testing and the PSRR, but I think it's a bit late in the game to do anything about this. The datasheet specs ~50 dB of PSRR on the negative rail, but we have a couple of decoupling caps close to the IC and this IC is itself in a feedback loop with the low noise AD8671 IC so maybe this won't be much of an issue.

For the purposes of this discussion, I think both Satellite Amp and Coil Driver chassis can be driven with +/- 24 V DC.


On a side note - after the upgrade will the "Satellite Amplifiers" be in the racks, and not close to the flange as they currently are? Or are we gonna have some mini racks next to the chambers? Not sure what the config is at the sites, and if the circuits are designed to drive long cables.

  15700   Thu Dec 3 11:02:35 2020 ranaUpdateElectronicsElectrical LO signal for AS WFS

looks good to me.

The thing I usually look for is how much the downstream system (mixers, etc) can perturb the main oscillator. i.e. we don't want mixer in one chain to reflect back and disturb the EOM chain. But since our demods have amplifiers on the LO side we're pretty immune to that.

  15709   Fri Dec 4 19:23:40 2020 KojiUpdateElectronicsAA/AI board testing ongoing

I have the setup built for the AA/AI board testing around the PD testing area. Please let me leave it like that for a week or so.

12/4 TF Tested 5 PCBs
12/6 TF Tested 19 PCBs (12min/PCB) - found 1 failure (S2001479 CH1) -> Fixed 12/11
12/8 TF Tested 16 PCBs (12min/PCB)
       PSD Tested 4 PCBs (11min/PCB)
12/11 TF Tested 10 PCBs + 1 fixed channel (All channels checked)
       PSD Tested 10 PCBs (11min/PCB)
12/14 PSD Tested 4 PCBs (6.5min/PCB) fixed noise issue of 2 ch, TF issue of 1 ch
12/15 PSD Tested 32 PCBs (6.5min/PCB) fixed noise issue of 1ch
Temp dependence measurement
Crosstalk measurement
 

 

  15735   Tue Dec 15 12:38:41 2020 gautamUpdateElectronicsDC power strip

I installed a DC power strip (24 V variant, 12 outlets available) on the NW upright of the 1X1 rack. This is for the AS WFS. Seems to work, all outlets get +/- 24 V DC.

The FSS_RMTEMP channel is very noisy after this work. I'll look into it, but probably some Acromag grounding issue.

In the afternoon, Jordan and I also laid out 4x SMA LMR240 cables and 1x DB15 M/F cable from 1X2 to the NE corner of the AP table via the overhead cable trays.

  15741   Sat Dec 19 20:24:25 2020 gautamUpdateElectronicsWFS hardware install

I installed 4 chassis in the rack 1X2 (characterization on the E-bench was deemed satisfactory, I will upload the analysis later). I ran out of hardware to make power cables so only 2 of them are powered right now (1 32ch AA chassis and 1 WFS head interface). The current limit on the +24V Sorensens was raised to allow for similar margin to the limit with the increased current draw.

Remaining work:

  1. Make 2 more power cables for ISC whitening chassis and quad demod chassis.
  2. Make a 2x 4pin LEMO-->DB9 cable to digitize the FSS and PMC diagnostic channels with the new AA chassis. If RnD cables has a very short turnaround time, might be worth it to give this to them as well.
  3. Connect ADC1 on c1ioo machine to new AA chassis (transfer SCSI cable from existing AA unit to the new one). This will necessarily involve some model changes as well.
  4. Make a short cable to connect 55 MHz output from RFsource box to the LO input on the quad demod chassis.
  5. Install the WFS head on the AS table at a suitable location. Probably will need a focusing lens as well. 
  6. Connect WFS head to the signal processing electronics (the cables were already laid out by Jordan and I).
  7. Make the necessary CDS model changes (WFS filters, matrices, servos etc). I personally don't see the need for a new model but if anyone feels strongly about separating the IMC WFS and AS WFS we can set up another model.
  8. Commission the system.

While I definitely bumped various cables, I don't seem to have done any lasting damage to the CDS system (the RFM errors remain of course).

  15773   Wed Jan 20 10:13:06 2021 gautamUpdateElectronicsHV Power supply bypassing

Summary:

Installing 10uF bypass capacitors on the High Voltage power supply line for the HV coil driver circuit doesn't improve the noise. The excess bump around a few hundred Hz is still present. How do we want to proceed? 

Details

  • The setup is schematically shown in Attachment #1.
  • Physically, the capacitors were packaged into a box, see Attachment #2.
  • This box is installed between the HVPS and the 2U chassis in which the circuit is housed, see Attachment #3.
  • I measured the noise, (using the same setup as shown here to avoid exposing the SR785 input to any high voltage), for a variety of drive currents. To make a direct comparison, I took two sets of measurements today, one with the decoupling box installed and one without.
  • The results are shown in Attachment #4. You can see there is barely any difference between the two cases. I've also plotted the expected noise per a model, and the measured Johnson noise of one of the 25kohm resistors being used (Ohmite, wirewound). I just stuck the two legs of the resistor into the SR785 and measured the differential voltage noise. There is a slight excess in the measured Johnson noise compared to what we would expect from the Fluctuation Dissipation theorem, not sure if this is something to be worried about or if it's just some measurement artefact.

Discussion:

So what do we do about this circuit? For the production version, I can make room on the PCB to install two 10uF film capacitors on the board itself, though that's unlikely to help. I think we've established that 

  1. The excess noise is not due to the Acromag or the input Acromag noise filtering stage of the circuit, since the excess is present even when the input to the HV stage is isolated and shorted to ground.
  2. There was some evidence of coherence between the supply rails and the output of the HV stage (with input isolated and shorted to ground). The coherence had the "right shape" to explain the excess noise, but the maximum value was only ~0.1 (could have been because I was not measuring directly at the PA95's supply rail pins due to space constraints).
  3. The impedance of 10uF at 100Hz is ~150 ohms. idk what the impedance of the supply pins of the PA95 are at this frequency (this will determine the coupling of ripples in the HVPS output to the PA95 itself.

Do we have any better bipolar HV supply that I can use to see if that makes any difference? I don't want to use the WFS supplies as it's not very convenient for testing.


Not really related directly to this work but since we have been talking about current requirements, I attach the output of the current determining script as Attachment #5. For the most part, having 220ohm resistances on the new HAM-A coil driver boards will lead to ~half the DAC range being eaten up for the slow alignment bias. For things like MC1/MC3, this is fine. But for PRM/SRM/BS, we may need to use 100ohms. Chub has ordered all manner of resistances so we should have plenty of choices to pick from.

Attachment 1: bypassCaps.pdf
bypassCaps.pdf
Attachment 2: IMG_9079.jpg
IMG_9079.jpg
Attachment 3: IMG_9078.jpg
IMG_9078.jpg
Attachment 4: HVampNoise_driven_chassis.pdf
HVampNoise_driven_chassis.pdf
Attachment 5: printCoilCurrents.pdf
printCoilCurrents.pdf
  15786   Mon Feb 1 12:30:21 2021 gautamUpdateElectronicsMore careful characterization

Summary:

  1. Swapping out the KEPCO HV supplies (linear) I was using for a pair of HP6209s I borrowed from Rich has improved the noise performance somewhat.
  2. However, there is still an excess relative to the model. I confirmed that this excess originates from the PA95 part of the circuit (see details).
  3. The bypass capacitors don't seem to have any effect on the measured ripple from these HP6209s. Maybe they're internally fitted with some 10uF or similar bypass caps?
  4. The production version of this board, with several improvements (after discussions with Koji and Rich), are on the DCC. They're being fabbed right now and will arrive in ~1 week for more bench testing. 

Power supply bypassing [updated 10pm]:

As mentioned earlier in this thread, I prepared a box with two 10uF, 1kV rated capacitors to bypass the high-voltage rails (see inset in the plot), to see if that improves the performance. However, in measuring the voltage ripple directly with the SR785 (no load connected), I don't see any significant difference whether the decoupling caps are connected or not, see Attachment #1. For this, and all other HV measurements made, I used this box to protect the SR785. One hypothesis is that this box itself is somehow introducting the excess noise, maybe because of leakage currents of the diode pair going into the 1Mohm SR785 input impedance, but I can't find any spec for this, and anyway, these diodes should be at ground potential once the transient has settled and the DC blocking capacitor has charged to its final value.

Note that the 10uF caps have an ESR of 7.2 mOhms. The HP6209 has a source impedance "<20mOhm" when operated as a CV source, per the datasheet. So perhaps this isn't so surprising? The same datasheet suggests the source impedance is 500 mOhms from 1kHz to 100 kHz, so we should see some improvement there, but I only measured out to 2 kHz, and I didn't take much effort to reduce these crazy peaks so maybe they are polluting the measurement out there. There must also be some continuous change of impedance, it cannot be <20 mOhm until 1 kHz and then suddenly increase to 500 mOhms. Anyways, for this particular circuit, the nosie DC-1kHz is what is important so I don't see a need to beat this horse more. 

Simplified circuit testing:

I decided to see if I can recover the spec'd voltage noise curve from the PA95 datasheet. For this, I configured the PA95 as a simple G=31 non-inverting amplifier (by not stuffing the 15 uF capacitor in the feedback path). Then, with the input grounded, I measured the output voltage noise on the circuit side of the 25kohm resistor (see inset in Attachment #2). To be consistent, I used the DC blocking box for this measurement as well, even though the output of the PA95 under these test conditions is 0V. Once again, there is considerable excess around ~100 Hz relative to a SPICE model. On the basis of this test, I think it is fair to say that the problem is with the PA95 itself. As far as I can tell, I am doing everything by the book, in terms of having gain > 10, using a sufficiently large compensaiton cap, HV rail decoupling etc etc. Note that the PA95 is a FET input opamp, so the effects of input current noise should be negligible. The datasheet doesn't provide the frequency dependence, but if this is just shot noise of the 1200 pA input bias current (for 300 V rails, per the spec), this is totally negligible, as confirmed by LTspice.

In the spirit of going step-by-step, I then added the feedback capacitor, and still, measured noise in excess of what I would expect from my model + SR785 measurement noise.

Integrated circuit testing:

After the above simplified test, I stuffed a full channel as designed, and tested the noise for various drive currents. To best simulate the operating conditions, an Acromag XT1541 was used to set the DC voltage that determines the drive current through the 25 kohm resistor. The measurements were made on the circuit side of this resistor (I connected a 20ohm resistor to ground to simulate the OSEM). As shown in Attachment #3, the noise with these HP6209 supplies is significantly better than what I saw with the KEPCO supplies, lending further credence to the hypothesis that insufficient PSRR is the root of the problem here. I've added subplots in a few different units - to be honest, I think that reaching this level of measured displacement noise at the 40m at 100 Hz would already be pretty impressive.

So what's next?

The main design change is that a passive R-C-R (4k-3uF-20k) replaces the single 25kohm resistor at the output of the PA95. 

  • This allows similar current drive range.
  • But adds an LPF to filter out the observerd excess noise at 100 Hz. 

Let's see if this fixes the issue. Not that I've also added a pair of input protection diodes to the input of the PA95 in the new design. The idea is that this would protect the (expensive) PA95 IC from, for example, the unit being powered with the +/- 18V rail but not the +/- 300 V rail. As I type this, however, I wonder if the leakage current noise of these diodes would be a problem. Once again, the datasheet doesn't provide any frequency dependence, but if it's just the shot noise of the 1nA expected when the diodes are not reverse biased (which is the case when the PA95 is operating normally since both inputs are at nearly the same potential), the level is ~20 fA/rtHz, comparable to the input current noise of the PA95, so not expected to be an issue. In the worst case, the PCB layout allows for this component to just be omitted. 

Attachment 1: HVPS.pdf
HVPS.pdf
Attachment 2: HV_testckt.pdf
HV_testckt.pdf
Attachment 3: totalNoise.pdf
totalNoise.pdf
  15798   Wed Feb 10 14:14:58 2021 gautamUpdateElectronicsCustom cables received

We received the custom cables to test the new suspension electronics. They are under my desk. So we are ready.

This batch was a small one - the company says that they can make molded cables if we have a minimum order, something to consider I gues.s.


Update 1900 11 Feb: I verified that the pin outs of the cables are as we intended (for one set of each type of cable). Because this was a small order, the connectors have metal shells, and so for cable #2 (sat box to flange), the two shells are shorted to each other. I can't verify if the shield is isolated from the shell on J5 without cutting open the cable. One thing that occurred to me is that we should give pins 5,8,11 on J4 and 16,20,24 on J5 (respectively) unique identifiers. They should only be shorted to GND on the circuit board itself. To be fixed for the next iteration. I uploaded some photos here.

I was unable to measure the capacitance of the cable using the LCR meter, and didn't opt to try any other method.

Attachment 1: satWiring.pdf
satWiring.pdf
  15802   Wed Feb 10 21:14:03 2021 gautamUpdateElectronicsProduction version of the HV coil driver tested

Summary:

I did what I consider to be a comprehensive set of tests on the production version of the high voltage coil driver circuit. I think the performance is now satisfactory and the circuit is ready for the production build. Barring objections from anyone, I will ask Chub to place an order for components to stuff the 4 necessary units + 1 spare on Friday, 12 Feb (so that people have a full day to comment). A big thanks to Chub and the folks at JLCPCB for dealing with my incessant order requests and patiently supporting this build and letting me turn this around in 10 days - hopefully this is the end of this particular saga.

Schematic is here. All references to component designations are for v4 of the schematic.

Important design changes:

  1. All I/O to this board will be via D9 connectors. This will allow bypassing the high voltage stage in future suspensions while retaining the same cable config in the suspension drive, if that is desired. Some re-arrangement of the grouping of coils was also done for consistency with the planned upgrade.
  2. Differential receiving for the input signal from the Acromag. The nominal quad opamp is LT1125 but if we find noise issues (which I didn't), the OP497 has compatible PCB footprint.
  3. Added input protection dual diode D6 to protect the PA95 as recommended in the datasheet. This should protect the IC if (for example) the HV line isn't plugged in but the Acromag input is non-zero.
  4. Increased the feedback resistance from 30kohms to 12kohms. R16 through R21 are now 20k, while the old design had 5k. The purpose is to reduce the current demand in the feedback path, hopefully this opens up the number of DCPS we can use. To keep the overall gain of 31, the resistor R15 was upped from 1kohms to 4kohms.
  5. Feedback capacitance reduced from 15 uF to 3 uF. This was largely for space saving / ease of layout on the PCB. The resulting corner frequency is increased slightly from 0.35 Hz to 0.45 Hz but this doesn't have any imapct on the performance of the circuit at frequencies of interest (1/2/pi/R/C had R=30k, C=15uF, now R=120k, C=3uF).
  6. Added an R-C-R network at the output of the PA95, before the fast actuation signal is summed and sent to the OSEM coil.
    • This is probably the most important change, noise-performance wise.
    • The purpose of the network is to passively filter out the excess noise we saw at ~100 Hz (the corner from the 4kohm resistor + 3uF cap is at ~13 Hz, so factor of 10 filtering at 100 Hz, which was deemed sufficient, see earlier elogs in the thread). 
    • The Johnson noise contribution of the 20 kohm resistor is slightly higher than the original design which had a 25 kohm series resistor (but no R-C-R passive filter at the output of the PA95). But once again, this was deemed to have negligible effect on the performance at frequencies of interest to us.
    • The total current driving capability of the circuit is almost unchanged since the 20kohm + 4kohm nearly equals the old 25kohm resistance.
  7. Made the Vmon paths for monitoring the HV output of the PA95 differential sending, seems like a good practise to follow for all new designs.
  8. Added on-board bypass capacitors (2 x 10uF WIMA film caps) for cleaning up the HV supply noise.

Tests:

A series of tests were done. Note that only 1 channel was stuffed (I am out of PA95s), and the HP power supplies borrowed from Rich were used for the HV rails. For the +/-18V, a regular bench-top unit was used.

  1. Low voltage stage tests
    • TF of the differential receiving stage was measured and the overall unity gain and corner at 24kHz were verified, see Attachment #1.
    • With the input of the circuit grounded, I measured the noise of the circuit at various points (see legends on Attachment #2). I didn't bother to do a detailed verification against a SPICE model as the levels seemed roughly what is expected.
  2. Overall noise performance with HV stage enabled
    • For a range of drive currents, generated by applying the appropriate voltage using an Acromag XT1541 DAC module to the J1 connector, I measured the voltage on the circuit side of the 20 kohm resistor (by clipping onto the resistor leg. Note that the path to ground for the current was provided by connecting a 20 ohm resistor between pins 1 and 6 on J3a, and then grounding pin 6.
    • Results are shown in Attachment #3
    • For the drive currents at the edge of the range of operation, there is a small excess relative to lower drive currents. My best hypothesis for why this is happening is that the HV rail is too close to the requested output voltage (the HP units are rated for 320V, which is borderline if we want 300V at the output of the PA95). In any case, the R-C-R passive filter means that above ~60 Hz, there is excellent agreement between model and measurement.
  3. Time domain tests
    • Used a function generator. to drive a 50 mHz, 3Vpp sine wave to the "Bias Input" (=J1), and monitored (i) pickoff of drive signal, (ii) High voltage output at the circuit side of the 20kohm resistor, and (iii) the Vmon output (=pins 1/6 on J4), all on a 100 MHz Tektronix scope.
    • Results shown in Attachment #4. Once again, I see no red flags.
    • While I had the unit hooked up to the scope, I also checked the time domain signal with the scope set to 100 ns/div time base. I saw no evidence of any oscillatory features, either when no input signal was applied, or when a DC signal was provided (in which case the scope was set to AC coupling). 👍 
  4. Checked that the protection diodes at various locations in the circuit work.
  5. Checked the pin-mapping on all 6 D9 connectors is consistent with the top level diagram in the schematic.

PCB remarks:

As I was stuffing the board, I noticed a few improvements that can be made. Just noting these here for documentation purposes - these changes are mostly aesthetic and I personally see no need to order another set of PCBs.

  1. In some places, the silkscreen designators don't have the correct "orientation" relative to the component they are designating. I didn't find any serious ambiguity in terms of being misled to stuff the wrong component onto the wrong pads, but in the spirit of doing a professional job...
  2. The current limiting resistors on the +/-430V LEDs (R37/R38) have footprints for leaded components rather than SMT (which is what the +/-15V LEDs have).
  3. R45 and R46, the current limiting resistors for the rear panel power indicator LEDs, have 0805 footprint rather than 1206.
  4. When I drew up the PCB, R23, the 4kohm resistor in the R-C-R network, was set up as a 1W resistor. Let's say there can be 15 mA flowing through this resistor - the power dissipated is 15e-3 ^2 * 4e3 is 0.9W, which is uncomfortably close to the limit. For all the tests above, I used a 3W resistor, and didn't find any serious noise issues. The drilled holes are a little tight for this higher power rated resistor, but I don't think this is a showstopper.

Communications with Apex:

I've been talking to support at Apex, and pointed out that I couldn't match the SPICE model performance even for a simple non-inverting amplifier with the PA95. The feedback I got from them was that 

  1. They don't optimize the SPICE models for input noise and so it was a nice coincidence that model and measurement are somewhat close (but not exactly).
  2. They recommend the PA194, which is actually advertised as "low-noise". The PA95 is apparently not a "low-noise" part, with its 2uVrms input noise. 

Whiel the PA194 is compatible with our voltage and current requirements for this application, it is ~3x the cost, and seems like the R-C-R output filter allows us to realize the goal of 1pA/rtHz, so I'm inclined to stick with the PA95.

Production assembly:

I'd prefer to get as much of the board stuffed by Screaming Circuits as possible. It took me ~3 hours to stuff 1 channel + the power supply parts, standoffs etc. So I estimate it'll take me ~6 hours to stuff the entire board. So not the end of the world if we have to do it in-house.

Attachment 1: inputDiffRecTF.pdf
inputDiffRecTF.pdf
Attachment 2: LVnoises.pdf
LVnoises.pdf
Attachment 3: totalNoise.pdf
totalNoise.pdf
Attachment 4: timeDomainTests.pdf
timeDomainTests.pdf
  15815   Thu Feb 18 03:20:09 2021 KojiSummaryElectronicsCurrent Rack Map

For your planning:

Attachment 1: rack_plan.pdf
rack_plan.pdf
  15820   Thu Feb 18 20:24:48 2021 KojiSummaryElectronicsA bunch of electronics received

Todd provided us a bunch of electronics. I went to Downs to pick them up this afternoon and checked the contents in the box. Basically, the boxes are pretty comprehensive to produce the following chassis

  • 8 HAM-A coil driver chassis
  • 7 16bit Anti-Aliasing chassis
  • 4 18bit Anti-Imaging chassis
  • 5 16bit Anti-Imaging chassis

Some panels are missing (we cannibalized them for the WFS electronics). Otherwise, it seems that we will be able to assemble these chassis listed.
They have placed inside the lab as seen in the attached photo.


HAM-A COIL DRIVER (Req Qty 28+8)

- 8 Chassis
- 8 Front Panels
- 8 Rear Panels
- 8 HAM-A Driver PCBs
- 8 D1000217 DC Power board
- 8 D1000217 DC Power board

16bit AA (Req Qty 7)
- 7 CHASSIS
- 6 7 Front Panels (1 missing -> [Ed 2/22/2021] Asked Chub to order -> Received on 3/5/2021)
- 7 Rear Panels
- 28 AA/AI board S2001472-486, 499-511
- 7 D070100 ADC AA I/F
- 7 D1000217 DC Power board

18bit AI (Req Qty 4)
- 4 CHASSIS
- 4 Front Panels
- 4 Rear Panels
- 8 AA/AI board S2001463-67, 90-92
- 4 D1000551 18bit DAC AI I/F
- 4 D1000217 DC Power board
- bunch of excess components

16bit AI (Req Qty 5)
- 5 CHASSIS
- 4 5 Front Panels (D1101522) (1 missing -> [Ed 2/22/2021] Asked Chub to order -> Received on 3/5/2021)
- 3 5 Rear Panels (D0902784) (2 missing -> [Ed 2/22/2021] Asked Chub to order -> Received on 3/5/2021)
- 10 AA/AI board S2001468-71, 93-98
- 5 D1000217 DC Power board
- 5 D070101 DAC AI I/F

Internal Wiring Kit

[Ed 2/22/2021]
Asked Chub to order:
- Qty 12 1U Hamilton Chassis
- Qty 5 x Front/Rear Panels/Internal PCBs for D1002593 BIO I/F (The parts and connectors to be ordered separately)

  -> Front/Rear Panels received (3/5/2021)
  -> PCBs (unpopulated) received (3/5/2021)
  -> Components ordered by KA (3/7/2021)

Attachment 1: IMG_0416.jpeg
IMG_0416.jpeg
  15828   Sat Feb 20 10:01:48 2021 gautamSummaryElectronicsA bunch of electronics received

Will we also be receiving the additional 34 Satellite Amplifier PCBs?

  15830   Sat Feb 20 16:46:17 2021 KojiSummaryElectronicsA bunch of electronics received

We received currently available sets. We are supposed to receive more coil drivers and sat amps, etc. But they are not ready yet.

 

  15846   Fri Feb 26 16:31:02 2021 gautamUpdateElectronicsProduction version of the HV coil driver tested with KEPCO HV supplies

Koji asked me to test the production version of the coil driver with the KEPCO HV supplies. See Attachment #1 for the results. For comparison, I've added a single trace from the measurements made with the HP supplies. I continue to see excess noise with the KEPCO supplies. Note that in the production version of the board that was tested, there are a pair of 10uF bypass capacitors on the board for the HV supply lines. It is possible that one or both KEPCO supplies are damaged - one was from the ASY setup and one I found in the little rack next to 1X2. The test conditions were identical to that with the HP supplies (as best as I could make it so).

Attachment 1: totalNoise_KEPCO.pdf
totalNoise_KEPCO.pdf
ELOG V3.1.3-