I wonder why POY11 has the dark noise level of 90nV/rtHz that is 5 times larger than that of POX (18nV/rtHz)
even though the Q are the same (~15) and the transimpedance is better (3.9k instead of 2k).
What cause this high noise level?
What is the expected dark noise level?
[Koji and Kevin]
I measured the shot noise of POY and fit the data to determine the RF transimpedance at 11 MHz and the dark current. The transimpedance is (3.860 +- 0.006) kΩ. I realize that there are not many data points past the dark current but I did not want to take any further data because the light bulb was getting pretty bright. If this is a problem, I can try to redo the measurement using a lens to try to focus more of the light from the bulb onto the photodiode.
I also measured the spectrum and recorded a time series of the RF signal with the light to the photodiode blocked. These measurements do not show any large oscillations like the ones found for POX.
The plots of the measurements are on the wiki at http://lhocds.ligo-wa.caltech.edu:8000/40m/Electronics/POY.
This diagram shows the setup of the analog Mixer-Frequency Discriminator (MFD).
The idea is similar to the one of the Schnupp Asymmetry for our Michelson interferometers. The signal from the PD (or any signal source for which you want to know the frequency) is split into two legs; one leg is much longer than the other. The two legs are recombined at a mixer/demodulator. The demodulator output varies sinusoidally with the phase difference of two legs, the same as when we try to measure the phase noise of an oscillator, for example. This is the same concept as the digital frequency discriminator that Aidan and Joe put into the GFD FE system recently.
With a ~1m cable length asymmetry, we get 180 deg of phase shift for a ~100 MHz signal (recall that the speed of light in most of our cables is ~2 x 10^8 m/s). The mixer gives a linear output at 50 MHz (and 150 MHz, 250 MHz, etc.).
This single mixer based setup is fine for most everything we do. In order to get even more resolution, one can just use 2 mixers by splitting the signal with a 4-way instead of 2-way mixer. One setup can have a 0.5-1 m asymmetry to have a large range. The other can have a ~10-30m asymmetry to get a comb of linear readouts.
Typically, we will have some kind of weak signal at the photodiode and will use a 20 or 40 dB gain RF amp to get the signal into the mixer. In this case, the mixer output noise will be at the level of tens of nV/rHz. Any usual low noise audio amplifier (SR560 variety) will be enough to read out the signal.
Why the 50 Ohm terminator? If you look at the specs of the BLP-1.9 filter from Mini-Circuits (its the same for almost all of their LP filters) you see that there's ~90 dB of attenuation above ~30 MHz (where our signals 2*f product will show up). If we use an RF input signal of ~0 dBm, this means that we get a high frequency product of -95 dBm (~10 uVrms) which is OK. But the return loss is 0 dB above 5 MHz - this means that all of the high frequency content is reflected back into the mixer! The 50 Ohm terminator is there to absorb the RF signals coming out of the mixer so as to prevent them from going back into the mixer and mixing with the RF/LO signals. The 50 Ohm terminator does attenuate the DC/audio frequency signals we get out of the mixer by a factor of two, but that's OK since we are not limited by the mixer's thermal noise.
To checkout the noise, we used a 6m RG-58 cable in one leg. We used the DS345 signal generator for the source. We adjusted the frequency to (~21 MHz) give a ~zero mean signal at the demod output. The 6m cable makes the demod output's peak-peak swing correspond to ~16 MHz. We then used an SR560, DC coupled, G=1000, low-noise, 2pole low pass at 1 kHz, to get the signal into the ADC.
The attached plot shows the noise. We have caibrated the digital gain in the channel to make the output into units of Hz. The high frequency noise floor is ~0.3 Hz/rHz and the 1/f knee is at 10 Hz. This setup is already good enough for all of the green locking work at the 40m. In order to make this useful for the reference cavity work or the gyro, we will have to use a longer cable and a lower noise audio amplifier.
As can be seen from the plot, the ADC noise is below the measured noise. The noise of the SR560 with the input terminated is shown in grey - the measured noise of the MFD is very close to this. In order to improve the performance, the next step should be to use a longer cable. There's clearly going to be some trade-off between the temperature dependent effects which come with long cables (dphi/dT gets bigger) and trying to use a high gain ~1 nV/rHz amplfier at the mixer output.
Temperature Drift of the long cable:
This 24-hour minute-trend shows the frequency wander as well as the room temperature. This is not proof of a temperature dependence, but if it is then we get ~3 kHz/deg for the sensitivity. If this is actually the cable and not the amplifier, then we'll have to hunt for a lower tempco cable and put it in a box to isolate it.
I swapped over to a 3x longer cable (old 65 ft. Pasternak cable from ancient 40m days). The old one was 6m, the new one is 18.2 m. It was already coiled up so I put it into a tupperware box to shield it somewhat from the HVAC wind.
The noise went down nearly proportional to the length (after I recalibrated the DAQ channel for the ~3x higher phase->voltage gain). With this length, the peak-peak mixer range is 5.5 MHz, so still enough to go an FSR here.
I give credit to the low frequency improvement entirely to Tupperware for their excellent containers. The current noise limit is most likely the SR560.
This is with reference to Kevin and Jenne's elogs # 3890, 4034 and 4048 .
While the electronics are working okay, there is no DC signal from the photodiode.
Since the solderings and tracks on the PCB were fine I took a close look at the exposed front face of the photodiode.
As we can see, one of the thin wires on the top surface of the photodiode is broken. We can see some wipe marks closer to the lower left edge..
Something seems to have brushed across the exposed face of the photodiode and dislodged the wire.
The new photodiode still has its protective can intact. Do we need to remove the can and expose the photodiode before istallation?
A new photodiode ( Perkin and Elmer Model no. C30642GH Sl No.1526) has been installed in the place of the old photodiode. The datasheet of this model is attached.
The 68pF capacitor which was present in parallel with the photodiode has been removed. Here is a picture of the PCB ( in all its gory detail!) and the photodiode after replacement.
I also checked to see if we have a DC output from the new PD. With 375mW of 1064nm light incident we have 15mV of output. Which matches well with the typical Reponsivity of 0.8V/A reported in the datasheet and our REFL11 ckt . The schematic of the ckt is also attached here for easy reference. The various factors are
V_dc = 0.375 mW x 0.8 V/A x 10 Ohm x 5 = 15mV
The last factor is the gain of the last stage on the DC route.
When I reassembled the box I noticed that there is problem with the SMA connectors popping out of the box. The holes seem misplaced so I enlarged the holes to remove this concern.
375 mW is way too much light. We must never put more than 100 mW on any of these diodes. We don't want to blow up more diodes like we did with the WFS. The InGaAs diodes often show an excess dark noise before they finally let go and completely fail. This one may show excess during the shot noise testing.
We should ensure that the beam paths are engineered so that none of these new detectors ever sees such high light levels.
The DC path should be made to let us see a 10V from the differential EPICS readout when there is 100 mA of photocurrent (i.e. an effective 100 Ohms transimpedance):
0.1 A * 10 V/A * 5 V/V * 2V/V
The last factor of 2 is from the single to differential conversion.
If we really only get 15 mV from 375 mW, then this diode or the circuit is broken.
Suresh is saying 375mW and 0.375mW. Let's wait for his update of the actual power.
Also he is not using EPICS, there may be the factor of two missing for now.
It is 0.375 mW as in the calculation. The total diode output is just 1mW and it is divided with a 50/50 beam splitter... There are a couple of lenses along the way which may account for the ~12% loss.
I used a handheld multimeter to measure the output.
We hooked up the VCO Driver output to the MFD. We adjusted the levels with attenuators to match up to the Level 7 mixer that's being used.
The mixer the input to the SR560 is going in to the XARM_COARSE_OUT channel and the SR560 (AC coupled, Low Noise, G=1000, LP@1kHz) 600 Ohm output goes into XARM_FINE_OUT.
We calibrated these channels by putting in a 10 mVpp sine wave at 0.22 Hz into the Wideband Input of the VCO Driver box (which has been calibrated to have 1.75 MHz/V for f < 1.6 Hz). This should correspond to 17.5 kHz_pp.
To increase the sensitivity, we also added a 140 ft. BNC cable to the setup. We also added some extra short cable to make the overall phase shift be ~90 deg and zero out the mixer output.
I used the time series data in DTT to then calibrate the channels by changing the GAIN field in their filter modules. So now the DAQ channels are both calibrated as 1 count/Hz.
There were several parts in this box which did not have shunting capacitors across their input power lines. Only the four RF amps (ZHL-2) had them.
I soldered two capacitors (100 microF electrolytic and 150pF dipped mica) across the power supply lines of each of the following units: 11MHz oscillator, 29.5 MHz oscillator, Wenzel 5x frequency multiplier and the 12x RF amplifier (ZHL-1HAD).
It was quite difficult to reach the power inputs of these units as some of them were very close to the inner walls of the box. To access them I undid the front panel and found that there were several very taut RF cables which prevented me from moving the front panel even a little.
I had to undo some of the RF cables and swap them around till I found a solution in which all of them had some slack. At the end I checked to make sure that the wiring is in accordance with the schematic present here.
The Distribution box is several steps nearer to completion.
1) Soldered capacitors and DC power lines for four units of the distribution box.
2) mounted all the components in their respective places.
3) Tomorrow we prepare the RF cables and that is the last step of the mechanical assembly.
4) we plan to test both the generator and distributon parts together.
Kevin took a transfer function of the newly assembled PD and noticed that the frequency has shifted to 14.99 freom 11. MHz.
We needed to find the current RLC combination. So we removed the ferrite core from L5 rendiring it to its aircore value of 0.96/muH. We then used this to find the Capacitance of the PD (117pF)
We used this value to compute the inductance required to achieve 11.065MHz which turned out to be 1.75microH.
This was not reachable with the current L5 which is of the type 143-20J12L (nominal H=1.4 micro Henry).
We therefore changed the inductor to SLOT 10 -3-03. It is a ferrite core, shielded inductor with a plasitc sleeve. Its nomial valie is 1.75 microH
We then tested the DC output to see if here is a response to light. There was nonel. l
The problem was traced to the new inductor. Surprisingly the inductor coil had lost contact with the pins.
I then replacd the inductor and checked again. The elecronics seems to work okay.. but there is a very small signal 0.8mV for 500microW.
There seems to be still something wrong with the PD or its electronics.
This is the 140 ft. MFD measurement of the VCO phase noise. It is open loop and so should be a good measurement. The RMS is 30 Hz integrated down to 2 mHz.
I don't know why this doesn't agree with Suresh's measurements of the same thing which uses the PLL feedback method.
In BLUE, I also plot the frequency noise measured by using a Stanford DS345 30 MHz func. generator. I think that this is actually the noise of the FD (i.e. the SR560 preamp) and not the DS345. Mainly, it just tells you that the PINK VCO noise measurement is a real measurement.
I calibrated it by putting in a 5 kHz_pp triangle wave on the sweep of the DS345 and counting the counts in DV.
Today I was working on RF distribution box.
So far I almost finished to electronically isolate voltage regulators from the box wall by inserting mica sheet, sleeve, and washers.
The problem I found is the resistance between wall and the voltage regulator is order of M ohms
I checked my isolation (mica sheet and sleeve and washer) but there is no problem there.
But I found that the power switch is not completely isolated from the wall.( around 800 kohm)
and that the resistance between the regulator and the wall is smaller for the regulator closer to the power switch
and greater for the regulator less closer to it.
So I think we need to put washer or sleeve to isolate the powersitch electronically from the box wall.
Suresh or I will fix this problem
[ To Suresh, I can finish the isolation when I come tomorrow. Or you can proceed to finish isolation.]
Frank put his low noise preamp info here.
I suggest that we build these (using Altium) but replace the cheapo transistors with the high class MAT03 matched BJT pair from Analog Devices.
This will allow us to have a pre-amp better matched to the noise of the mixers down to low frequency.
Most of the RF cables required for the box are done. There are two remaining and we will attend to these tonight.
We expect to have finished the mechanical assembly by Sunday and start a quality test on Monday.
The mechanical assembly of RF distribution box is 99% complete. Some of the components may be bolted to the teflon base plate if needed.
All RF cables and DC voltage supply lines have been installed and tested. I removed the terminal block which was acting as a distribution box for the common zero voltage line. Instead I have used the threaded holes in the body of each voltage regulator. This allows us to keep the supply lines twisted right up to the regulator and keeps the wiring neater. The three regulator bodies have been wired together to provide a common zero potential point.
I did a preliminary test to see if everything is functioning. All units are functioning well. The output power levels may need to be adjusted by changing the attenuators.
The 2x frequency multiplier outputs are not neat sine waves. They seem to produce some harmonics, unlike the rest of the components.
I will post the measured power output at each point tomorrow. The RF power meter could not be found in the 40m lab. We suspect that it has found its way back to the PSL lab.
Frank is recommending these PhaseTrack-210 as phase stable low loss rf coax cables.
We wish to have roughly 2 dBm of output power on each line coming out of the RF distribution box. So I adjusted the attenuators inside the box to get this.
I also looked at why the 2x output looked so distorted and found that the input power was around 17 dBm whereas the maximum allowed (as per the datasheet of Minicircuits MK-2) is 15dBm. So I increased the attentuation on its input line to 5dBm (up by 2dBm) The input power levels are around 14.6dBm now and the distortion has come down considerably. However the net output on the 2x lines is now down to 0.7dBm. We will have to amplify this if we need more power.
The schematic and the power output are now like this:
[Kevin, Rana, Koji]
I calculated the dark noise of POX and POY due to Johnson noise and voltage and current noise from the MAX4107 op-amp using nominal values for the circuit components found in their data sheets. I found that the dark noise should be approximately 15.5 nV/rtHz. The measured dark noise values are 18.35 nV/rtHz and 98.5 nV/rtHz for POX and POY respectively. The shot noise plots on the wiki have been updated to show these calculated dark noise sources.
The measured dark noise for POY is too high. I will look into the cause of this large noise. It is possible that the shot noise measurement for POY was bad so I will start by redoing the measurement.
This experiment deals with measuring the total harmonic distortion (THD) contribution of mixers in a circuit.
(a circuit diagram is attached) where:
Mixer: ZFM-3-S+ at +7dBm
Attenuator: VAT-7+, at +7dB
Low-pass filter: SLP-1.9+, which is set to DC-1.9MHz
The total harmonic distortion can be calculated by the equation:
where Vn represents the voltage of the signal at a certain harmonic n.
In this experiment, only the voltages of the first three harmonics were measured, with the first harmonic at 400Hz, the second at 800Hz, and the third at 1.2kHz. The corresponding voltages were read off the spectrum analyzer after it had time averaged 16 measurements. (picture of the general shape of the spectrum analyzer output is attached)
(results for this mixer's particular configuration are on the pdf attached)
There really isn't that much correlation between the modulations and the resulting THD.
We won't know how good these numbers are until more experiments on other mixers are done, so they can be compared. Since the rest of the mixers are relatively high levels (+17dBm, +23dBm in comparison to this experiment's +7dBm), an RF amplifier will need to be hooked up first to do any further measurements.
The previous measurement for the shot noise of POY had the dark noise at ~100 nV/rtHz. I redid the measurement and got 26 nV/rtHz for the dark noise. I think that when I made the previous measurement, the spectrum analyzer had automatically added some attenuation to the input that I failed to remove. This added attenuation raised the noise floor of the measurement making the dark noise of POY appear larger than it is.
The updated measurement can be found on the wiki at http://lhocds.ligo-wa.caltech.edu:8000/40m/Electronics/POY.
Finished calculations for harmonic distortion at each of the 10 outputs of the RF distribution box. The diagram can be found on Suresh's post http://nodus.ligo.caltech.edu:8080/40m/4342
THD calculation consisted of gather data on the dBm at harmonics of the fundamental frequency. These dBm values were converted into units of power and plugged into the appropriate THD equation pulled from Wikipedia:
On the table, the number 1-6 correspond to the harmonic number of the input frequency used. For example, the first five PD's listed used an 11MHz source, while the second set of five PD's listed used a 55MHz source. Values listed under certain harmonics are dBm measurements at the corresponding frequency. The P-subscript values are essentially the dBm measurements converted to units of power (Watts) for ease of calculation in the equation above. THD is then calculated using these power units; I have converted the ratios to percentages.
It should be noted that as with all THD calculations, the more data points collected, the more precise the THD % will be.
By the way, the outputs on the physical RF distribution box for REFL165 and AS165 are actually labeled as REFL166 and AS166.
Fast work indeed! It would be nice if we could have the following details filled in as well
a) A short title and caption for the table, saying what we are measuring
b) the units in which this physical quantity is being measured.
It is good to keep in mind that people from other parts of the group, who are not directly involved in this work, may also read this elog.
I measured the transfer function, shot noise, and dark spectrum of AS55.
From the shot noise measurement, the RF transimpedance is (556.3 +- 0.8) Ohms and the dark current is (2.39 +- 0.01) mA. The dark noise agrees with the approximate value calculated from the circuit components.
There are no anomalous oscillations when there is no light on the photodiode. I am working on fitting the transfer function in LISO but the other plots are on the wiki at http://blue.ligo-wa.caltech.edu:8000/40m/Electronics/AS55
We want to increase gain in the lower frequencies, so a circuit must be designed (a passive low pass filter).
First, measurements were taken at the X arm for impedance and capacitance, which were 104.5kOhms and 84.7pF respectively. Kiwamu decided to make the circuit resemble a voltage divider for ease of calculation, such that Vout/Vin would be a ratio of some values of the equivalent circuit reactance values. After a few algebra mistakes, this Vout/Vin value was simplified in terms of the R, C measured and the R', C' that would be needed to complete the circuit.
Since the measured C was very small and the measure R was fairly high, the simplified form allowed us to pick values of R' and C' that would make the critical frequency occur at 0.1Hz: set the R' resistance to 1MOhm and C' capacitance to 10uF, which would yield a gain ~1.
With these values a circuit we can start actually making the circuit.
[Steve, Suresh, Larisa]
The following cables were laid today: ETMYT, ETMY, IFOPO, MC1, OMCR, AS Spare, and MC2T.
Though the paper suggested 135' for the MC2T, we used a 110'. This is too short: need at least another 15' for the MC2T.
The RCR cable wasn't crossed off on the list, but a cable exists at the RCR cable which is black and is labeled (old label, 75 ohms)
There was no indication of which length was needed for MC1, so a 95' cable was used.
This is the continuation of http://nodus.ligo.caltech.edu:8080/40m/4402
The first picture is of the actual component, where the resistor is 1M and capacitor is 10uF.
But before the component can be put into place, its transfer function had to be checked to make sure it was doing what we calculated it would do. The results of these are in the graphs generated: frequency vs. gain, and frequency vs. phase.
According to these graphs, we are not achieving the targeted cutoff frequency: need to recalculate and compensate for the extra 100k resistance being encountered.
For bode plot:
USE LOG-LOG plot for the amplitude
USE LOG-LINEAR plot for the phase
Search "Bode Plot" on web
I have prepared several diagrams outlining the current state of the RF System.
These are uploaded into the svn40m here and will be kept uptodate as we complete various parts of the task. These plans have taken into account
the new priorities of the LSC (set out by Koji here )
We (Koji, Kiwamu and I) took stock of the RF cables which we have inherited from the earlier RF system and have made new plans for them.
I took stock of the filters purchased for the modifying the demod boards. We have pretty much everything we need so I will start modifying the boards right away. The following table summarises the modifications
LP Filter (U5)
We seem to have a spare SHP-175. I was wondering where that is supposed to go.
This is the status and tentative schedule for completing the various tasks. I have put the dates based on priority and state of the hardware.
The RF Cable layout plans are drawn on top of a Lab Layout. The various subsystems are drawn (not to scale) on separate layers. The graffle files are located here . I thought they might come in handy for others as well.
[Steve, Kiwamu, Larisa]
Having finished laying new cable last week, we moved on to connecting those on PSL table and AP table.
--RCR, RCT, PMCR (all three are blue)
--OMCR (blue cable, ***now has a camera***), PMCT, IMCR, REFL, AS (white cable), OMCT (***now has camera***)
Unless otherwise noted, the cables are black on the AP table. Also on the AP table: cables were connected directly to the power source.
The wiki has been updated accordingly.
Steve noted that MC2T and POP cameras are not there.
Ooh. Can you explain the purpose of the resistors which are connected to the (+) inputs? It looks like some real electronics ninjitsu.
51 Ohm for CLC409
The datasheet of CLC409 uses 25Ohm there. This is to cancel the input bias current of the two inputs of the opamp.
The source impedance (series) of SGD444 is 50Ohm. So I used 50Ohm for the + input shunting.
However, I could probably use anything between 0-50Ohm as the datasheet itself tells that the bias currents are
not related between the two inputs. In addition, I am not sure how much the real series resistance of the PD is.
1kOhm for OP27
This resister is to ensure the (+) input to have a high impedance at high frequencies.
As far as OP27 is behaving as an ideal opamp, the (+) input has a high impedance.
Also if the inductor behaves as the ideal inductor, no photocurrent comes to the AF path.
However, if both of the op27 and the inductor show similar impedances to the RF transimpedance of 240Ohm,
the AF path absorbs some photocurrent and affects the RF transimpedance of the RF output.
We know that the inductor has a self resonance where the shunt capacitance take over the impedance of the coil.
Above that frequency, the inductor is no longer the inductor. The self resonant freq of this inductor is ~300MHz. It is OK, but not
too far from the freq of interest if we like to see clear cut off at around f>100MHz.
Also OP27 is an AF amplifier and I had no confidence about the input impedance of the OP27 at 100~300MHz.
If I put 1k in the (+) input of the OP27, I can ensure the entire AF path has the impedance of ~1k (at least 500Ohm even when L and OP27 are shorted).
I think the chip resister easily works as a resister up to 1GHz.
I brought TTFSS set #7 to 40m and kept it in the electronic cabinet.
note that Q4 transistor has not been replaced back to PZT2907A yet. It's still GE82.
Q3 is now pzt3904, not PZT2222A.
The (-) input has been decoupled by the capacitor. So the series resistance of the PD is not the matter.
In this sense, we should use 0Ohm for the (+) input shunting.
However, I could probably use anything between 0-50Ohm as the datasheet itself tells that the bias currents are
not related between the two inputs. In addition, I am not sure how much the real series resistance of the PD is.
The following Video MUX inputs(cameras) and outputs(monitors) have been checked:
MC2F, FI, AS Spare, ITMYF, ITMXF, ETMYF, ETMXF, PSL Spare, ETMXT, MC2T, POP, MC1F/MC3F, SRMF, ETMYT, PRM/BS, CRT1(MON1), ETMY Monitor, CRT2(MON2), CRT4(MON4), MC1 Monitor, CRT3(MON3), PSL1 Monitor, PSL2 Monitor, CRT6(MON6), CRT5(MON5), ETMX Monitor, MC2 Monitor, CRT9, CRT7(MON7), CRT10, and Projector.
Their respective statuses have been updated on the wiki: (wiki is down at the moment, I will come back and add the link when it's back up)
Plotting the data points yielded by the spec analyzer of my first LPF yielded a result that was not expected: the desired cutoff frequency wasn't achieved because of some extra 100k resistance that wasn't taken into consideration. (see here ). I have redrawn the Bode graphs for this configuration so that it is easier to see that it is wrong (first attachment)
After some calculation adjustments, it was found that the capacitor value could remain at 10uF, but the resistance needed to be changed to 100k to maintain a gain of 0.5 and critical frequency at 0.1Hz. Second attachment is the Bode graph that results from this configuration.
Note: Bode graphs are both in Log-Linear scales (Wikipedia said so)
The performance plots for POX_11 in the wiki are horrendous and the schematic is missing.
I opened up the box and found all kinds of horrors. There were multiple tunable parts and a flurry of excess nonsense.
The top 2 worst offenders:
1) The main tunable inductor was busted. I removed it and found that the coil was open. Too much indelicate soldering in its vicinity had melted the wire. Someone had put extra inductors and capacitors around it to make it seem as if the PD was working fine, but the noise performance was off by a factor of ~100.
2) The MAX4107 had a 1.4k series resistor. This make the output go through a 1450/50 voltage division which is not nice for the SNR. I removed it.
I then struggled for awhile to get a sensible response. It turned out that the TEST IN input was not giving me a sensible TF. Jenne and I fired the Jenne laser at it and found that the 11 MHz main resonance is there. In the morning I'll finish this off and post more results. I think its going to end up being fine.
I used the Jenne AM laser to tune up the PD (used to be POX_11 but now is called REFL_11). In addition to the notch at 22 MHz, I have also put in a LC notch at 5*f = 55.3 MHz. The transfer function below shows the RF OUT of the PD v. the drive to the laser. I didn't divide out by the 1811 because its not on the EE bench.
A new 90 deg splitter, PSCQ-2-51W, has been installed on another demod board called AS55.
It shows a reasonably close 90 degree separation between the I and Q signals at 55 MHz with various LO and RF power.
So far we have ordered only three PSCQ-2-51Ws for test. Now we will order some more for the other demodulators.
Some plots will be posted later.
Figure.1 I-Q relative phase measurement as a function of LO power.
Blue curve : relative phase of AS55 that I have modified today (#4572).
Red curve : relative phase of AS11 that I had modified a week ago (#4554). Just for comparison.
The relative phase of AS55 agrees approximately what we expected according to the datasheet of PSCQ-2-51W. We expected 85 degree.
Figure.1 I-Q amplitude imbalance as a function of LO power.
From - 5 dBm to 5 dBm in LO power the imbalance is within 3 %.
But the precision of the measurement is also about 2 % (because I used an oscilloscope). Even so the imbalance is still good.
Some plots will be posted later.
I found that all the Heliax cables landing on the bottom of 1Y2 were too loose.
Due to this loose connection the RF power at 55 MHz varies from -34 dBm to 3 dBm, depending on the angle of the Heliax's head.
The looseness basically comes from the fact the black plate is too thick for the Heliax cable to go all the way. It permits the Heliax's heads to rotate freely.
What we should do is to make countersinks on the black plate like this:
The countersink gives rise to another problem when we mount the N-type-to-SMA bulkhead adaptor. As we are making a circular hole in the plastic strip (instead of a hole with two flat sections) the adaptor is free to turn when we tighten it with a wrench. We currently hold the smooth circular part on the other side with a gripping pliers and while tightening. If that part disappears into the countersink (as seen in the pics) we will not be able to tighten the adaptor sufficiently and consequently we will also not be able to get the heliax connector to be tight.
A better solution would be to use the 1/4-inch plastic L-angle beam which Steve has used on the AS table. In addition to solving this loose connector problem, the beam is also more rigid than the plastic strip.
I was charge with making a Non-Inverting Op Amp Low Pass Feedback circuit for Jenne, which may somehow be integrated into a seismometer project she's working on.
Circuit diagram is attached. Calculations show that R1, R2 and C have the following relationship: if R1=10^n, R2=10^(n+1), C=10^(-n-4). For the particular circuit being modeled by the transfer function, R1=100 Ohm, R2=1k Ohm, and C=1uF.
Attached also is the circuit's Bode plot, showing frequency versus gain and phase, respectively. The frequency versus gain graph is true to what the circuit was calculated to generate: a gain of +20 and a cutoff frequency at 200Hz. Not sure what's going on with the frequency verus phase plot.
This is a continuation of this
The low pass filter is finally acceptable, and its Bode graph is below (on a ~3Hz frequency span that shows the cutoff frequency is at 0.1Hz)
My observation wasn't accurate enough.
The looseness came from the fact that the N-SMA bulk heads were slipping on the black plate.
This is actually what Suresh pointed out (see here). So the thickness of the black plate doesn't matter in this case.
Somehow I was able to tighten the bulk heads using two wrenches and I think they are now tight enough so that the heliax's heads don't move any more.
Building on what was posted previously
The configuration has now evolved into an Inverting Op Amp Feedback Low Pass Filter circuit.
Had to change out some components to satisfy conditions: R1=1k Ohm, R2=10k Ohm, C=0.1uF. These were changed in order to decrease the magnitude of the current passing through the op amp by a factor of 10 (10V supplied through the R1 resistor yields about 10mA). The configuration itself was changed from non-inverting to inverting in order to get the frequency vs. gain part of the Bode plot to continue to decrease across higher frequencies instead of leveling off around 4kHz.
Having finished the bulk of the work for the LPF itself ( see here ), I have begun trying to design the seismometer box to Jenne's specifications.
Currently looking into what the voltage buffer amplifier might look like for this.
Suggestions/corrections would be much appreciated!
Yesterday we found that MC3 OSEM LL PD did not have a sensible signal - the readback was close to zero and it was making MC move around. I disabled the PD LL so that the damping is done with just three face plus side PDs. There still no signal from MC3 LL PD today. It needs debugging.
The schematic for the seismometer box from this last time has been updated...
Koji was helpful for coming up with a general diagram for the voltage buffer amplifier, which has now been added to the configuration pictured below.
The only thing that remains now before I try to plot it with Eagle/LISO is to pick an op amp to use for the voltage buffer itself. Someone suggested THS4131 for that (upon Googling, it hit as a "high speed, low noise, fully-differential I/O amplifier"). It looks good, but is it the best option?
[Valera / Kiwamu]
It was because of a loose connection. Pushing the connector solved the issue.
We really have to think about making reliable connections and strain reliefs.