Looks like there was a mysterious loss of data overnight; since there's nothing in the elog I assume that its some kind of terrorism. I'm going to call Rolf to see if he can come in and work all night to help diagnose the issue.

 again

Friday: 

In addition to the other fixes, Alex rebuilt the daqd process. I failed to elog this. When he rebuilt it, he needed change the symmerticom gps offset in the daqdrc file (located in /opt/rtcds/caltech/c1/target/fb). 

On Friday night, Kiwamu contacted me and let me know the frame builder had core dumped after a seg fault.  I had him temporarily disable the c1ass process (the only thing we changed that day), and then replaced Alex's rebuilt daqd code with the original daqd code and restarted it.  However, I did not change the symmetricom offset at this point.  Finally, I restarted the NDS process.  At that point testpoints and  trends seemed to be working.

Sunday:

The daqd process was restarted sometime on Sunday night (by Valera i believe).  Apparently this restart finally had the symmetricom gps offset kick in (perhaps because it was the first restart after the NDS was restarted?).  So data was being written to a future gps time.

Monday:

Kiwamu had problems with testpoints and trends and contacted me.  I tracked down the gps offset and fixed it, but the original daqd process only started once successfully, after that is was segfault, core dump non-stop. I tried Alex's rebuilt daqd (along with putting the gps offset to the correct value for it), and it worked.  Test points, trends, excitations were checked at the point and found working.

I still do not understand the underlying causes of all these segmentation faults with both the old and new daqd codes.  Alex has suggested some new open mx drivers be installed today.

-- Alignment

     - C1ASS

     - MC WFS 

    - OPLEVs

-- Optimization of Suspended Optics

  - diagonalizations

  - damping control

  - upgrading of electronics

   - weekly check

-- Input Optics

   - PMC

         * mode matching
         * epics LO HI values

   - FSS and ISS

        * recover FSS
        * make ISS working

   - EOM

   - doubling and RFPD for green

-- Length Sensing and Control

   - digital system and electronics

-- Green Locking

    - X end station

    - Y arm green locking ==> Suresh/Bryan

    - digital control systems

      - noise budget

   - cavity scan and handing off

-- misc.

- Put priority on the list

- Put names on the items

- Where is the CDS TO DO ==> Joe

-

- Remote disconnection of the greeen PDH 

- What is the situation of the PD DC for the LSC PDs?

- SUS Satelite box Resister replacement ==> Jamie

- IMC mode matching ==> Jamie/Larisa 

- Mechanical shutters everywhere

- SRM OPLEV Connection

- MC OAF

- Better LSC whitening boards

- DAFI 

CDS To Do:

1) Get ETMY working - figure out why signals are not getting past the AI board (D000186) to the coils.

2) Get TDS and command line AWG stuff working

3) Get c1ass and new c1lsc (with Koji) fully integrated with the rest of the system.

4) Get CDS software instructions up to date and well organized.

5) Redo cabling  and generally make it a permanent installation instead of hack job:

   a) Measure cable lengths, check connectors, wire with good routes and ensure strain relief.  Ensure proper labeling

   b) Get correct length fiber for c1sus RFM and timing.

   c) Fix up final BO adapter box and DAC boxes.

   d) Make boxes for the AA filter adapters which are currently just hanging.

   e) Get two "faceplates" for the cards in the back of the ETMY IO chassis so they can screwed down properly.

   f) Remove and properly store old, unused cables, boards, and anything else.

6) Create new documentation detailing the current 40m setup, both DCC documents and interactive wiki.

7)  Setup an Ubuntu work station using Keith's wiki instructions

Simulated Plant To Do:

1)Create simulated plant to interface with current end mass controls (say scx).

2) Create proper filters for pendulum and noise generation, test suspension.

3) Propagate to all other suspensions.

4) Working on simulated IFO plant to connect to LSC.  Create filters for near locked (assume initial green control perhaps) state.

5) Test LSC controls on simulated IFO.

6) Fix c code so there's seamless switching between simulated and real controls.

CDS Status:

Here is a partial list of stuff which is being packed at LLO to be shipped to CIT.  The electronics ckt boards are yet to be added to this list.  Will do that tomorrow.

Its been well noted in the past that sweeping the PMC at high power leads to a distortion of the transmitted power curve. The explanation for this was coating absorption and thermo-elastic deformation of the front face of the mirrors.

Today, I did several sweeps of the PMC. I turned off its servo and tuned its PZT so that it was nearly resonating. Then I drove the NPRO via the HV driver (gain=15) with 0-150 V (its 1.1 MHz/V) to measure the PMC transmitted light. I adjusted the NPRO pump diode current from 2A on down to see if the curves have a power dependent width.

In the picasa web slideshow:

There are 3 significant differences between this measurement and the one by John linked above: its a new PMC (Rick says its the cleanest one around), the sweep is faster - since I'm using a scope instead of the ADC I feel free to drive the thing by ~70 MHz in one cycle. In principle, we could go faster, but I don't want to get into the region where we excite the PZT resonance. Doing ~100 MHz in ~30 ms should be OK. I think it may be that going this fast avoids some of the thermal distortion problems that John and others have seen in the past. On the next iteration, we should increase the modulation index for the 35.5 MHz sidebands so as to get a higher precision calibration of the sweep's range.

By eye I find that the FWHM from image #4 is 11 ms long. That corresponds to 300 mV on the input to the HV box and 15 V on the PZT and ~16.5 MHz of frequency shift. I think we expect a number more like 4-5 MHz; measurement suspicious.

We have one single O-ring on the 40m vacuum envelope. It is on the OOC west side, facing the AP table. This O-ring has to be protected from the force of this

door. There should be 3 shims  ~120 degrees apart to carry the full load, so it is not the O-ring that is getting squashed.

This morning I found only one of these shims in place.

Today was a banner day for Simulated Plants.

Joe and Jamie have been working to get it all happening and this afternoon we started stuffing filters into the plant to make it act like the:

We put in the following features so far:

1. Anti-Imaging filters (these are hacked to be approximate since the real ones are 7570 Hz LP filters and the SimAI only can have filters up to 8192 Hz).
2. Dewhitening filters (copied from the SimDW in the SUS-ETMY screens)
3. Coil Driver transimpedance (1 / 200 Ohms)
4. Magnet-coil force constant (0.016 N/A)
5. Conversion from Coil to DOF Basis
6. All DOFs of the mechanical model are represented as simple harmonic oscillators with Q~100 and f ~ measured free swinging peaks.
7. Signals/Noise can be injected either as force noise on the test mass or as displacement noise at the suspension point.
8. Conversion from DOF to Shadow Sensor basis.
9. Optical Levers (with whitening)
10. Shadow Sensors have 2V/mm readout gain and whitening filters before being digitized by the SimADC.

We have also changed the switching logic for the SUS and SimETMs for the shadow sensor whitening. It used to be that either the hardware OR the software whitening was on. Now we have made it like all of the other whitening/antiwhitening in LIGO and the whitening/antiwhitening come on together. Joe and Jamie are going to propagate this to the other SUS. The hardware filter is a 30,100:3 (poles:zeros) whitening filter. The digital filter used to also be 30,100:3 with a DC gain = 1. I've changed the FM1 filter in the XXSEN filter banks into a 3:30 for the ETMY so that it now comes on and just compensates the hardware filter. This change should be propagated to all other SUS and the MEDM screens updated to show the new situation.

After this change, we decided to calibrate the {UL,UR,LL,LR,SD}SEN channels into units of microns. To do this we have made an FM6 filter called 'cts2um' that accounts for the OSEM gain and the ADC conversion factors. These channels are now in units of microns without applying any calibration in the DTT or Dataviewer. The plot below shows the OSEM shadow sensor time series with all damping loops ON and a very rough version of seismic noise being injected in all 6 DOFs (note that the y-axis is microns and the x-axis is seconds).

Next, Jamie is adding the angular calibrations (so that SUSPIT and SUSYAW are in rads) and Chris is making vectift quality seismic noise injectors.

We also need to add coating thermal noise, suspension thermal noise, substrate thermal noise, ADC/DAC noise and a lot of MEDM screen indicators of what state we're in. I myself can't tell from the OSEM time series if its real or Sim.

Just for a record. We got 4 new laser pointers (2 greens, 1 blue, and 1 green and red combination). Don't lose them.

They reside in a bucket on the SP table, where IR viewers and sensor cards also reside.

- Plan for this week

  * Intensity stabilization for the end green laser (Matt / Kiwamu)

  * Hand off the servo from Green to Red (Matt / Kiwamu)

  * Y end green locking (Suresh / Bryan) (rough schedule)

  * Reconnect the X end mechanical shutter to 1X9 (Kiwamu)

  * Connect the end DCPD signal to a DAC (done)

  * Make a LPF in a Pomona box for the temperature (Larisa)

  * Clean up and finalize the X end setup (Kiwamu)

  * Make a item lists for electronics. Order the electronics. (Aidan / Kiwamu)

Some tasks for the daytime tomorrow.

  * Beam profile measurements of the Y end laser  (Suresh / Bryan)

  * Taking care of CDS and the simulated plant (Jamie / Joe)

  * Reconnect the X end mechanical shutter to 1X9 (Kiwamu)

  * LPF for the X end temperature feedback (Larisa)

This is the grand plan we talked about in the beginning of the meeting.

• (Kiwamu) X-end Green cleaning up / Prep for DRMI
• (Bryan) Y-end Green
• (Suresh) Help Bryan / RF (w. Kevin)
• (Jenne) MC WFS / Y-arm IR alignment / MC adaptive feedforward (incl. CDS)
• (Koji) LSC
• (Joe) CDS cleaning up
• (Jamie) Help Joe / Noise Budget
• (Larisa) PMC scan / PSL photo&diagram
• (Barbarela) ASS

With the exception of a 2" mirror mount, I've confirmed that we have everything for the Y-end green production and mode-matching.

We need to calculate a mode-matching solution for the Lightwave laser so that it gives the correct beam size in the doubling crystal.

Additionally, Rana has suggested that we change the pedestals from the normal 1" diameter pedestal+fork combo  to the 3/4" diameter posts and wider bases that are used on the PSL table (as shown in the attached image).

There was a 2" mirror mount among the spares on the PSL table.  It has a window LW-3-2050 UV mounted in it.  I
have moved it to the Y-end table.  We seem to have run out of 2" mirror mounts ...

Skip to final thought ...

Kiwamu and I have set about measuring the contrast of the signal on the RF PD. We can only do this when the end green laser is locked to the cavity. This is because the green transmission through the cavity, when unlocked, is too low. Unfortunately, once we lock the green beam to the cavity, we can't keep the beatnote on the RF PD stable to within a few hundred Hz of DC (remember that the cavity is swinging around by a couple of FSRs). So we also lock the PSL to cavity.

At this point we're stuck because we can't get both of these beams resonant within the cavity AND have the frequency difference between them be less 1kHz - when the lasers are locked to the cavity, their frequencies are separated by an integer number of FSRs + a fixed frequency offset, f_offset, that is set by the phase difference on reflection from the coating between the two wavelengths (532nm and 1064nm). We can never get the frequency difference between the lasers to be less than this offset frequency AND still have them both locked to the cavity.

So our contrast measuring method will have to use the RF signal.

So this is our method. We know the incident power from each beam on the RF PD (see Kiwamu's elog entry here), but to recap,

P_green_PSL = 72 uW (as measured today)

P_green_XARM = 560 uW (as measured by Kiwamu last week).

The trans-impedance of the RF PD is 240 Ohms. We'll assume a responsitivity of 0.25 A/W. So, if the XARM transmission and PSL green beams are perfectly matched then the maximum value of the RF beat note should be:

RF_amplitude_max = 2* SQRT(P_green_PSL*P_green_XARM) * responsivity * transimpedance = 240*0.25*2*(72E-6*560E-6)^(1/2) (volts)

= 24 mV = -19.5 dBm (or 27.5dBm after the +47 dB from the two  ZFL-1000LN+ amplifiers - with +15V in - that protrude from the top of the PD)

The maximum RF strength of the beat-note that we measure is around -75 dBm (at the RF output of the PD). This means the contrast is down nearly 600x from optimal. Or it means something is broken.

Final thought: at the end of this procedure we found that the RF beat note amplitude would jump to a different and much higher amplitude state. This renders a lot of the above useless until we discover the cause.

 I gutted one of the $2 red laser pointers to build a laser source whose amplitude we could modulate at RF frequencies. Basically, I cut off the bulk of the housing from the pointer and soldered a BNC connection into the two terminals that the 2x 1.5V batteries were connected to. When I applied 3V across this BNC connector the diode still worked. So far so good.

Next I added a bias tee to the input. I put 3V across the DC input of the bias tee and added a -3dBm signal into the RF port of the tee. The laser beam was incident on a PDA100A (bandwidth of 1.7MHz) and, sure enough, Kiwamu and I could see a flat response in the amplitude at a given drive frequency out to around 1.7MHz.

We should check the response on a faster PD to see how fast the laser diode is, but we should be able to use this now to check the RF response of the green beat note PD. 

TO DO:

1. Add some capacitors across the DC input of the bias tee.

2. Do something about the switch on the laser diode.

3. Attach some labels to the laser that specify what is the required DC voltage and the maximum acceptable RF modulation amplitude.

I measured the DC response of the Green PD

Power into PD at DC (green laser pointer) = 285 uW
Voltage out of PD = 552mV/(100x SR560gain) = 5.52mV
Photocurrent = 5.52mV/(241 Ohms)*3 = 68.7uA
Responsivity = 68.7/285 = 0.24 A/W

Therefore, since the responsivity is in the correct range for a Silicon PD at 532nm, the DC output is giving us sensible response to an input signal.


But, there is a 2.12MHz, 328mV oscillation on the DC output irrespective of the incident power.
 

According to the measurement done by Aidan and me, there are two beat-note state.

One gave us a small beat signal and the other gave us a bigger signal by approximately 20 dB.

 A possible reason for this phenomenon is that the end laser is operating at a special temperature that somehow drives the laser with two different modes at the same time.

So that it permits the laser sometimes locked with one of the two modes and sometimes with the other mode.

For the first step we will change the temperature such that the laser can run with a single stable mode.

Then for investigating it we will put a scanning cavity on the X end table to see if it really exhibits a two modes or not.

Mike Caton of Konecranes inspected and loadtested all 3  of the 40m cranes at max reach trolley positions with 1 ton.

 Low power green-red laser pointers are in. High power green, red and blue pointers are confiscated.

Since last Friday I have been testing the broadband RF photodetector in order to figure out the capability of S3399 with the similar circuit as Matt's BBPD
We also like to figure out if it has sufficient performance for the 40m green locking.

The circuit diagram is shown in the first attachment. The RF amplifier is attached at the diode while the reverse bias voltage is applied at the other side of the diode. The amplifier's input impedance is used as the transimpedance resister. Note that the bandwidth of this configuration is limited by the RC filter that consists of the junction capacitance of the diode, the series resistance of the diode, and the transimpedance resister. This cut off freq is in general lower than that cut off obtained with the usual transimpedance amplifier which has the readout resister at the feedback path of the opamp.

The transfer function of the PD is measured using Jenne's laser. At the reverse bias voltage of 30V, the -3dB bandwidth of 178MHz was obtained. This is quite high bandwidth for the most of the applications at the 40m.

Because of the low transimpedance the low-noise level of the RF amplifier is very crucial. Recently we can obtain an ultra low noise RF amplifier like Teledyne Cougar AC688 which has the NF of 0.9dB with the bandwidth between 10MHz - 600MHz. Next step will be to obtain this kind of amplifier to test the noise performance.

Last Thursday, I tested Newport Servo Controller LB1005 with the X_arm green PDH servo.

The setup and the settings I could lock the arm is depicted in the attached figure.
To lock the cavity, follow the steps below

1) Toggle the switch to the "lower" position. This disengages the servo and reset the integrator.

2) Toggle the switch to the "middle" position. The zero freq is set to the "PI corner" freq. At the low freq the gain is limited
at the value of "LF Gain Limit". This gives us a single pole at the low freq.

3) Once the lock is acquired, toggle the switch to the "upper" position. This moves the pole freq to DC, resulting in the complete integration of the signal at the low frequency.

I measured the openloop transfer function (attachment 2). The amp is quite fast and exhibits almost no phase delay upto 100kHz.
The UGF was 10kHz with the phase mergin of ~45deg. I had to tune the input offset carefully to stay at the center of the resonance.

I moved the Hartmut Green PD to the Jenne laser bench to try to determine if the response at RF was reasonable or somehow very much smaller than it should be. It was set up as shown in the attached diagram. The first pass at this was by comparing the ratio of the RF photocurrent of the green PD to the RF photocurrent of the New Focus 1611 InGaAs PD. That ratio (at a sufficiently low frequency) should be the same as the ratio the DC photocurrents of the two PDs.

Using the network analyzer I measured the ratio of the voltages of the two RF signals (and then scaled each of these by the respective transimpedances of the PDs: 700 Ohms for the 1611 and 240 Ohms for the Harmut PD). The resulting ratio is shown in the attached plot.

I measured the DC voltages from each PD and scaled those by the transimpedances to get the photocurrent (10 kOhm for the 1611 and 80 Ohm effective for the Harmut PD). The ratio of the DC photocurrents was 0.37. This is roughly 3x the ratio of the RF photocurrents at 500kHz (=0.14). This discrepancy is uncomfortably large.

 The full set of measurements is given in the table below:

There is one other troubling point: using the estimate of responsivity on the Harmut PD * incident power * transimpedance at DC = (0.02A/W) * (0.28mW) * (80 V/A) = 0.45 mV.

But the measured DC voltage is 6.5mV = inconsistent.

 Konecrane representative gave crane operator training in the 40m. Koji has become a qualified, trained crane operator of the 40m lab.

I think I had underestimated the responsivity of the Silicon PD at 1064nm. The previous value was based on a rough search online for the responsivity of Silicon (I couldn't find the product number of the actual PD we are using). For instance, the PDA100A Si detector from Thorlabs has a responsivity of 0.35-0.4A/W at 1064nm. 

If we calculate the responsivity of the Hartmut PD from the measurements I made today (input power = 0.300mW, output voltage = 5.56mV, effective transimpedance = 80 Ohms), then the responsivity at 1064nm is 0.23 A/W which is not an unreasonable number given the response of the Thorlabs detector.

Responsivity of SGD-444A

Having convinced myself that the green Hartmut PD is giving an acceptable response at RF frequencies I decided to double-check the beatnote at IR (fiber transmission from the X-end beating with the PSL). This took a while because I had to realign the beam into the fiber at the X-end (I had a PD monitoring the output from the fiber on the PSL table and 40m of BNC cable giving me the signal from it at the X-end).

Eventually, I managed to get a beatnote on the PD. At first there was no signal at the temperature calculated using Koji and Suresh's calibration, but it turned out that the mode-overlap wasn't good enough on the PD. Now I can clearly see beats between a couple of modes, one of which is much stronger than the other. I think we should use a frequency discriminator on the output from the IR PD to servo the end laser and keep the strong beat note within <100MHz of DC.

An RF combiner should be included in the triple resonant box because it eases impedance mismatching and hence lowers undesired RF reflections.

Therefore we should use three cables to send the RF signals to the box and then combine them in the box.

(RF combiner)

 With proper terminations an RF combiner shows 50 Ohm input impedance.

But it still shows nearly 50 Ohm input impedance even if the source port is not properly terminated (i.e. non 50 Ohm termination).

This means any bad impedance mismatching on the source port can be somewhat brought close to 50 Ohm by a combiner.

  The amount of deviation from 50 Ohm in the input impedance depends on the circuit configuration of  the combiner as well as the termination impedance.

For example a resistive 3-way splitter shows 40 Ohm when the source port is shorten and the other ports are terminated with 50 Ohm.

Also it shows 62.5 Ohm when the source port is open and the other ports are terminated with 50 Ohm.

In this way an RF combiner eases  impedance mismatching on the source port.

(RF signal transfer at the 40m)

 According to the prototype test of the resonant box it will most likely have a non-50 Ohm input impedance at each modulation freqeucy.

If we install the resonant box apart from the combiner it will create RF reflections due to the mismatch (Case 1 in the diagram below)

The reflection creates standing waves which may excite higher harmonics and in the worst case it damages the RF sources.

 To reduce such a reflection one thing we can do is to install the combiner as a part of the resonant box (Case 2).

It will reduce the amount of the mismatching in the input impedance of the resonant circuit and results less reflections.

A rule we should remember is that a cable always needs to be impedance matched.

 The input impedance of the resonant box was measured when an RF combiner was attached to the box.

Indeed the combiner makes the impedance more 50 Ohm and reduces the reflection.

**** measurement conditions ****

* The output of box, where the EOM will be connected,  was open so that the box tries resonating with a parasitic capacitor instead of the real EOM.

* ZFSC-3-13, a 3-way combiner from mini circuit, was used.

* The S-port of the combiner was directly attached to the box with a short connector (~ 30 mm).

* Port 1 and 2 are terminated by 50 Ohm.

* The input impedance was measured on port 3 with AG4395A net work analyzer.

* Reflection coefficient 'Gamma' were calculated from the measured impedance 'Z' by using an equation Gamma = (50-Z)/(50+Z).

The resonances are found at 11, 29 and 73 MHz (55 MHz resonance was shifted to 73 MHz because of no EOM).

Note that the resonances are at frequencies where the notches appear in the reflection coefficient plot.

Don't be confused by a peak at 70 MHz in the impedance. This is an extra resonance due to a leakage inductance from the transformer in the circuit.

I realized that my impedance matching theory on an RF combiner was wrong !

In fact an RF combiner behaves more like an attenuator according to a reflection measurement that I did today.

A 3-way combiner reduces power of an input signal by a factor of 4.8 dB because it can be also considered as a 3-way splitter.

So it is just a lossy component or in other words it is just an attenuator.

(reflection measurement)

To check my speculation that I posted on #4504 I measured reflection coefficients for both cases.

In the measurement I used a heliax cable, which goes from 1X2 rack to the PSL table with a length of about 10 m. Note that this is the cable that had been used as '33 MHz EOM'.

At the input of the heliax cable it was connected to a direction coupler to pick off reflections and the reflected signal was sampled in AG4395A.

The other end of the cable (output side of the cable) was basically connected to the resonant box.

Then I did a reflection measurement for both cases as drawn in this entry (see #4504).

  - case 1 -  the combiner was inserted at the input side of the heliax cable.

  - case 2 - the combiner was directly attached to the resonant box

On the combiner, ZFSC-3-13, the port 1 and 2 were terminated with 50 Ohm, therefore the port 3 was used as an input and the source port is the output.

Here is a resultant plot of the reflection measurements.

Note that whole data are calibrated so that it gives 0 dB when the output side of the heliax is open.

There are two things we can notice from this plot:

 (1) The reflection coefficient at the resonant frequencies (where notches appear) are the same for both cases.

 (2) Over the measured frequency range the reflections were attenuated by a factor of about 9.6 dB , which is twice as large as the insertion loss of the combiner.

These facts basically indicates that  the RF combiner behaves as a 4.8 dB attenuator.

Hence the location of the combiner doesn't change the situation in terms of RF reflections.

Here is the idea how we upgrade the demodulation boards.

Basically we go ahead with two steps as depicted in the cartoon diagram below.

Once we finish the first step of upgrade, the board will be ready to install although the circuit won't be awesome in terms of noise performance.

* * * (details) * * *

 First of all we will replace the home-made 90 degree splitter (see this entry) by a commercial splitter, PSCQ-2-51-W+ from Mini circuit. This is the step 1 basically.

At this point the boards will be ready to use in principle. I asked Steve to get three 90 degree splitters so that we can have at least three demodulators for the dual-recycled Michelson locking.

If they work very fine we will buy some more 90 degree splitters for full locking.

While we try to lock the dual-recycled Michelson once we will get a Cougar amplifier, remove all ERA-5s and install it such that we don't have to gain up and down in the circuit. This is the last step.

This is an overview of how I got (almost) all the CDS tools running on pianosa, the new Ubuntu 10.04 control room work station.

This is machine is experiment in minimizing the amount of custom configuration and source code compiling. I am attempting to install as many tools as possible from existing packages in

available packages

I was able to install a number of packages directly from the ubuntu archives, including fftw, grace, and ROOT:

apt-get install \
 libfftw3-dev \
 grace \
 root-system

LSCSOFT

I installed all needed LSCSOFT packages (framecpp, libframe, metaio) from the well-maintained UWM LSCSOFT repository.

$ cat /etc/apt/sources.list.d/lscsoft.list
deb http://www.lsc-group.phys.uwm.edu/daswg/download/software/debian/ squeeze
deb-src http://www.lsc-group.phys.uwm.edu/daswg/download/software/debian/ squeeze contrib

sudo apt-get install lscsoft-archive-keyring
sudo apt-get update
sudo apt-get install ldas-tools-framecpp-dev libframe-dev libmetaio-dev lscsoft-user-en

You then need to source /opt/lscsoft/lscsoft-user-env.sh to use these packages.

EPICS

There actually appear to be a couple of projects that are trying to provide debs of EPICS. I was able to actually get epics working from one of them, but it didn't include some of the other needed packages (such as MEDM and BURT) so I fell back to using Keith's pre-build binary tarball.

Prereqs:

apt-get install \
 libmotif-dev \
 libxt-dev \
 libxmu-dev \
 libxprintutil-dev \
 libxpm-dev \
 libz-dev \
 libxaw7-dev \
 libpng-dev \
 libgd2-xpm-dev \
 libbz2-dev \
 libssl-dev \
 liblapack-dev \
 gfortran

Pulled Keith's prebuild binary:

cd /ligo/apps
 wget https://llocds.ligo-la.caltech.edu/daq/software/binary/apps/ubuntu/epics-3.14.10-ubuntu.tar.gz
 tar zxf epics-3.14.10-ubuntu.tar.gz

GDS

I built GDS from svn, after I fixed some broken stuff [0]:

cd ~controls/src/gds
svn co https://redoubt.ligo-wa.caltech.edu/svn/gds/trunk
cd trunk
#fixed broken stuff [0]
source /opt/lscsoft/lscsoft-user-env.sh
./bootstrap
export GDSBUILD=online
export ROOTSYS=/usr
./configure --prefix=/ligo/apps/gds --enable-only-dtt --with-epics=/ligo/apps/epics-3.14.10
make
make install

dataviewer

I installed dataviewer from source:

cd ~controls/src/advLigoRTS
svn co https://redoubt.ligo-wa.caltech.edu/svn/advLigoRTS/trunk
cd trunk/src/dv
#fix stupid makefile /opt/rtapps --> /ligo/apps
make
make install

I found that the actual dataviewer wrapper script was also broken, so I made a new one:

$ cat /ligo/apps/dv/dataviewer
#!/bin/bash
export DVPATH=/ligo/apps/dv
ID=$$
DCDIR=/tmp/${ID}DC
mkdir $DCDIR
trap "rm -rf $DCDIR" EXIT
$DVPATH/dc3 -s ${NDSSERVER} -a $ID -b $DVPATH "$@"

environment

Finally, I made a environment definer file:

$ cat /ligo/apps/cds-user-env.sh
# source the lscsoft environment
. /opt/lscsoft/lscsoft-user-env.sh

# source the gds environment
. /ligo/apps/gds/etc/gds-user-env.sh

# special local epics setup
EPICS=/ligo/apps/epics
export LD_LIBRARY_PATH=${EPICS}/base/lib/linux-x86_64:$LD_LIBRARY_PATH
export LD_LIBRARY_PATH=${EPICS}/extensions/lib/linux-x86_64:$LD_LIBRARY_PATH
export LD_LIBRARY_PATH=${EPICS}/modules/seq/lib/linux-x86_64:$LD_LIBRARY_PATH
export PATH=${EPICS}/base/bin/linux-x86_64:$PATH
export PATH=${EPICS}/extensions/bin/linux-x86_64:$PATH
export PATH=${EPICS}/modules/seq/bin/linux-x86_64:$PATH

# dataviewer path
export PATH=/ligo/apps/dv:${PATH}

# specify the NDS server
export NDSSERVER=fb 

[0] GDS was not compiling, because of what looked like bugs. I'm not sure why I'm the first person to catch these things. Stricter compiler?

To fix the following compile error:

TLGExport.cc:1337: error: ‘atoi’ was not declared in this scope

I made the following patch:

Index: /home/controls/src/gds/trunk/GUI/dttview/TLGExport.cc
===================================================================
--- /home/controls/src/gds/trunk/GUI/dttview/TLGExport.cc	(revision 6423)
+++ /home/controls/src/gds/trunk/GUI/dttview/TLGExport.cc	(working copy)
@@ -31,6 +31,7 @@
 #include <iomanip>
 #include <string.h>
 #include <strings.h>
+#include <stdlib.h>
 
 namespace ligogui {
    using namespace std;

To fix the following compile error:

TLGPrint.cc:264: error: call of overloaded ‘abs(Int_t&)’ is ambiguous

I made the following patch:

Index: /home/controls/src/gds/trunk/GUI/dttview/TLGPrint.cc
===================================================================
--- /home/controls/src/gds/trunk/GUI/dttview/TLGPrint.cc	(revision 6423)
+++ /home/controls/src/gds/trunk/GUI/dttview/TLGPrint.cc	(working copy)
@@ -22,6 +22,7 @@
 #include <fstream>
 #include <map>

I tried the idea that the PRC can resonate f1 and f2 at the same time if the arm gives the reflection phase to f1 and f2 with the ratio of 1 vs 5.

The details are described on wiki. The point is this removes all of the PRC/SRC/asymmetry mumbo jumbo.

The calculated cavity lengths for f_mod of 11.065399MHz are:

• Arm Length: 37.7974 [m]

• PRC Length: 6.7538 [m]

• SRC Length: 5.39915 [m]

• Asymmetry (lx-ly): 0.0342 [m]

Here is the actual values derived from the photos.

• Arm Length: 37.54 [m] (0.26m too short)

• PRC Length: 6.760 [m] (6mm too long)

• SRC Length: 5.415 [m] (16mm too long)

• Asymmetry (lx-ly): 0.0266 [m] (8mm too long)

[Rana/Jamie/Kiwamu]

 Since we've got the PRMI locked we now should be able to do more qualitative measurements.

Here is a task list that we will measure/develop in the PRMI condition.

 - Optical gain measurements

 - Characterization of control loops

 - MICH and PRC calibrations

 - Noise budget

 - Development of automatic noise budget scripts

 - Arm loss measurement 

 - Shnupp asymmetry measurement

The attached plot shows 2 day trends of the PMC and MC reflected and transmitted power, the PSL POS/ANG QPD signals, and the temperature measured by the dust counter.

The power step in the middle of the plot corresponds to Koji/Jenne PMC realignment yesterday.

It looks like everything is following the day/night temperature changes.

Comparison between Hamamatsu S3399 and Perkin Elmer FFD-100

These are the candidates for the BB PD for the green beat detection as well as aLIGO BB PD for 532nm/1064nm.

FFD-100 seems the good candidate.

Basic difference between S3399 and FFD-100

- S3399 Si PIN diode: 3mm dia., max bias = 30V, Cd=20pF

- FFD-100 Si PIN diode: 2.5mm dia., max bias = 100V, Cd=7pF

The circuit at the page 1 was used for the amplifier.

- FFD-100 showed 5dB (= x1.8) larger responsivity for 1064nm compared with S3399. (Plot not shown. Confirmed on the analyzer.)

- -3dB BW: S3399 180MHz, FFD-100 250MHz for 100V_bias. For 30V bias, they are similar.

I think that the gain ramping time (_TRAMP) should be set to 1 second for all filter modules by default. We don't want them to switch instantaneously except in a few special cases.

So Jamie and I wrote a script (in scripts/general/) which sets all of these fields to 1 for a given system. The name of the system is an argument to the script. e.g.

>  setTRAMP LSC 1

The idea is that we set it once and then from then on, its captured by the autoBURT. Of course, we have to run this script each time we add new filter modules to a model.

Minicircuits ERA-5SM was used for the RF amp of the BBPD. This amp is promising as a replacement of Teledyne Cougar AP389
as ERA-5SM gave us the best performance so far among the BBPDs I have ever tested for the aLIGO BBPD/Green.

The -3dB bandwidth of ~200MHz and the noise floor at the shotnoise level of 0.7mA DC current were obtained.

1. Introduction

The aLIGO BBPD candidate (LIGO Document D1002969-v7) employs Teledyne Cougar AP389 as an RF amplifier.
This PD design utilizes the 50Ohm termination of the RF amp as a transimpedance resistance at RF freq.

However, it turned out that the bandwidth of the transimpedance gets rather low when we use AP389, as seen in the attachment2.
The amplifier itself is broadband upto 250MHz (the transfer function was confirmed with 50Ohm source).
The reason is not understood but AP389 seems dislike current source. Rich suggested use of S-parameter measurement
to construct better model of the curcuit.

On the other hand, the RF amplifiers from Minicircuits (coaxial type like ZFL-1000LN+), in general, exhibit better compatibility with PDs.
If you open the amplifier case, you find ERA or MAR type monolithic amplifiers are used.

So the question is if we can replace AP389 by any of ERA or MAR.

2. Requirement for the RF amp

- The large gain of the RF amp is preffered as far as the output does not get saturated.

- The amplifier should be low noise so that we can detect shot noise (~1mA).

- The freq range of the useful signal is from 9MHz to 160MHz.

The advanced LIGO BBPD is supposed to be able to receive 50mW of IR or 15mW of 532nm. This approximately corresponds to
5mA of DC photocurrent if we assume FFD-100 for the photodiode. At the best (or worst) case, this 5mA has 100% intensity modulation.
If this current is converted to the votage through the 50Ohm input termination of the RF amp, we receive -2dBm of RF signal at maximum.

This gives us a dilemma. if the amp is low noise but the maximum output power is small, we can not put large amount of light
on the PD. If the amp has a high max output power (and a high gain), but the amp is not low noise, the PD has narrow power range
where we can observe the shotnoise above the electronics noise.

What we need is powerful, high gain, and low noise RF amplifier!

Teledyne Cougar AP389 was almost an ideal candidate before it shows unideal behavior with the PD.
Among Minicircuits ERA and MAR series, ERA-5 (or ERA-5SM) is the most compatible amplifier.

Considering the difference of the gain, they are quite similar for our purpose. Both can handle upto -2dBm,
which is just the right amount for the possible maximum power we get from the 5mA of photocurrent.

3. Test circuit with ERA-5SM

A test circuit has been built (p.1 attachment #1) on a single sided prototype board. 

First, the transfer function was measured with FFD-100. With the bias 100V (max) the -3dB bandwidth of ~200MHz was observed.
This decreases down to 75MHz if the bias is 25V, which is the voltage supplied by the aLIGO BBPD circuit. The transimpedance
at the plateau was ~400Ohm.

Next, S3399 was tested with the circuit. With the bias 25V and 30V (max) the -3dB bandwidth of ~200MHz was obtained although
the responsivity of S3399 (i.e. A/W) at 1064nm is about factor of 2 smaller than that of FFD-100.

The noise levels were measured. There are many sprious peaks (possibly by unideal hand made board and insufficient power supply bypassing?).
Othewise, the floor level shows 0.7mA shotnoise level.

The emphasis of this annual safety audit  was on  safe  electrical housekeeping on March 3, 2011

The RF amplifier of the prototype BBPD has been replaced from ERA-5SM to MAR-6SM.
The bandwidth is kept (~200MHz for S3399 with 30V_bias), and the noise level got better while the maximum handling power was reduced.

MAR-6SM is a monolithic amplifier from Minicircuits. It is similar to ERA-5SM but has lower noise
and the lower output power.

The noise floor corresponds to the shotnoise of the 0.4mA DC current.
Now the mess below 50MHz and between 90-110MHz should be cleaned up.
They are consistently present no matter how I change the PD/RF amp (ERA<->MAR)/bias voltage.
I should test the circuit with a different board and enhanced power/bias supply bypassing.

Discussion on the RF power (with M. Evans)

- Assume 5mA is the maximum RF (~50mW for 1064nm, ~15mW for 532nm). This is already plenty in terms of the amount of the light.

- 100% intenisty modulation for 5mA across 50Ohm induces -2dBm RF power input for the amplifier.

- Assume if we use MAR-6 for the preamplifier. The max input power is about -18dBm.
  This corresponds to 16% intensity modulation. It may be OK, if we have too strong intensity modulation, we can limit the power
  down to 0.8mA in the worst case. The shot noise will still be above the noise level.

- In the most of the applications, the RF power is rather small. (i.e. 40m green beat note would expected to be -31dBm on the RF amp input at the higherst, -50dBm in practice)
So probably we need more gain. If we can add 10-12dB more gain, that would be useful.

- What is the requirement for the power amplifier?

• Gain: 10~12dB
• Output (1dBcomp): +3dBm +Gain (13dBm~15dBm)
• Noise level / Noise Figure: 3nV/rtHz or NF=14dB
  The output of MAR-6 has the votage level of ~7nV/rtHz. If we bring the power amplifier with input noise of ~3nV/rtHz,
  we can surppress the degradation of the input equivalent noise to the level of 10%. This corresponds to N.F. of 14dB.

Search result for Freq Range 10-200MHz / Max Gain 14dB / Max NF 15dB / Min Power Out 13dBm
GVA-81 is available at the 40m. ERA-4SM, ERA-6SM, HELA-10D are available at Downs.

Conversion between nV/rtHz and NF (in the 50Ohm system)

SN1: Connect signal source (50Ohm output) to a 50Ohm load.
Power ratio between the noise and the signal

SN1 = (4 k T (R/2)) / (S/2)^2

SN2: Connect signal source (50Ohm output) to an RF amp.
Only the voltage noise was considered.

SN2 = (4 k T (R/2) + Vn^2) / (S/2)^2

10 Log10(SN2/SN1) = 10. Log10(1 + 2.42 (Vn / 1nVrtHz)^2)

e.g.
Vn: 0 nVrtHz ==> 0dB
Vn: 0.5 nVrtHz ==> 2dB
Vn: 1 nVrtHz ==> 5dB
Vn: 2 nVrtHz ==> 10dB
Vn: 3 nVrtHz ==> 13.5dB

  1064 nm transmison were measured of 40m safety glasses as shown . Their performance did not degrade. They are as good as their labels.

I hope that new daqd code will fix the problem with non-aligned at 16 seconds frame file GPS times.

I have compiled new daqd program under /opt/rtcds/caltech/c1/core/release/build/mx and installed it under

target/fb/daqd, then restarted daqd process on "fb" computer. It was installed with the ownership of user root

and I did chmod +s on it (set UID on execution bit). This was done in order to turn on some code to renice daqd process

to the value of -20 on the startup. Currently it runs as the lowest nice value (high priority).

controls@fb /opt/rtcds/caltech/c1/target/fb $ ls -alt daqd
-rwsr-sr-x 1 root controls 6592694 Jun  2 10:00 daqd

Backup daqd is here:

controls@fb /opt/rtcds/caltech/c1/target/fb $ ls -alt daqd.02jun11
-rwxr-xr-x 1 controls controls 6768158 Feb 21 11:30 daqd.02jun11

Measurement of Schnupp asymmetry

This was done by measuring the relative phase between the sidebands reflected from the two arms while the arm cavities are locked.

The Schnupp asymmetry is measured to be:   L_sa = 3.64 ± 0.32 cm

Description:

As a phase reference we use the zero crossing of the response function for the out-of-phase control signal for the single arm cavity lock [0]. The difference in the RD rotation phase of the response zero crossings indicates the phase difference in the sideband signals reflected from the arms. Assuming the asymmetry is less than half the RF modulation wavelength [1], the asymmetry is given by the following formula:

       \Delta \phi   c   1 
L_sa = ----------- ----- -
           360     f_RSB 2

We use a LSC digital lock-in to measure the response of the arm cavity at a single-frequency drive of it's end mirror.

[0] The locations of the zero crossings in the out-of-phase components of the response can be determined to higher precision than the maxima of the in-phase components.

[1] f_RSB = 55 MHz,     c/f_RSB/2 = 2.725 m

Procedure:

1. Lock/tune the Y arm only.
   
   • We use AS55_I to lock the arms.
2. Engage the LSC lock-in.
3. Tune the lock-in parameters:

lock-in freq: 103.1313 Hz
I/Q filters:  0.1 Hz low-pass
phase:        0 degrees

4. Set as input to the lock-in the out-of-phase quadrature from the control RFPD.  In this case AS55_Q->LOCKIN.
5. Drive the arm cavity end mirror by setting the LOCKIN->Y_arm element in the control matrix.
6. Note the "RD Rotation" phase between the demodulated signals from the control PD (AS55)
7. For some reasonable distribution of phases around the nominal "RD Rotation" value, measure the amplitude of the lock-in I output.
   • Assuming the Q output is nearly zero, it can be neglected.  In this case the Q amplitude was more than a factor of 10 less than the I amplitude.
   • Here we take 5 measurements, each separated by one over the measurement bandwidth (as determined by the lock-in low pass filter), in this case 10 seconds.  The figure above plots the mean of these measurements, and the error bars indicate the standard deviation.

The data and python data-taking and plotting scripts are attached.

Error Analysis:

To to determine the parameters of the response (which we know to be linear) we use a weighted linear least-squares fit to the data:

y = b X

where:

X_0j = 1
X_1j = x_j              # the measurement points
y = y_i                 # the response
b = (b₀, b₁)     # line parameters

The weighting is given by the inverse of the measurement covariance matrix. Since we assume the measurements are independent, the matrix is diagonal and W_ii = 1/\sigma_i² The
estimated parameter values are given by:

\beta  =  ( X^T W X )^-1 X^T W y  =  ( X'^T X' )^-1 X'^T y'

where X' = w X, y' = w y and w_ii = \sqrt{W_ii}.

The X' and y' are calculated from the data and passed into the lstsq routine. The output is \beta.

The error on the parameters is described by the covariance matrix M^\beta:

M^\beta = ( X^T W X)^-1 = ( X'^T X')^-1

with correlation coefficients \rho_ij = M^\beta_ij / \sigma_i / \sigma_j.

The x-axis crossing is then given by:

X(Y=0) = - \beta₁ / \beta₀

References:

Valera's LLO measurement
http://en.wikipedia.org/wiki/Weighted_least_squares
http://en.wikipedia.org/wiki/Linear_least_squares_(mathematics)#Weighted_linear_least_squares
http://en.wikipedia.org/wiki/Error_propagation