The RF multiplexer is configured as shown in the figure. It is now effectively a 15x1 RF mux.

To select a required channel:

Run the script as shown below 

/opt/rtcds/caltech/c1/scripts/general/rfMux.py

>python rfMux.py ch11

For channel 10 to 16, you can just enter the required channel number and it is routed to the output.

For channel 1 to 8, you only need to input the required channel number as above. No need to run the code again to select ch9 after selecting ch1-8

How the NI-8100 controller works:

Whenever any channel of one switch is selected, the output of the other switch is set to its ch0 (ch1 and ch9 in the figure).

So selecting ch1-8 will automatically select ch9 as output for the other switch. IF you send a command to select ch9 afterwards, the first switch will be automatically set to ch1 and not stay on what you had selected before.

I finally made good on the LIFE TIME WARRANTY on the ancient, Jamie-made 1811 power supply with the faulty switch:

Back to fully working form.  Hopefully I'll still be around the next time it breaks in 16 years.

 Placed in PD cabinet in Y arm, next to the OTHER Jamie-made 1811 power supply from 1998.

We need to work farther on checking out the end transmission QPD electronics situation. 

In bullet-point form, we need to:

* Ensure that the Weiss QPD head modifications have been made on these diodes.  (cf. Rai W's LLO elogs on QPDs)

* Ensure that the QPD biases are somewhere in the range of 10-15V, not the old 100V.  (Because we only need HV to make the capacitance low for RF use. Low voltage means less power dissipation in the head)

* Ensure the Rana/Rob modifications have been propagated to the whitening boards, so that we have full dynamic range.  (Steve is looking for the marked up paper schematics)

* Replace signal path resistors with low noise metal film resistors.

* Check QPDs / whitening boards for oscillation (with a scope probe), ensure that we chose an appropriate analog gain.

In thinking about the transimpedances that we want, we thought about the current that we expect.  We should get about 100 mW of light transmitted through the ETMs when we have full IFO lock.  There is a 50/50 BS to split the light between the QPD and the Thorlabs transmission diode, so we have about 50 mW incident on the QPDs, which is about 13 mW per quadrant.  With a sensitivity of about 0.15 Amps/Watt for silicon, this means that we're expecting to see about 2 mA of current per quadrant once we have the IFO fully resonant. We want this to correspond to about 5V, which means we want a transimpedance gain of around 2.5 kOhm. 

For the things that need checking, each quadrant has:

Photodiode  ------  Gain Switch 1 ----- Gain Switch 2  ------ Gain Switch 3 ------ Variable Gain Amplifier ------- Whitening stage 1 (z @ 4 Hz, p @ 40 Hz)  ------- Whitening stage 2 (z @ 4 Hz, p @ 40 Hz)

We want to check on the status of each of these switches, and whether they actually do what they say on the QPD Head screens.  Q has checked out and fixed the bit outputs for the whitening stages, but the rest still needs to be checked out.  Also note that the Switch 1, Switch 2 and Switch 3 are common to all 4 quadrants (i.e. enabling switch 1 on one quadrant enables it on all quadrants), but the variable gains and the whitening stages are individual for each quadrant.

Today, me and Manasa went inside the lab to figure out a place for the place for the FC. The whole setup will be placed in a chassis box . The chassis in figure(setupforFC.pdf) will be placed in the highlighted(red) box in the figure(setup.png). All the cables will be routed to the computers from behind the box and the RF cables from the beat box will be routed from the front end of the box. The two raspberry Pi boxes will be placed inside the box and the Frequency counters will be mounted as shown so that the frequency count can be seen from outside. 

 [Nichin, Manasa]

AIM: Taking DC output readings with multimeter for each PD to create a database (required for transimpedance calculations), by taking off the table tops. Also, making sure each PD is illuminated properly.

What we did:

• In rack 1Y1: Diode laser controller was set to 150.0 mA at all times. This gave powers in the neighbourhood of 1mW at the end of fibers illuminating all PDs. The laser outputs light of 1064nm wavelength. The laser was switched off in the end.
• Checked the collimation of the fiber for each PD. In some cases they were not focused to give a sharp spot, so we had to unmount the fibers and fix it and mount them back. Manasa did it initially and I learnt how it was done properly. Eventually I got better and did it myself (under her supervision)
• Set the mount alignment for maximum illumination of the PD.
• Record the power falling on the laser and also the DC voltage output. Any light that did not come from my fiber was blocked when taking the readings and then unblocked. I also took care of offset voltage present when taking the DC readings.

Recorded measurements:

REFL11:   P_inc = 0.91 mW         VDC = 34.9 mV 

REFL33:   P_inc = 0.83 mW         VDC = 33.2 mV 

REFL55:   P_inc = 1.08 mW         VDC = 42.7 mV 

REFL165: P_inc = 0.79 mW         VDC = 115.3 mV

AS55:         P_inc = 0.78 mW         VDC = 31.3 mV

POX11:      P_inc = 0.83 mW         VDC = 34.7 mV

POP22**:   P_inc = 1.08 mW         VDC = 5.82 mV

POY11:      Not illuminated; there was no optical fiber mount. Although, there was a fiber near it with a cap on the end. It also looks like there is no space to put in a new mount near the PD. 

REF PD:    P_inc = 1.19 mW         VDC = 8.2 V     (REF PD = New focus 1611)

**Note: The current POP 22 PD does not have 2 different outputs for DC and RF signals. I unplugged the RF cable from the output, took readings with the multimeter and then plugged back the RF cable.

Further work:

I will calculate the responsivity for each PD and compare it to the expected values. 

[Rana, Jenne]

We have decided to keep better track (using new-fangled digital "computers") of our modifications to electronics boards. 

The idea will be to create a new DCC document for every electronics board (when we pull a board and modify it, it should receive this treatment) that we have, and that document will become a history of the board's life.  Version 1 will be a copy of the original drawing.  Version 2 should be a modified version of that drawing with the current situation.  All future versions should be modified from the most recent version, to reflect any changes.  Notes for each updated version should include an elog reference to the work, so that we know why we did things, and have a place to find photos of the actual modifications.  Elogs should also include a link to the DCC version.  DCC titles should include the phrase "40m Revisions" for ease of searching.

Patient Zero for this new system will be the PMC servo card.  The DCC number is D1400221.  As of this moment, this just has the V1 original drawing with no modifications.

This has been included in the 40m's DCC document tree that Jamie started back in November 2012.

The following values are going to be entered in the param_[PDname].yml file for each PD. I am elogging them for future reference.

I got the values from combing schematics and old Elog entries. Please let me know if you believe the values are different.

• AS55: 66.2 ohms
• REFL11 : 66.2 ohms 
• REFL33 : 50.2 ohms
• REFL55: 50 ohms (Elog 4605)
• REFL165: 50.2 ohms
• POY11: 66.2 ohms
• POX11: 50.2 ohms
• REF (NF1611): 700 ohms
• POP22: ?? (This is currently a Thorlab BBPD )
  

I have configured a NEW Prologix GPIB-Ethernet controller to use with HP8591E Spectrum analyzer that sits right next to the control room computers.

Static IP: 192.168.113.109

Mask: 255.255.255.0

Gateway: 192.168.113.2

I have no clue how to give it a name like "something.martian" and to update the martian host table (Somebody please help!!)

The instructions for adding a name to the martian DNS table are in the wiki page that I pointed you to:

https://wiki-40m.ligo.caltech.edu/Martian_Host_Table

The instructions at https://wiki-40m.ligo.caltech.edu/Martian_Host_Table   are outdated!

The name server configuration is currently at /etc/bind/zones/martian.db [ source: elog:10067 ]

Anyway, I need superuser access to edit the files, which I don't have. Even if I did know the password, I don't think it's a good idea for me to be messing around. So any of the 40m folks please update the martian table to include:

santuzza.martian  192.168.113.109

Main Problem:

The frequency counter (FC) takes in an analog RF input(signal) and outputs the frequency of the signal(Ranging from 1 MHz- 6000 MHz) in the digital domain (into a processor). The FC samples the data with a given sample rate( user defined) which ranges from 0.1 s to 1 s(faced problems in fixing this initially).  For data acquisition, we have been using a Raspberry Pi(as a processor) which is connected to the martian network and can communicate with the computers inside the 40m.  The ultimate challenge which I faced( and been knocking my head off from past two-three weeks) is the synchronization of clocks between the Raspberry Pi and the FC i.e the clock which the FC uses to sample and dump data( every 'x' s) and the clock inside the raspberry pi( used  in the loop to wait for a particular amount of time the frequency counter takes to dump successive data).

Steps Taken:

• To address this problem, first I added an external clock circuit which monitors the Raspberry Pi and the FC to dump and read data at a particular rate(which is equal to the sampling rate of the FC)In detail: http://nodus.ligo.caltech.edu:8080/40m/10129. 
• While doing so, at first the level trigger algorithm was used which means that the external clock frequency was half as that of  the reciprocal of the sampling rate and a trigger was seen every time the level shifts from +DC to -DC(of the external square wave).
• But this did not completely mitigate the issues and there were still few issues on how quickly the ADC reads the signal and R Pi processes it.
• To minimize these issues completely, an edge trigger algorithm which detects a pos edge(rising)  of the clock was used. The clock  frequency is now equal to the reciprocal of the sampling rate. This algorithm showed better results and greatly minimized the drift of the sampling time.

Psuedo Code(code attached):

open device : FC via USB-HID;

open device : ADC via I2C;

always(for t= recording time):

            read data from ADC(external clock);

            if pos edge detected:

                    read data from FC and store it in a register;

             else read data from ADC;

end

write data stored in the register to a file( can be an Epics channel or a text file);

Results:

The attached are the plots showing the time between samples for a large number of samples taken for different sampling times of the FC. The percentage error is the percentage of standard error in the timing between two samples for the data for the entire measurement. It can be inferred that this error has been cut down to the order of ms.

To do next:

• I have started taking phase measurements( analysis and plots will follow this elog) and also the PSD plots with the improved timing characteristics.

Looks good. Now you have the internal timer to verify the external clock.
If you can realize the constant rate sampling without employing the external clock, that's quite handy.

 [Nichin, Jenne]

The martian lookup tables are located at /etc/bind/zones/martian.db  and etc/bind/zones/rev.113.168.192.in-addr.arpa

Jenne updated these to include santuzza.martian  192.168.113.109

The method to restart named server given at  https://wiki-40m.ligo.caltech.edu/Martian_Host_Table  also does not work.

I restarted it using  >sudo /etc/init.d/bind9 restart

The named server is now updated and works fine. :)  I will update the 40m wiki now.

The attached are the PSD plots with improved FC timing(with the same code as in http://nodus.ligo.caltech.edu:8080/40m/10151). More plots(Phase and PSD) to follow.

 I went into the lab and connected the RF cables to the Mux. Will take measurements for each PD henceforth.

A test run was conducted on the PDFR system last afternoon and transimpedance plots were generated for 6 of the PDs. The laser was shut down after the test run.

I have not verified (yet) if the transimpedance values indicated by the plots are correct or not. The values mostly look INCORRECT. But the peaks are exactly where they need to be. *phew!*

Reasons: Incorrect calibration, Light other than from the PDFR system fibers on the PDs

Will have to work on debugging all this.

Log-log ... 

I found the VCO driver, that Rana asked me to locate, inside the 40m. I already have one VCO from PSL lab. Now, I have kept both of them inside the 40m lab(one on the cart in the side of the Y-arm and the other near the X-arm electronics table).

Goal:

To estimate the transfer function and the noise in the FC that is a part of the FOL-PID loop.

Measurements Taken:

The setup used for the measurements is described in my previous elogs.

The input modulation signal and the FC output were recorded simultaneously for a certain period of time and the phase and gain are estimated from the data.

Analysis(Data and code attached):

The recordings must contain equal number of data points(around 6000 data points in my measurements) for analysis.

The steps I followed to generate these plots are:

• Took the FFT of both FC out data(from FC) and Modulation input(from SRS via ADC).
• Estimated the phase angles at the particular modulation frequencies from the FFT data(in Matlab using  angle(x) for phase at the frequency f(x);x: is the frequency bin)
• Then for the phase of the system at a particular modulation frequency, 

                              Phase(system) =Phase(FC Signal) - Phase(Input Signal)

• Plotted the acquired phase vs the modulation frequency on a Semi-log graph.

Results:

From the plots its can be inferred that :the delay of the FC is almost 0 until the modulation of 0.1 Hz. Then there are phase shifts of  +/- 180 degrees showing that the system has multiple poles and zeroes(will be estimated after I have phase plots at few more carrier frequencies).

To Do Next: 

Phase plots for varying carrier frequencies and different sampling times.

Installation of FC inside the 40m.

If I assume 1sample delay for 0.1s sampling rate, the delay is Exp[-I 2 pi f T], where T is the sampling period.

This means that you expect only 36 deg phase delay at 1Hz. In reality, it's 90deg. Huge!

Also there are suspicious zeros at ~1.6Hz and ~3Hz. This may suggest that the freq counter is doing some
internal averaging like a moving average.

It would be interesting to apply a theoretical curve on the plot. It's an intellectual puzzle.

I took back he VCO driver that Reetika brought over to the 40m from the PSL lab.  

In a attempt to debug the values of transimpedance generated by the PDFR system, I did a manual measurement for REFL11 PD.

• Took the tops off AS and POY tables. (REFL11 and REF PD) Under the supervising eye of Manasa
• Verify that no extra light is falling on REFL11.
• Retake DC voltage readings, power readings.
• Manually set the sweep parameters and record readings from network analyzer.
• Put the tops back on the tables
• Calculate transimpedance 

Results:

REF PD(1611):

Pinc = 1.12 mW                 T_dc = 10000 V/A (datasheet)

Vdc = 7.68 V                      T_rf = 700 V/A (datasheet)

Calculated Responsivity = 0.68 A/W (Which matches perfectly with the datasheet value of 0.68 A/W) 

REFL11:

Pinc = 0.87 mV             T_dc = 66.2 V/A (schematic)

Vdc = 32.5 mV      

Calculated Responsivity = 0.56 A/W

Network analyzer reading at 11 MHz : 0.42

Calculated RF Transimpedance = 460 V/A

40m Wiki : RF Transimpedance = 4 kV/A

I ran the same measurement using PDFR system and got the same results.

Attached: the automatic data and plot obtained.

Conclusion:  The PDFR system and manual measurements agree with each other. However the values do not match with 40m Wiki. I have no clue about which measurement is correct or any mistakes I might be making in the calculations. 

What is the coupling factor between the RF in and the RF mon of the demodulator?
I don't assume you have the same amount RF power at those two points unless you have an RF amplifier in the mon path.

Rack 1Y2, I took transfer function measurements for each of the following demodulator boards: REFL11, REFL33, REFL55, REFL165, AS55, POP22, POX11 and POY11.

What I did:

1) Removed the wire at PD Input to demodulator board.

2) Put the MOD output from network analyzer into PD input of board.

3) Ran a sweep from 100kHz to 100MHz.

4) Measured the transfer function between PD RF MON and PD Input. (The PD RF MON signal came out of the RF multiplexer, so the mux is included as well )

5) Put the original wire back at PD Input.

Results:

The plots clearly show an attenuation of 20dB (factor of 10) for all the demodulator boards. This explains why my transimpedance measurements are off by 10 times.

Note: for REFL 165, there was an extra 100MHz high pass filter installed at PD Input. I did not remove this and made my measurements along with this.

To Do:

a) Modify the PDFR system to calibrate out this attenuation.

b) Measure the transfer function between the input and output of RF mux, so that we can have just the transfer function between PD input an PD RF MON (for documentation's sake)

 Thanks Koji , for your  hint for the brain teasing puzzle. I was looking into Filters that are usually used in devices like counters, DSO and other scopes. I found that , to improve the quality of the measurement one of the best approach  is averaging. I looked deeper into averaging and found out this:

There are two general use-cases for averaging . The first, successive sample averaging, takes a single acquisition and averages between its samples. The second, successive capture averaging, combines the corresponding  samples of multiple captures to create a single capture. Successive sample averaging is also called boxcar filtering or moving average filtering. In an implementation of this type of averaging each output sample represents the average value of M consecutive input samples. This type of averaging removes noise (one of the reasons the noise level was not bad: http://nodus.ligo.caltech.edu:8080/40m/10151) by decreasing the device's bandwidth(could be one of the reasons why the FC operates in 4 different frequency ranges). It applies an LPF function with a 3dB point approximated by  0.433 * s / M, where M is the number of samples to be averaged, and s is the sample rate in samples per second. 

Now I tried verifying the 3 dB points in the gain plots I generated :

For 1 s Sampling time : the 3 dB point for such a Boxcar filter should be at 0.433* 1/M. If we assume that it averages for 2 samples, M=2 which gives the 3dB point at 0.288 Hz but occurs somewhere between 0.3 and 0.4 Hz.  (http://nodus.ligo.caltech.edu:8080/40m/140619_120548/GainVsFreq.png)

For 0.1s Sampling time: the 3dB point should be at 2.17 Hz and in reality is 2.5 Hz(http://nodus.ligo.caltech.edu:8080/40m/140701_211904/gain.png).

Also, This type of filter will have very sharp nulls at frequencies corresponding to signals whose periods are integer sub-multiples of M/s. As seen my previous plots (http://nodus.ligo.caltech.edu:8080/40m/10118 , http://nodus.ligo.caltech.edu:8080/40m/10070) there are sharp nulls at frequencies

0.4 Hz for 1S sampling time and

at 1.5 Hz,3 Hz for 0.1 S sampling time as correctly predicted.

The moving average filter is  L-sample moving average FIR, with the frequency response as:   H(ω) = (1/L) (1 − e^{− jω L})/(1 − e^{− jω})..

There is an overall delay of (M - 1)/2 samples from such a length-M causal FIR filter. 

The expected bode plots for such a filter with L= 5 is attached(attachment 2).

Patient One for this new system will be the MC servo card. The DCC number is D1400242. Currently, v1 is just the original drawing with no modifications. I've updated the DCC document tree at E1400326 accordingly.

It looks like we can use Jenne's information in 40m:9892 to deduce the modifications that have been made (alternatively, someone can just pull the board and examine it on the bench).

The expected bode plots for such a filter with L= 4 is attached and compared with the measured.

RXA: When comparing two things, please put them onto the same plot so that they can be compared.

The attached zip file has a modified schematic of the MC servo card (011/MC), as deduced from Jenne's photos. Someone should go through and verify that the schematic is correct. Then it can go on the DCC as D1400242-v2.

To modify the schematic, I used Inkscape (the svg files for each sheet are included in the zip file). Then to generate the pdf, I ran

for i in sheet*.svg; do inkscape -A "${i/svg/pdf}" "$i"; done

pdftk sheet*.pdf cat output D1400242

I repeated the exact steps above and made sure everything was back where it should be after I was done.

Reason I had to retake the measurements:

My script for acquiring data from the AG4395A network analyzer was such that it first acquired the magnitude data from channel 1 and then recorded phase data from channel 2 without holding its trace. Hence the phase and magnitude data were not exactly in sync with each other. So, when I tried to fit the data to a model using vector fitting, I ended up with very bad results.

I have now changed every single script relating to the network analyzer to just get the real and imaginary data in one go and then calculate the phase using this data.

The fitting process is now in progress and results will be up shortly.

 As a part of installation of two(X-ARM and Y-ARM) frequency counters in the 40m, I have tested their performance when using them both on a single Raspberry Pi. The timing plots are attached. There are almost no timing issues in this configuration and it can be said that there is no harm using both of the FCs on the same platform.

We will be installing the FC box inside the lab and carry out few tests with RF mon beat note inputs.

The plots in the previous Elog includes delay and a little attenuation by RF cables and the RF mux.

Today I separately calculated the delay and attenuation for an RG405 cable (550 cm) and the RF mux(using really small RF cables). These delays should be accounted for when fitting the transfer function of Demodulator boards and transimpedance of PDs.

The plots are in both semilogx and linear.

 A time delay can be modeled as the exponential transfer function :  e^(-sT_d)  as seen HERE . Therefore the slope of the phase gives us the time delay.

A RG405 coaxial cable, exactly 5.5 meters in length, was fit to an ideal delay function e^(-sT_d) , with T_d = 150 ns.

The plots shows the actual data, fit data and data after correction using the ideal model stated above.

Conclusion:

Delay in RG405 cables is approximately 27.27 ns per meter. This value can be used to correct the phase in measurements of transimpedance for each PD by dividing out the ideal transfer function for time delay.

[EDIT: This looks like we have about 12 % the speed of light inside the RF cables. Too small to be true. I will check tomorrow if the Network analyzer itself has some delay and update this value.]

The varying attenuation of about 1dB due to the cable is not compensated by this. We need to separately include this.

Things to do:

1) Get the length of RF cables that is being used by each PD, so that the compensation can be made.

2) Calculate the attenuation and delay caused by RF multiplexer and Demodulator boards. Include these in the correction factor for transimpedance measurements. 

A time delay can be modeled as the exponential transfer function :  e^(-sT_d)  as seen HERE . Therefore the slope of the phase gives us the time delay.

The transfer function of RF multiplexer in rack 1Y1 (NI PXI-2547) was fit to an ideal delay function e^(-sT_d) , with T_d = 59 ns.

The plots shows the actual data, fit data and data after correction using the ideal model stated above.

Conclusion:

Delay the RF Multiplexer is approximately 59 ns. This value can be used to correct the phase in measurements of transimpedance for each PD by dividing out the ideal transfer function for time delay.

 I used vector fitting to fit the transfer functions between RF input and PD RF MON of demodulator boards. These fittings can certainly do a lot better on LISO, but for the time being I will assume these to be good enough and change the main PDFR scripts to calibrate out this factor and get a decent reading of PD transimpedance. Then it will just be a matter of changing the transfer function parameters. A lot of work needs to be done on the PDFR interface and plot features.

Attached: The plots showing data and fits.

 Koji asked me to get the calibration of the PZT counts to Volts for the the X and Y ends. Yesterday, I went inside the lab and took some measurements from the digital readout of the PZT by giving in a DC offset(-5 to +5 volts) to PZT_Out and read out from these channels:

For X-end:  C1:ALS-X-SLOW_SERVO1_IN1

For Y-end:  C1:ALS-Y-SLOW_SERVO1_IN1

Since a 20dB attenuator was placed in the path of X-arm readout while taking the Transfer functions(Detail), I did the calibration measurements without removing it from the path. However, for the Y arm there was no attenuator in the readout path.

The obtained calibration values are :

X- arm PZT : [146.3 +/- 2.37 ]  counts/Volt 

Y- arm PZT :  [ 755.1 +/- 3.6]    counts/Volt

The attached are the fit and data plots for the above calibration.

1) Don't be brainless. Redo the fitting of the Y arm. Obviously the fit is not good.

2) How can you explain the value from the ADC bit and range?

e.g. +/-10V range 16bit ADC => 2^16/20 = 3276.8 count/V

 The PZT seems to saturate at around +/- 3500 counts. So for the Y arm, I excluded the saturated points and fitted the data points again.

As for the calibration number, we expect the 3276.8 count/V for +/- 10 V range of a 16 bit ADC but the number is ~800 count/V. I couldn't figure out a reason why the number is so different.

The new calibration values are :

X- arm PZT : [146.3 +/- 2.37 ]  counts/Volt   (with a 20 dB attenuator included in the path)

Y- arm PZT :  [ 797 +/- 3.6]    counts/Volt  

I will get the calibration in MHz/V of PZT actuation and check whether these numbers make any sense.

 The PZT actuation on the laser frequency in MHz/V ( assuming the previous calibration here of the PZT count/V) is :

X- arm: 33.7 MHz/V

Y- arm: 14.59 MHz/V

This number seems to be wrong by a factor of 10. 

So we[I and EricQ] decided to trace the cables that run into the ADC from the PZT Out. We found a black LEMO box in the path to ADC,which is  an anti-aliasing filter for each input channel. However,in theory the response of this filter should be flat up until a few kHz i.e. for  the DC gain it should be 1. But we will manually test it and look at the DC gain of the LEMO box.

Sitting down to work on the IFO, I couldn't lock the Yarm.  I looked at the error signal as well as the transmission on Dataviewer, as usual, and saw that the POY error signal was almost non-existant. 

Since there was work on the POY table today (Steve removed the oplev test setup, elog 10489 and Q centered the SRM oplev after doing SRMI alignment, no elog yet), I went out to have a look at the table. 

There was nothing occluding the POY beam, which I traced back to the edge of the table.  The beam looked nice and round, so I decided that wasn't it.  I jiggled the PD cables, and lo and behold, the POY RF out cable almost came off in my hand it was so loose.  My suspicion is that whomever was the last to put the POY RF out back didn't tighten the cable and then the work today jiggled the cable loose.  I tightened the cable, and by the time I was back to the control room the arm was locked and Koji was already running the alignment scripts.

In April, Koji logged that he had made some changes to the Yend QPD whitening board (elog 9854).  Today, I pulled the Xend board to see if it had the same modifications.  The filter shapes all seem to be the same (as in, the capacitors at the output filters were removed, etc.), and the final gain is the same.  I just realized that I didn't explicitly check if the whitening switches were pulled to ground to permanently turn on the whitenening, but hopefully I'll be able to see that in the photo. 

I have not made any changes today (yet) to the board, so the overall gain is still accessible via EPICS.  I wanted to do a quick check that we won't be saturating things at full power with the maximum gain, before I make a change.

EDIT:  After comparing the photos here and in elog 9854, the X end board has the filter shape modifications that were done some time ago, but the whitening is not permanently enabled.  For the Yend board, Koji added a jumper wire connecting (for example) R97 and R106 to the grounded side of C69.  This jumper wire is not in place on the X qpd board.

Before I re-pull the board and modify it, I want to make sure I know what I'm going to do for the gain slider override.

Here is a little PDF of what I plan to do to both of the transmission QPD whitening boards later today.  The idea is to take away the remote gain slider inputs, and force the gains to always be at +30dB.

The red and blue notes are from Koji's elog 9854, and the green are my plans for today. 

I will cut the traces from the gain slider inputs, and pull the negative input of the AD620 to ground.  The positive input will be connected to the +5 voltage line, with a divider so that the positive input to the AD620 is about 666mV. 

The AD602 will be maxed out at +30dB with anything over 625mV. 

Unless there are objections, I will start these modifications in an hour or so.  I will also make the Xarm whitening always-on, just like Koji has already done for the Yend.

EDIT, JCD, 12Dec2014:  These are not the modifications that were made.  Please see ____ for actual modifications.

Okay, I have finished modifying the Xend QPD whitening board, although I will likely need to change the gain on Monday.

Rather than following my plan in elog 10782, I removed the AD602's entirely, and just use the AD620's as the amplifiers.  We don't need remotely adjustable gains, and the AD620s are a less noisy part.

I set the gain to be 30dB using a 1.65k resistor for R_G, which turns out to be too high.  After I installed the board and realized that my counts were much higher than they used to be, I realized that what we had been calling +30dB was in fact +13.2dB. ( I am assuming that the ADC for the gain sliders were putting out a maximum of +10V.  The AD620 used to have a 1/10 voltage divider at the input, and an overall gain of 1, so the output of the AD620 was 100mV.  This goes into pin 16 of the AD602, which has gain of 32*V_set + 10.  Which gives 32*0.1+10=13.2dB.  Ooops.  We've been lying to ourselves. )

Anyhow, before I made the gain realization, I was happily going along, setting the AD620s' gains all to 30dB. I also copied Koji's modification from April of this year, and permanently enabled the whitening filters.

Here is the schematic of what ended up happening.  The red modifications were already in place, and the greens are what I did today.

You can see the "before" picture in my elog Wednesday, elog 10774.  Here is an "after" photo:

Here is a spectrum comparing the dark noise of the Xend QPD after modification to the current Yend QPD (which is still using the AD602 as the main instrumentation amplifier).  I have given the Yend data an extra 16.8dB to make things match.

And, here is a set of spectra comparing both ends, dark noise versus single arm lock.  While I'll have to sacrifice a lot of it, there's oodles more SNR in the Xend now.  The Yend data still has the "gain fixing" extra 16.8dB.

The Xend quadrant input counts (before the de-whitening filters) now go up to peak values of about 1,000 at single arm lock.  If (optimistically) the we got full power recycling and the arms got to powers of 300, that would mean we would have 300,000 counts, which is obviously way more than we actually have ADC range for.  Currently, the Yend quadrant input counts go as high as 50, which with arm powers of 300 would give 15,000 counts.  I think I need to bring the Xend gain down to about the level of the Yend, so that we don't saturate at full arm powers.  I can't remember right now - are the ends 14-bit or 16-bit ADCs?  If they're 16-bit, then I can set the gain somewhere between the current X and Y values.

 Finally, I added a section of the 40m's DCC document tree for the QPD whitening:  E1400473, with a page for each end.  Xend = D1400414, Yend = D1400415.

 16 bit. There aren't any 14-bit ADCs anywhere in LIGO. The aLIGO suspensions have 18-bit DACs.

The Y-End gains seem reasonable to me. I think that we only use TRX/Y as error signals once we have arm powers of >5 so we should consider if the SNR is good enough at that point; i.e. what would be the actual arm motion if we are limited only by the dark noise?

I seem to remember that the estimate for the ultimate arm power is ~200, considering that we have such high losses in the arms.

Details later - empty entry for a reply.

Short story - Yend is now same as Xend filters-wise and lack of gain sliders -wise.  Both ends have 13.7k resistors around the AD620 to give them gains of ~4.5.

Xend seems fine.

Yend seems not fine.  Even the dark noise spectrum sees giganto peaks.  See Diego's elog 10801 for details on this investigation.

 [Jenne, Rana, Diego]

We did some test on the modified QPD board for the Yend; we saw some weird oscillations at high frequencies, so we went and check more closely directly within the rack. The oscillations disappear when the cable from the QPD is disconnected, so it seems something is happening within the board itself; however, looking closely at the board with an oscilloscope in several locations, with the QPD cable connected or disconnected, there is nothing strange and definitely nothing changing if the cable is connected or not. In the plots there are the usual channels we monitor, and the 64kHz original channels before they are downsampled.

Overall it doesn't seem being a huge factor, as the RMS shows at high frequencies, maybe it could be some random noise coming up, but anyway this will be investigated further in the future.

Here's a summary of the changes made to the D990511 serial 115 (formerly known as REFL 33), as well as a short procedure. It needed tuning to 29.5MHz and also had some other issues that we found along the way. 

So here's a picture of it as built:

The changes made are:

1. U11 and U12 changed from 5MHz LP to 10 MHz LP filters.

2. Resistors R8 and R9 moved from their PCB locations to between pins 1 (signal) and 3 (ground) of U11 and U12, respectively. These were put in the wrong place for proper termination so it made sense to shift them while I was already replacing the filters.

Also, please note- whoever labeled the voltages on this board needed an extra cup of coffee that day. There are two separate 15V power supplies, one converted from 24V, one directly supplied. The directly supplied one is labeled 15A. This does NOT mean 15 AMPS.

Transfer functions:

Equipment: 4395A, Signal generator (29.5 MHz), two splitters, one mixer

You can't take the TF from PD in to I/Q out directly. Since this is a demod board, there's a demodulating (downconverting) mixer in the I and Q PD in paths. Negligible signal will get through without some signal applied to the L input of the mixer. In theory, this signal could be at DC, but there are blocking capacitors in the LO in paths. Therefore, you have to upconvert the signal you're using to probe the board's behavior before it hits the board.  Using the 4395A as a network analyzer, split the RF out. RFout1 goes to input R, RFout2 goes to the IF port of the mixer. Split the signal generator (SG). SG1 goes to LO in, SG2 goes to the L port of the mixer. The RF port of the mixer (your upconverted RFout2) goes to PD in, and the I/Q out goes back to the A/B port of the 4395A - at the same frequency as the input, thanks to the board's internal downconversion. 

Phase measurement:

Equipment: Signal generator (29.5 MHz), signal generator (29.501 MHz), oscilloscope

Much simpler: 29.5 MHz to the LO input (0 dBm), 29.501 MHz to the PD input (0 dBm), compare the phases of the I/Q outputs on the oscilloscope. There are four variable capacitors in the circuit that are not on the DCC revision of the board - C28-31. On the LO path, C28 tunes the I phase, C30 tunes the Q phase. On the PD path, C29 and 31 appear to be purely decorative - both are in parallel with each other on the PD in Q path, I'm guessing C29 was supposed to be on the PD in I path. Fortunately, C28 and C30 had enough dynamic range to tune the I/Q phase difference to 90 degrees.

Before tuning:

After tuning:

The RF amplifier panel on the IOO rack (Attachment 1) will be used to also hold the RF amplfiers for the frequency counters. The amplifiers mounted on it right now are getting +15 (orange wire) and GND (black wire) from the rack power supply (Attachment 2).

Proposed plan to install RF amplifiers:

1. Disconnect the D sub connector that powers the amplifiers and pull out the panel. The rack power supplies will NOT be shut down for this. 

2. Mount the RF amplifiers with bypass capacitors. There will be 4 amplifiers ZFL-500LN mounted on the same panel (2 for each frequency counter).

3. While putting back the panel on the IOO rack, we will need to shut down the +15V and -15V sources to connect the amplifiers to the rack power supply.

I will do this over this weekend so that we dont lose any locking time. If anybody has any concerns, let me know

I have re-enabled the second whitening stage switching on each quadrant of each end's QPD whitening board, to try and avoid saturations at full power. Looking at the spectra while single arm locked, I confirmed that the FM2 whitening switch works as expected. FM1 should be left on, as it is still hard-wired to whiten. 

The oscillations in the Y QPD still exist. Jenne is updating the schematics on the DCC.

Went to zero CARM offset on ALS; transmission QPDs are still saturating :(

Maybe we need to switch off all whitening.